JP4355712B2 - Switching power supply - Google Patents

Description

  The present invention relates to a switching power supply device configured to extract a switching output obtained by switching a DC input voltage to an output winding of a power conversion transformer.

  Conventionally, various types of switching power supply devices have been proposed and put into practical use. Most of them are systems in which a DC input voltage is switched by a switching operation of a switch circuit connected to an input winding of a power conversion transformer and a switching output is taken out to an output winding of the power conversion transformer. Along with the switching operation of such a switch circuit, the voltage appearing in the output winding is rectified by the rectifier circuit, then converted into direct current by the smoothing circuit and output.

  In this type of switching power supply device, an output rectifier element such as an output rectifier diode is connected in series with the power transmission line in the rectifier circuit. Therefore, reducing the loss in the output rectifier diode is extremely effective in improving the efficiency of the switching power supply device.

  In order to reduce the loss in the output rectifier diode, a diode having a small forward voltage drop may be used. However, a diode with a low forward voltage drop has a low reverse withstand voltage. For this reason, when a diode with a low forward voltage drop is used as the output rectifier diode, it is particularly necessary to suppress the reverse voltage.

  In this type of switching power supply, the most important thing to consider as the reverse voltage is the surge (spike) voltage caused by the parasitic elements accompanying the on / off operation of the switch circuit. It is applied as a reverse voltage. Therefore, various attempts have been made to suppress such a surge voltage.

  For example, the present applicant has proposed a snubber circuit using LC resonance in Patent Document 1. According to this snubber circuit, the surge voltage can be suppressed to a predetermined voltage or less by using LC resonance.

  Patent Documents 2 to 4 also disclose a switching power supply device provided with a circuit for suppressing the surge voltage.

Japanese Patent No. 3400443 US Pat. No. 5,198,969 US Pat. No. 6,466,459 US Pat. No. 6,650,551

  Here, the predetermined voltage in Patent Document 1, that is, the maximum value (peak value) of the surge voltage to be suppressed is 4 × Vin / value as described in paragraphs [0062] to [0065] of the same document. n (Vin; DC input voltage, n: ratio of the primary side winding and the secondary side winding of the power conversion transformer). This value is for the center tap type rectifier circuit. When the rectifier circuit is a full bridge type, this value is half of this value, that is, 2 × Vin / n. As described above, according to the snubber circuit of Patent Document 1, it is possible to suppress the surge voltage to some extent, but there is still room for improvement in suppressing the maximum value.

  In the circuits disclosed in Patent Documents 2 to 4, the parasitic inductance and parasitic capacitance of the wiring are not taken into consideration. Moreover, the rise of the voltage generated on the secondary side of the power conversion transformer is steep. Therefore, when the wiring on the secondary side of the power conversion transformer is long, for example, when the parasitic inductance or capacitance of the wiring is large, the energy stored in the parasitic inductance increases, and the surge when the energy is released The voltage will increase. That is, with this circuit, it is difficult to sufficiently suppress the surge voltage without depending on the device configuration.

  The present invention has been made in view of such problems, and an object of the present invention is to provide a switching power supply device that can more effectively suppress a surge voltage applied to an output rectifying element without depending on the device configuration. It is to provide.

The switching power supply device of the present invention includes four switching elements, and includes a full bridge type bridge circuit that generates an input AC voltage based on a DC input voltage, a primary side winding, and a secondary side winding. A transformer that transforms the input AC voltage to generate an output AC voltage, and is provided on the secondary side of the transformer and includes a plurality of first rectifier elements, and the plurality of first rectifier elements An element formed by rectifying the output AC voltage to generate a DC output voltage, and a parallel connection of a second rectifier element and a first capacitor element connected in parallel to the bridge circuit and connected in reverse direction A surge voltage suppression circuit including two pairs; a resonance inductor that forms a first resonance circuit together with the first capacitor; and a driving circuit that drives the bridge circuit. With the door, but the recovery time of the resonance time and the first rectifier element of the first resonant circuit and is set so as to satisfy the following condition (1). However, {2π × (L × C) ^1/2 } is a resonance time for one cycle in the first resonance circuit, L is an inductance of the resonance inductor, and C is a capacitance value of the first capacitor element. , Trr1 is the recovery time of the first rectifying element.
1/4 × {2π × (L × C) ^1/2 }> Trr1 (1)

  In the switching power supply device of the present invention, an input AC voltage is generated from a DC input voltage input to the bridge circuit, and an output AC voltage is generated by transforming the input AC voltage by a transformer. The output AC voltage is rectified by the first rectifier element in the rectifier circuit and output as a DC output voltage. In addition, the first capacitive element and the resonance inductor cooperate to function as an LC series resonance circuit (first resonance circuit), so that a resonance operation is performed between them. Here, since the resonance time of the first resonance circuit and the recovery time of the first rectifying element are set so as to satisfy the conditional expression (1), the rise of the reverse voltage applied to the first rectifying element is Regardless of the configuration, it becomes more gradual compared to the prior art.

In the switching power supply device of the present invention, it is further preferable that the resonance time of the first resonance circuit and the recovery time of the second rectifying element are set so as to satisfy the following conditional expression (2). However, Trr2 is the recovery time of the second rectifying element. When comprised in this way, the rise of the reverse voltage added to a 2nd rectifier becomes loose, and the raise of the surge voltage in a 2nd rectifier is suppressed.
1/4 × {2π × (L × C) ^1/2 }> Trr2 (2)

  In the switching power supply device of the present invention, in the surge voltage suppression circuit, two element pairs may be connected in series with each other. In this case, the resonant inductor may be arranged on the primary side of the transformer, and the primary side winding of the transformer is connected to one of the two switching elements of the four switching elements. An H-bridge connection is made to one bridge circuit composed of a switching element and two element pairs, and the resonance inductor is connected to the other two switching elements and two element pairs connected in series among the four switching elements. H bridge connection may be made to the other bridge circuit composed of The resonance inductor may be arranged on the secondary side of the transformer.

  In the switching power supply device of the present invention, the transformer and the resonance inductor may be provided magnetically independent from each other. Further, an auxiliary winding may be provided on the primary side of the transformer, and the auxiliary winding and the resonance inductor may share a magnetic flux.

  The switching power supply device of the present invention includes a second capacitor element connected in parallel to each of the four switching elements, and the resonance inductor forms a second resonance circuit together with the second capacitor element. Also good. When comprised in this way, the short circuit loss in a switching element is suppressed by resonance operation | movement by a 2nd resonance circuit. Further, the switching element may be constituted by a field effect transistor, and the second capacitor element may be constituted by a parasitic capacitance of the field effect transistor. When configured in this way, the number of elements used is reduced, and the circuit configuration is simplified. Further, the first rectifying element may be constituted by a parasitic diode of a field effect transistor.

  In the switching power supply device of the present invention, the rectifier circuit may be a center tap type configured to include two first rectifier elements, or a full bridge type configured to include four first rectifier elements. Is possible. In this case, the maximum value (peak value) of the surge voltage is, for example, about 2 × Vin / n in the case of the center tap type, and about 1 × Vin / n in the case of the full bridge type. It can be made lower than in the past.

  According to the switching power supply device of the present invention, the first resonance circuit is constituted by the first capacitor element and the resonance inductor, and the resonance time of the first resonance circuit and the recovery time of the first rectifier element are expressed by the above conditional expression. Since (1) is satisfied, the rising of the reverse voltage applied to the first rectifying element can be made slower than before, and the surge voltage can be more effectively suppressed without depending on the device configuration. It becomes possible to do.

  Hereinafter, the best mode for carrying out the present invention (hereinafter simply referred to as an embodiment) will be described in detail with reference to the drawings.

  FIG. 1 shows a configuration of a switching power supply apparatus according to an embodiment of the present invention. This switching power supply device is a DC-DC converter that converts a high-voltage DC input voltage Vin supplied from the high-voltage battery 10 into a lower DC output voltage Vout and supplies it to a low-voltage battery (not shown) to drive the load 7. It functions.

  This switching power supply device includes an input smoothing capacitor 11, a bridge circuit 1 and a surge voltage suppression circuit 2 provided between a primary high voltage line L1H and a primary low voltage line L1L, a resonance inductor Lr, And a transformer 3 having a secondary winding 31 and secondary windings 32A and 32B. A DC input voltage Vin output from the high voltage battery 10 is applied between the input terminal T1 of the primary high voltage line L1H and the input terminal T2 of the primary low voltage line L1L. The switching power supply device also includes a rectifier circuit 4 provided on the secondary side of the transformer 3, a smoothing circuit 5 connected to the rectifier circuit 4, and a drive circuit 6 that drives the bridge circuit 1.

  The input smoothing capacitor 11 is for smoothing the DC input voltage Vin input from the input terminals T1 and T2.

  The bridge circuit 1 includes four switching elements S1 to S4, capacitors C1 to C4 and diodes D1 to D4 connected in parallel to the switching elements S1 to S4, respectively, and has a full bridge circuit configuration. It has become. Specifically, one ends of the switching elements S1 and S2 are connected to each other, and one ends of the switching elements S3 and S4 are connected to each other. Further, the other ends of the switching elements S1 and S3 are connected to each other and the other ends of the switching elements S2 and S4 are connected to each other, and these other ends are connected to the input terminals T1 and T2, respectively. With this configuration, the bridge circuit 1 converts the DC input voltage Vin applied between the input terminals T1 and T2 into an input AC voltage in accordance with the drive signals SG1 to SG4 supplied from the drive circuit 6. ing.

  The switching elements S1 to S4 are composed of switching elements such as a field effect transistor (MOS-FET) or an insulated gate bipolar transistor (IGBT). Further, when MOS-FETs are used as these switching elements, the capacitors C1 to C4 and the diodes D1 to D4 can each be constituted by a parasitic capacitance or a parasitic diode of the MOS-FET. Further, the capacitors C1 to C4 can be configured by junction capacitances of the diodes D1 to D4, respectively. When configured in this manner, it is not necessary to provide the capacitors C1 to C4 and the diodes D1 to D4 separately from the switch elements, and the circuit configuration can be simplified.

The surge voltage suppression circuit 2 has a pair of diodes D5 and D6 connected in the reverse direction and capacitors C5 and C6 connected in parallel to the diodes D5 and D6, respectively. The anode of the diode D5 is connected to the connection point P3, and the cathode is connected to the primary side high voltage line L1H. The anode of the diode D6 is connected to the primary side low-voltage line L1L, and the cathode is connected to the connection point P3. With such a configuration, the surge voltage suppression circuit 2 forms an LC series resonance circuit (first resonance circuit) between the capacitors C5 and C6 and an inductor Lr, which will be described later, and uses the resonance characteristics of the LC series resonance circuit. Thus, a surge voltage applied to rectifier diodes 4A and 4B in the rectifier circuit 4 to be described later is suppressed. Specifically, in the switching power supply of the present embodiment, the resonance time of the first resonance circuit and the recovery time of the rectifier diodes 4A and 4B are set so as to satisfy the following conditional expression (1), and the rectifier diode The reverse voltage of 4A and 4B reaches a voltage according to the turn ratio of the input voltage gently according to resonance at 1/4 of the resonance time, and during this period, recovery ends gently. The surge voltage applied to the diodes 4A and 4B is suppressed.
1/4 × {2π × (L × C) ^1/2 }> Trr1 (1)

However, {2π × (L × C) ^1/2 } is a resonance time for one period in the first resonance circuit, L is an inductance of the inductor Lr, and C is a parallel combined capacitance value of capacitors C5 and C6 ( C = (C5 + C6)), and Trr1 is the recovery time of the rectifier diodes 4A and 4B. Here, this recovery time means the time described below. That is, when these rectifier diodes 4A and 4B are PN junction diodes, the diodes are in conduction by holes injected from the P layer to the N layer, but the forward current is reduced and a reverse voltage is applied. In the process, the holes accumulated in the N layer return to the P layer or disappear by recombination, and as a result, a current flows in the reverse direction in the rectifier diodes 4A and 4B until the depletion layer spreads. This is called the recovery current, and the time during which the recovery current flows is called the recovery time. Note that when these rectifier diodes 4A and 4B are metal-semiconductor junction Schottky barrier diodes, no recovery current is generated in principle. However, even in this case, since the junction capacitance exists, in the process in which the reverse voltage is applied, a current flows in the reverse direction while the junction capacitance is charged. Therefore, in the case of a Schottky barrier diode, the time during which the reverse current flows in this way can be considered as corresponding to the recovery time.

  The inductor Lr has one end connected to the connection point P1 and the other end connected to the connection point P3. That is, the inductor Lr is H-bridge connected to a bridge circuit composed of switching elements S1, S2, diodes D5, D6, and capacitors C5, C6. With such a configuration, the inductor Lr forms an LC series resonance circuit (second resonance circuit) together with the capacitors C1 to C4 in the bridge circuit 1, and uses the resonance characteristics of the LC series resonance circuit to be described later. In addition, short circuit loss in the switching elements S1 to S4 is suppressed. Further, as described above, the LC series resonance circuit (first resonance circuit) is configured together with the capacitors C5 and C6 in the surge voltage suppression circuit 2, and the surge voltage applied to the rectification diodes 4A and 4B in the rectification circuit 4 is suppressed. It is like that. Note that the inductance of the inductor Lr is set to be very small compared to the inductance of the primary winding 31 of the transformer 3 described later.

  The transformer 3 includes a primary winding 31 and a pair of secondary windings 32A and 32B. Of these, the primary winding 31 has one end connected to the connection point P3 and the other end connected to the connection point P2. That is, the primary winding 31 is H-bridge connected to a bridge circuit composed of switching elements S3 and S4, diodes D5 and D6, and capacitors C5 and C6. On the other hand, one ends of the secondary windings 32A and 32B are connected to each other by a center tap CT, and the center tap CT is led to the output terminal T3 through the smoothing circuit 5 on the output line LO. That is, the rectifier circuit 4 described later is a center tap type. With such a configuration, the transformer 3 steps down the input AC voltage generated by the bridge circuit 1 and outputs an output AC voltage having a phase difference of 180 degrees from each end of the secondary windings 32A and 32B. It has become. In this case, the degree of step-down is determined by the turn ratio between the primary side winding 31 and the secondary side windings 32A and 32B.

  The rectifier circuit 4 is a single-phase full-wave rectifier type composed of a pair of rectifier diodes 4A and 4B. The cathode of the rectifier diode 4A is connected to the other end of the secondary winding 32A of the transformer 3, and the cathode of the rectifier diode 4B is connected to the other end of the secondary winding 32B of the transformer 3. The anodes of these rectifier diodes 4A and 4B are connected to each other and connected to the ground line LG. That is, the rectifier circuit 4 has a center tap type anode common connection configuration, and each half wave period of the output AC voltage from the transformer 3 is individually rectified by the rectifier diodes 4A and 4B to generate a DC voltage. To get.

  The rectifier diodes 4A and 4B may each be composed of a MOS-FET parasitic diode. Further, when the rectifier diodes 4A and 4B are each composed of a MOS-FET parasitic diode, the MOS-FET itself is turned on in synchronism with the period in which the MOS-FET parasitic diode is conducted. It is preferable to do. This is because rectification can be performed with a smaller voltage drop.

  The smoothing circuit 5 includes a choke coil 51 and an output smoothing capacitor 52. The choke coil 51 is inserted and arranged in the output line LO, one end of which is connected to the center tap CT, and the other end is connected to the output terminal T3 of the output line LO. The smoothing capacitor 52 is connected between the output line LO (specifically, the other end of the choke coil 51) and the ground line LG. An output terminal T4 is provided at the end of the ground line LG. With such a configuration, the smoothing circuit 5 generates the DC output voltage Vout by smoothing the DC voltage rectified by the rectifier circuit 4, and supplies this to the low voltage battery (not shown) from the output terminals T3 and T4. It has become.

  The drive circuit 6 is for driving the switching elements S <b> 1 to S <b> 4 in the bridge circuit 1. Specifically, drive signals SG1 to SG4 are supplied to the switching elements S1 to S4, respectively, and the switching elements S1 to S4 are controlled to be turned on / off. In addition, the drive circuit 6 performs switching phase control on the switching elements S1 to S4 as described later, and stabilizes the DC output voltage Vout by appropriately setting the switching phase difference. Yes.

  Here, the capacitors C1 to C4 correspond to a specific example of “second capacitance element” in the present invention, the inductor Lr corresponds to a specific example of “resonance inductor” in the present invention, and the rectifier diodes 4A and 4B This corresponds to a specific example of “first rectifier element” in the present invention. Capacitors C5 and C6 correspond to a specific example of “first capacitor element” in the present invention, and diodes D5 and D6 correspond to a specific example of “second rectifier element” in the present invention. Capacitor C5 and diode D5, capacitor C6, and diode D6 each correspond to a specific example of “element pair” in the present invention. The switching elements S3 and S4 correspond to a specific example of “one two switching elements” in the present invention, and the switching elements S1 and S2 correspond to a specific example of “the other two switching elements” in the present invention. To do.

  Next, the operation of the switching power supply device configured as described above will be described. First, the basic operation of the switching power supply device will be described.

  The bridge circuit 1 switches the DC input voltage Vin supplied from the high voltage battery 10 via the input terminals T <b> 1 and T <b> 2 to generate an input AC voltage, and supplies this to the primary winding 31 of the transformer 3. From the secondary windings 32 </ b> A and 32 </ b> B of the transformer 3, an output AC voltage that has been transformed (here, stepped down) is taken out.

  The rectifier circuit 4 rectifies the output AC voltage by the rectifier diodes 4A and 4B. As a result, a rectified output is generated between the center tap CT (output line LO) and the connection point (ground line LG) between the rectifier diodes 4A and 4B.

  The smoothing circuit 5 smoothes the rectified output generated between the center tap CT and the rectifier diodes 4A and 4B, and outputs the DC output voltage Vout from the output terminals T3 and T4. The DC output voltage Vout is supplied to a low voltage battery (not shown) and the load 7 is driven.

  Next, with reference to FIG. 2 to FIG. 15, the operation for suppressing the surge voltage applied to the rectifier diodes 4A and 4B in the rectifier circuit 4, which is the main feature of the present invention, will be described in detail.

Here, FIG. 2 is a timing waveform diagram (timing t0 to t10) showing the voltage waveform or current waveform of each part in the switching power supply device of FIG. 1, and (A) to (D) in the drawing are driving. The voltage waveforms of the signals SG1 to SG4, (E) to (G) are the potentials VP1 to VP3 of the connection points P1 to P3, and (H) is the connection between the connection points P1 and P3 with reference to the potential VP3 of the connection point P3. The potential difference V _{P1 -P3} , (I) the potential difference V _P3 -P2 between the connection points P3 and P2 with reference to the potential VP2 of the connection point _P2 , (J) the current Ir flowing through the inductor Lr, (K) Is a current I31 flowing through the primary winding 31 of the transformer 3, and (L) and (M) are currents I5 flowing through parallel connection portions of the diodes D5 and D6 and the capacitors C5 and C6 in the surge voltage suppression circuit 2, respectively. I6, (N), (P) Reverse voltages V4A and V4B applied between the anodes and cathodes of the rectifier diodes 4A and 4B, (O) and (Q) indicate currents I4A and I4B flowing through the rectifier diodes 4A and 4B, respectively, and (R) indicates a choke coil 51. Each flowing current I51 is shown. The direction of each voltage is as shown by the arrow in FIG. 1, and the direction from “−” to “+” is the positive direction. In addition, the direction of each current is the positive direction as indicated by the arrow in FIG.

  3 to 14 show the operation state of the switching power supply device at each timing (timing t0 to t10) in FIG. 2, and FIG. 15 shows the timing after the timing shown in FIG. 2 (timing t10 to t20). (T0)) represents the voltage waveform or current waveform of each part. The timing shown in FIG. 2 and FIG. 15 respectively represents the half cycle of the operation of the switching power supply device, and the operation for one cycle is combined with these operations.

  First, the operation for the first half cycle will be described with reference to FIGS.

  Looking at the drive signals SG1 to SG4 (FIGS. 2A to 2D) of the switching elements S1 to S4, it can be seen that these switching elements are divided into two switching element pairs. Specifically, the switching elements S1 and S2 are both controlled to be turned on at a fixed timing on the time axis, and are referred to as “fixed-side switching elements”. The switching elements S3 and S4 are both controlled to be turned on at variable timing on the time axis and are referred to as “shift-side switching elements”.

  Further, these switching elements S1 to S4 are driven with a combination and timing at which the input terminals T1 and T2 to which the DC input voltage Vin is applied are not electrically short-circuited in any state of the switching operation. Specifically, the switching elements S3 and 4 (fixed side switching elements) are not turned on at the same time, and the switching elements S1 and S2 (shift side switching elements) are not turned on at the same time. The time interval that is taken in order to avoid these being turned on at the same time is referred to as dead time Td (FIGS. 2A and 2D).

  Further, the switching elements S1 and S4 have a period in which they are simultaneously turned on, and the primary side winding 31 of the transformer 3 is excited during the period in which the switching elements S1 and S4 are simultaneously turned on. The switching elements S1 and S4 operate so as to have a switching phase difference φ with reference to the switching element S1 (fixed-side switching element) (FIGS. 2A and 2D). Similarly, the switching elements S2 and S3 have a period during which the switching elements S2 and S3 are simultaneously turned on. During the period during which the switching elements S2 and S3 are simultaneously turned on, the primary winding 31 of the transformer 3 is excited in the opposite direction. These switching elements S2 and S3 operate so as to have a switching phase difference φ with reference to the switching element S2 (fixed-side switching element) (FIGS. 2B and 2C). Further, when switching phase difference φ between switching element S1 and switching element S4 and switching phase difference φ between switching element S2 and switching element S3 are controlled, switching element S1 and switching element S4 are simultaneously turned on. And the time during which the switching element S2 and the switching element S3 are simultaneously turned on respectively change. As a result, the duty ratio of the input AC voltage applied to the primary winding 31 of the transformer 3 changes, and the DC output voltage Vout is stabilized.

First, in the period from timing t0 to t1 shown in FIG. 3, the switching elements S1 and S4 are in the on state (FIGS. 2A and 2D), and the switching elements S2 and S3 are in the off state. (FIGS. 2B and 2C). Further, the potential VP1 = Vin (FIG. 2E) at the connection point P1 and the potential VP2 = 0V (FIG. 2F) at the connection point P2, and the inductance of the inductor Lr is 1 of the transformer 3 as described above. Since it is very small as compared with the inductance of the secondary winding 31, the potential VP3 of the connection point P3≈Vin (FIG. 2 (G)), and the potential difference V _P3-P2 between the connection points P3 and P2 with reference to VP2. Is substantially equal to Vin (FIG. 2 (I)). Therefore, the loop current Ia as shown in FIG. 3 flows through the bridge circuit 1 to excite the inductor Lr, and power is transmitted from the primary side to the secondary side of the transformer 3. Therefore, on the secondary side of the transformer 3, the loop current Ixa flows through the rectifier diode 4A and the choke coil 51, and the load 7 is driven. In this period, a forward voltage is applied to the rectifier diode 4A and the reverse voltage V4A = 0V (FIG. 2 (N)), while the reverse voltage V4B is applied to the rectifier diode 41B (FIG. 2). 2 (P)).

  Next, in the period from the timing t1 to the timing t2 shown in FIG. 4, the switching element S4 is turned off at the timing t1 (FIG. 2D). Then, the capacitors C3 and C4 and the inductor Lr cooperate to form an LC series resonance circuit (second resonance circuit), and the second resonance operation is performed. Therefore, the loop currents Ib and Ic as shown in FIG. 4 flow and the capacitor C3 is discharged, while the capacitor C4 is charged, so that the potential VP2 at the connection point P2 gradually rises at the timing t2. VP2 = Vin (FIG. 2F). At this time, the reverse voltage V4B of the rectifier diode 4B gradually decreases and becomes 0 V at timing t2 ((P) in FIG. 2).

  Here, as shown in FIG. 5, when VP2 = Vin at timing t2 (FIG. 2F), the diode D3 becomes conductive. Further, after VP2 = Vin and the diode D3 is turned on, as shown in FIG. 6, the switching element S3 is turned on at the timing t3 (FIG. 2C), so that zero volt switching ( ZVS (Zero Volt Switching) operation is performed, and as a result, the short-circuit loss in the switching element S3 is suppressed.

  In the period from the timing t2 to t4, the energy stored in the inductor Lr by being excited in the period from the timing t0 to t1 tends to circulate as a current in the circuit connected to both ends of the inductor Lr. . Specifically, as shown in FIG. 6, the potential difference between one end (connection point P3) of the inductor Lr and the other end (primary high voltage line L1H side) of the switching element S1 is equal to each other. Loop currents Id and Ie flow, respectively. Here, in the path of the loop current Id, this potential difference is the sum of the voltage V31 across the primary winding 31 of the transformer 3 and the voltage VS3 across the switching element S3. V31 is obtained by dividing the forward voltage drop of the rectifier diode 4A by the turn ratio n, where n is the turn ratio of the primary side winding and the secondary side winding of the transformer 3, and V31 is a switching element. When S3 is in the off state (period t2 to t3), the forward voltage drop of the diode D3 occurs, and when the switching element S3 is in the on state (period t3 to t4), it flows with the on-resistance of the switching element S3. It is the product of the current. On the other hand, in the loop current Ie path, the potential difference is a forward voltage drop of the diode D5.

  Here, the forward voltage drop values of the diodes 4A, D3, and D5 vary depending on the flowing forward current value and the ambient temperature, but the loop currents Id and Ie have the same potential difference. Flowing into. In addition, by dividing the current into the two loop currents Id and Ie in this way, the absolute value of the current I31 flowing through the primary winding 31 of the transformer 3 decreases (FIG. 2 (K)). Furthermore, the current I51 is equalized so that the ampere turns in the transformer 3 are equal and the sum of the currents flowing through the secondary windings 32A and 32B of the transformer 3 is equal to the current I51 flowing through the choke coil 51. The current is divided into a loop current Ixa flowing through the rectifier diode 4A and a loop current Ixb flowing through the rectifier diode 4B.

  Next, as shown in FIG. 7, at the timing t4, the switching element S1 is turned off (FIG. 2A). Then, the capacitors C1 and C2 and the inductor Lr cooperate to form an LC series resonance circuit (second resonance circuit), and the second resonance operation is performed. Therefore, loop currents If, Ig, Ih and Ii as shown in FIG. 7 flow. Accordingly, the capacitor C2 is discharged, while the capacitor C1 is charged. Therefore, the potential VP1 of the connection point P1 gradually decreases, and VP1 = 0V is reached at the timing t5 (FIG. 2E).

Here, as shown in FIG. 8, when VP1 = 0 V at timing t5 (FIG. 2E), VP3 = Vin (FIG. 2G) and V _P1-P3 = −Vin (FIG. 2). (H)), the diode D2 becomes conductive. In addition, after VP1 = 0V and the diode D2 is turned on in this way, as shown in FIG. 9, the switching element S2 is turned on at the timing t6 (FIG. 2B), and the ZVS operation is performed. As a result, the short circuit loss in the switching element S2 is suppressed.

  Next, during the period from the timing t6 to the timing t7 shown in FIG. 9, the energy stored in the inductor Lr is the loop current Im, as shown in FIG. 9, even after the charging and discharging of the capacitors C1 and C2 are completed. Il is regenerated to the input smoothing capacitor 11 by Il. As the input smoothing capacitor 11 is regenerated, the energy stored in the inductor Lr decreases, and the absolute value of the current Ir flowing through the inductor Lr and the current I31 flowing through the primary winding 31 of the transformer 3 are reduced accordingly. The absolute value also decreases (FIGS. 2 (J) and (K)). For this reason, the ampere-turns in the transformer 3 become equal, and the current I51 is such that the sum of the currents flowing through the secondary windings 32A and 32B of the transformer 3 is equal to the current I51 flowing through the choke coil 51. The current is divided into a loop current Ixa flowing through the rectifier diode 4A and a loop current Ixb flowing through the rectifier diode 4B.

  Also, during this period, the loop currents Im and Il flow so that the potential difference between one end of the inductor Lr (connection point P3) and the cathode of the diode D5 is equal to each other. The potential difference in the path becomes larger than the potential difference in the path of the loop current Il, and the diode D5 becomes non-conductive, so that the absolute value of the current Ir flowing through the inductor Lr and the primary winding 31 of the transformer 3 flow. The absolute value of the current I31 becomes equal (FIGS. 2 (J) and (K)). As described above, the potential difference in the path of the loop current Il is caused by the voltage V31 between the both ends of the primary side winding 31 of the transformer 3 (the forward voltage drop of the rectifier diode 4A is the primary side winding of the transformer 3). And the voltage VS3 across the switching element S3 (since the switching element S3 is on during this period, the on-resistance of the switching element S3) The potential difference in the path of the loop current Im becomes a forward voltage drop of the diode D5.

  Next, as shown in FIG. 10, at time t7, all the energy stored in the inductor Lr is regenerated, and the current Ir flowing through the inductor Lr = the current I31 = 0A flowing through the primary winding 31 of the transformer 3 (FIGS. 2 (J) and (K)) and the current I4A flowing through the rectifier diode 4A = the current I4B flowing through the rectifier diode 4B (FIGS. 2 (O) and (Q)). From this timing t7, the inductor Lr stores energy in the opposite direction as before, and the inductor Lr and the primary winding 31 of the transformer 3 have the opposite direction as shown in FIG. As the loop current In flows, the current Ir increases at a rate of Vin / L (L: inductance of the inductor Lr) (FIGS. 2 (J) and (K)). For this reason, the ampere-turns in the transformer 3 become equal, and the current I51 is such that the sum of the currents flowing through the secondary windings 32A and 32B of the transformer 3 is equal to the current I51 flowing through the choke coil 51. The current is divided into a loop current Ixa flowing through the rectifier diode 4A and a loop current Ixb flowing through the rectifier diode 4B. However, the current I4A flowing through the rectifier diode 4A gradually decreases, while the current I4B flowing through the rectifier diode 4B gradually increases (FIGS. 2 (O) and (Q)). When I4A = 0A, and the current flowing through the secondary winding 32B of the transformer 3 becomes equal to the current I51 flowing through the choke coil 51, the ampere turn in the transformer 3 does not increase any more, so I31 increases. However, the capacitors C5 and C6 of the surge voltage suppression circuit 2 and the inductor Lr cooperate to form an LC series resonance circuit (first resonance circuit), and the first resonance operation is started. This time corresponds to the timing t8.

Next, during the period from timing t8 to timing t9 shown in FIG. 12, loop currents Io and Ip flow by the first resonance operation. Accordingly, the capacitor C6 is discharged, while the capacitor C5 is charged, so that the potential VP3 at the connection point P3 gradually decreases along with the first resonance operation (FIG. 2 (G)). Accordingly, the absolute value of the voltage V31 between both ends of the primary side winding 31 of the transformer 3 increases, and voltages V32A and V32B are also generated in the secondary side windings 32A and 32B, respectively, and V32A = V32B = V31 / n (n: turn ratio of the primary side winding and the secondary side winding of the transformer 3), (potential of the cathode of the rectifier diode 4B) <(potential of the center tap CT) <(cathode of the rectifier diode 4A) ), (Current Ir flowing through the inductor Lr) = (current I31 flowing through the primary winding 31 of the transformer 3) + (current I5 flowing through the parallel connection portion of the diode D5 and the capacitor C5) + (diode D6 and The current I6) flows through the parallel connection portion with the capacitor C6. As described above, when VP3 gradually falls and VP3 = 0V and VP3 _-P2 = −Vin (FIGS. 2 (G) and (I)), it corresponds to timing t9.

  Here, in the switching power supply according to the present embodiment, the resonance time of the first resonance circuit and the recovery time of the rectifier diodes 4A and 4B satisfy the above-described conditional expression (1) during the period from the timing t8 to t9. Therefore, the generation of the recovery current in the rectifier diodes 4A and 4B is suppressed. Accordingly, the first resonance operation by the capacitors C5 and C6 and the inductor Lr is to be continued, but since VP3 = 0V (FIG. 2G), the voltage across the capacitor C6 and the diode D6 becomes 0V. , The current IC6 flowing through the capacitor C6 becomes 0A, and the diode D6 becomes conductive.

Therefore, in the period from timing t9 to t10 shown in FIG. 13, since the diode D6 is conductive and the switching element S3 is in the ON state (FIG. 2C), the primary side winding of the transformer 3 The voltage V31 at both ends of 31 (and the absolute value of VP3 _-P2 (FIG. 2 (I))) is clamped to Vin, so that the voltage V32B at both ends of the secondary winding 32B of the transformer becomes Vin / n ( n: The winding ratio of the primary side winding and the secondary side winding of the transformer 3). For this reason, the reverse voltage V4A applied to the rectifier diode 4A does not become larger than 2 × Vin / n because the rectifier circuit 4 has a center tap type configuration (FIG. 2 (N)). In other words, the reverse voltage V4A applied to the rectifier diode 4A is 2 × Vin / n or less at the maximum, and the surge voltage is suppressed from increasing.

  Further, during the period from the timing t9 to t10, the diode D6 conducts as described above, so (current Ir flowing through the inductor Lr) = (current I31 flowing through the primary winding 31 of the transformer 3) + ( The current ID6) flows through the diode D6, and the resonance current due to the first resonance operation is represented by the loop current Iq as shown in FIG. 13, while Ir is constant (FIG. 2 (J)). Further, as the choke coil 51 is excited by the voltage V32B across the secondary winding 32B of the transformer 3, the current I51 flowing through the choke coil 51 increases and I31 = (the secondary winding 32A is Since current I32A) + (current I32B flowing through secondary winding 32B) = I32B = I51, I31 also increases (FIG. 2 (K)). Furthermore, since Ir = I31 + ID6 and Ir are constant, ID6 decreases as I31 increases. When ID6 = I6 = 0V (FIG. 2 (M)), it corresponds to the timing t10 shown in FIG. Thus, the operation for the first half cycle is completed.

  Next, with reference to FIG. 15, an operation for a half cycle (timing t <b> 10 to t <b> 20 (t <b> 0)) after timing t <b> 0 to t <b> 10 illustrated in FIG. 2 will be described.

The operation for this half cycle is basically the same as the operation for the half cycle described with reference to FIGS. That is, in the period from timing t10 to t11, the switching elements S2 and S3 are in the on state (FIGS. 15B and 15C), and the switching elements S1 and S4 are in the off state (FIG. 15). (A), (D)). Further, the potential VP1 = 0V (FIG. 15E) at the connection point P1 and the potential VP2 = Vin (FIG. 15F) at the connection point P2, and the inductance of the inductor Lr is the primary side winding of the transformer 3. Therefore, the potential VP3 of the connection point P3 is almost 0V (FIG. 15G), and the potential difference V _P3−P2 between the connection points P3 and P2 with respect to VP2 is also almost 0V. They are equal (FIG. 15I). Therefore, a loop current flows through the bridge circuit 1 to excite the inductor Lr, and power is transmitted from the primary side to the secondary side of the transformer 3. Therefore, a loop current flows through the rectifier diode 4B and the choke coil 51 on the secondary side of the transformer 3, and the load 7 is driven. In this period, a forward voltage is applied to the rectifier diode 4B and the reverse voltage V4B = 0V (FIG. 15 (P)), while the reverse voltage V4A is applied to the rectifier diode 41A (FIG. 15). 15 (N)).

  Next, in a period from timing t11 to t12, the switching element S3 is turned off at timing t11 (FIG. 15C). Then, the capacitors C3 and C4 and the inductor Lr cooperate to form an LC series resonance circuit (second resonance circuit), and the second resonance operation is performed. Therefore, the capacitor C3 is charged by the two loop currents, while the capacitor C4 is discharged. Therefore, the potential VP2 at the connection point P2 gradually decreases, and VP2 = 0V is reached at timing t12 (FIG. 15 ( F)). At this time, the reverse voltage V4A of the rectifier diode 4A gradually decreases and becomes 0 V at timing t12 (FIG. 15 (N)).

  Here, when VP2 = 0 V at timing t12 (FIG. 15F), the diode D4 becomes conductive. Further, after VP2 = 0V and the diode D4 is turned on, the switching element S4 is turned on at timing t13 (FIG. 15D), so that the ZVS operation is performed. As a result, the switching element S4 Short circuit loss at is suppressed.

  In the period from the timing t12 to t14, as described above, the energy stored in the inductor Lr by being excited in the period from the timing t10 to t11 is circulated as a current in the circuit connected to both ends of the inductor Lr. Since the current is divided into two loop currents, the absolute value of the current I31 flowing through the primary winding 31 of the transformer 3 decreases (FIG. 15K). In addition, the current I51 is equal to the current I51 flowing through the choke coil 51 so that the ampere turns in the transformer 3 are equal and the sum of the currents flowing through the secondary windings 32A and 32B of the transformer 3 is equal. The current is divided into a loop current Ixa flowing through the rectifier diode 4A and a loop current Ixb flowing through the rectifier diode 4B.

  Next, at timing t14, the switching element S2 is turned off (FIG. 15B). Then, the capacitors C1 and C2 and the inductor Lr cooperate to form an LC series resonance circuit (second resonance circuit), and the second resonance operation is performed. Accordingly, four loop currents flow and the capacitor C2 is charged, while the capacitor C1 is discharged. Therefore, the potential VP1 at the connection point P1 gradually rises, and VP1 = Vin at timing t15 (FIG. 15). (E)).

Here, when VP1 = Vin at timing t15 (FIG. 15E), VP3 = 0V (FIG. 15G) and V _P1-P3 = Vin (FIG. 15H) at this time, The diode D1 becomes conductive. Further, after VP1 = Vin and the diode D1 is turned on, the switching element S1 is turned on at the timing t16 (FIG. 15A), so that the ZVS operation is performed. As a result, in the switching element S1 Short circuit loss is suppressed.

  Next, in the period from timing t16 to t17, the energy stored in the inductor Lr is regenerated to the input smoothing capacitor 11 by two loop currents even after the charging and discharging of the capacitors C1 and C2 are completed. As the input smoothing capacitor 11 is regenerated, the energy stored in the inductor Lr decreases, and the absolute value of the current Ir flowing through the inductor Lr and the current I31 flowing through the primary winding 31 of the transformer 3 are reduced accordingly. The absolute value also decreases (FIGS. 15 (J) and (K)). For this reason, the ampere-turns in the transformer 3 become equal, and the current I51 is such that the sum of the currents flowing through the secondary windings 32A and 32B of the transformer 3 is equal to the current I51 flowing through the choke coil 51. The current is divided into a loop current Ixa flowing through the rectifier diode 4A and a loop current Ixb flowing through the rectifier diode 4B. Further, during this period, the diode D6 is non-conductive, so that the absolute value of the current Ir flowing through the inductor Lr is equal to the absolute value of the current I31 flowing through the primary side winding 31 of the transformer 3 (FIG. 15 ( J), (K)).

  Next, at timing t17, all the energy stored in the inductor Lr is regenerated, and the current Ir flowing through the inductor Lr = the current I31 = 0A flowing through the primary winding 31 of the transformer 3 (FIG. 15 (J), ( K)), and current I4A flowing through rectifier diode 4A = current I4B flowing through rectifier diode 4B (FIGS. 15 (O) and (Q)). Then, from this timing t17, the inductor Lr stores energy in the reverse direction, and a loop current in the opposite direction flows through the inductor Lr and the primary winding 31 of the transformer 3. At the same time, the current Ir increases at a rate of Vin / L (L; the inductance of the inductor Lr) (FIGS. 15J and 15K). For this reason, the ampere-turns in the transformer 3 become equal, and the current I51 is such that the sum of the currents flowing through the secondary windings 32A and 32B of the transformer 3 is equal to the current I51 flowing through the choke coil 51. The current is divided into a loop current Ixa flowing through the rectifier diode 4A and a loop current Ixb flowing through the rectifier diode 4B. However, the current I4B flowing through the rectifier diode 4B gradually decreases, while the current I4A flowing through the rectifier diode 4A gradually increases (FIGS. 15 (O) and (Q)). When I4B = 0A and the current flowing through the secondary side winding 32A of the transformer 3 becomes equal to the current I51 flowing through the choke coil 51, the ampere turn in the transformer 3 does not increase any more, so I31 increases. However, the capacitors C5 and C6 of the surge voltage suppression circuit 2 and the inductor Lr cooperate to form an LC series resonance circuit (first resonance circuit), and the first resonance operation is started. This time corresponds to the timing t18.

Next, in the period from timing t18 to t19, two loop currents flow through the first resonance operation and the capacitor C6 is charged, while the capacitor C5 is discharged. The potential VP3 at the connection point P3 gradually rises (FIG. 15G). Along with this, the voltage V31 between both ends of the primary side winding 31 of the transformer 3 increases, and voltages V32A and V32B are also generated in the secondary side windings 32A and 32B, respectively. Thus, when VP3 gradually rises and VP3 = Vin and VP3 _-P2 = Vin (FIGS. 15 (G) and (I)), it corresponds to timing t19.

  In the switching power supply of the present embodiment, the resonance time of the first resonance circuit and the recovery time of the rectifier diodes 4A and 4B satisfy the above-described conditional expression (1) in the period from the timing t18 to t19. Therefore, the generation of the recovery current in these rectifier diodes 4A and 4B is suppressed. Therefore, the first resonance operation by the capacitors C5 and C6 and the inductor Lr is to be continued, but since VP3 = Vin (FIG. 15G), the voltage across the capacitor C5 and the diode D5 becomes 0V. , The current IC5 flowing through the capacitor C5 becomes 0A, and the diode D5 becomes conductive.

Therefore, in the period from timing t19 to t20, the diode D5 is conductive and the switching element S4 is in the ON state (FIG. 15D), so the voltage across the primary winding 31 of the transformer 3 V31 (and the absolute value of VP3 _-P2 (FIG. 15I)) is clamped to Vin, so that the voltage V32A across the secondary winding 32A of the transformer becomes Vin / n (n; (The turn ratio between the primary winding and the secondary winding). Therefore, the reverse voltage V4B applied to the rectifier diode 4B does not become larger than 2 × Vin / n because the rectifier circuit 4 has a center tap type configuration (FIG. 15 (P)). In other words, the reverse voltage V4B applied to the rectifier diode 4B is 2 × Vin / n or less at the maximum, and the surge voltage is suppressed from increasing.

  In the period from the timing t19 to t20, the diode D5 is turned on as described above, so Ir becomes constant (FIG. 15 (J)). Further, as the choke coil 51 is excited by the voltage V32A across the secondary winding 32A of the transformer 3, the current I51 flowing through the choke coil 51 increases and I31 also increases (FIG. 15 ( K)). Furthermore, since Ir = I31 + ID5 and Ir are constant, ID5 decreases as I31 increases. The time when ID5 = I5 = 0V (FIG. 15L) corresponds to the timing t20. Thus, the operation for the latter half cycle is completed, and the state becomes equivalent to the timing t0 in FIG.

  Next, referring to FIG. 16 to FIG. 18, the waveform of the surge voltage applied to the rectifier diode in the switching power supply of the present embodiment and the surge applied to the rectifier diode in the conventional switching power supply (Comparative Examples 1 and 2). The voltage waveform will be described while being compared.

  Here, FIGS. 16A to 16C show timing waveforms of the reverse voltage applied to the rectifier diode in the switching power supply according to the present embodiment and Comparative Examples 1 and 2, respectively. FIGS. 17 and 18 show the configurations of the switching power supply devices according to Comparative Examples 1 and 2, respectively. Specifically, in Comparative Example 1, a surge voltage suppression circuit 102 in which capacitors C5 and C6 are removed from the surge voltage suppression circuit 2 is provided instead of the surge voltage suppression circuit 2 of the present embodiment. In Comparative Example 2, a surge voltage suppression snubber circuit 202 including an inductor L7, a capacitor C7, and a diode D7 is provided on the secondary side of the transformer 3 instead of the surge voltage suppression circuit 2. is there. Specifically, in the snubber circuit 202, one end of the inductor L7 is connected between the choke coil 51 on the output line LO and the center tap CT, and the other end is connected to the cathode of the diode D7 and one end of the capacitor C7. Has been. The anode of the diode D7 is also connected between the choke coil 51 on the output line LO and the center tap CT, and the other end of the capacitor C5 is connected to the ground line LG. The reverse voltage waveforms shown in FIGS. 16A to 16C are voltage waveforms at the center tap CT on the secondary side of the transformer 3, and the reverse voltage actually applied to the rectifier diodes 4A and 4B is as follows. The value is doubled.

  First, in the reverse voltage waveform according to Comparative Example 2 shown in FIG. 16C, the maximum value (peak value) of the surge voltage is 83V. This is due to the result that the surge voltage is suppressed to some extent by the snubber circuit 202, and is about the DC input voltage Vin / n (n: the turn ratio of the primary side winding and the secondary side winding of the transformer 3). This corresponds to 2 times (2.02 times). On the other hand, in the reverse voltage waveform according to Comparative Example 1 shown in FIG. 16B, the maximum value of the surge voltage is 52 V, which corresponds to 1.26 times Vin / n. Further, in the reverse voltage waveform according to the comparative example 1, the rise time up to the maximum value is about 20 ns, which is steep due to the fact that the surge voltage suppression circuit 102 does not include a capacitor. You can see that it is standing up.

  On the other hand, in the reverse voltage waveform according to the present embodiment shown in FIG. 16A, the surge voltage suppression circuit 2 includes capacitors C5 and C6, and is composed of these capacitors C5 and C6 and an inductor Lr. Since the resonance time of the first resonance circuit and the recovery time of the rectifier diodes 4A and 4B are set so as to satisfy the conditional expression (1), the rectifier diodes 4A and 4B It can be seen that the generation of the recovery current is suppressed and the current rises gently due to the resonance operation of the first resonance circuit. Specifically, the maximum value of the surge voltage is 45.5 V, which corresponds to about 1 time (1.08 times) of Vin / n, and the rise time to this maximum value is about 100 ns. It has become. That is, it can be seen that the rising of the reverse voltage is moderate as compared with Comparative Examples 1 and 2 shown in FIGS. 16B and 16C, and as a result, the surge voltage is more effectively suppressed. .

  As described above, in the present embodiment, the capacitors C5 and C6 in the surge voltage suppression circuit 2 and the inductor Lr constitute the first resonance circuit, and the resonance time of the first resonance circuit and the rectifier circuit 4 Since the recovery time of the rectifier diodes 4A and 4B satisfies the above-mentioned conditional expression (1), the rising of the reverse voltage applied to the rectifier diodes 4A and 4B can be made slower than the conventional case, and the device configuration can be reduced. Without depending on it, it becomes possible to more effectively suppress the surge voltage. Specifically, for example, when the rectifier circuit 4 has a center tap type configuration as in the present embodiment, the maximum value (peak value) of the surge voltage is 2 × Vin / n (n; The ratio of the number of turns of the primary winding and the secondary winding) can be suppressed, and the maximum value can be reduced as compared with the conventional case where the maximum value is about 4 × Vin / n.

  In addition, since the surge voltage can be suppressed, the loss in the rectifying element can be reduced and the efficiency of the apparatus can be improved. Further, by reducing the loss in the rectifying element, it is possible to suppress heat generation in the element.

  Further, by suppressing the surge voltage from rising, a rectifying element (rectifying diode) having a low withstand voltage can be used, and the component cost can be reduced.

  Furthermore, since the surge voltage can be suppressed without depending on the device configuration, the degree of freedom in device design can be improved.

  While the present invention has been described with reference to the embodiment, the present invention is not limited to this embodiment, and various modifications can be made.

For example, in addition to the conditional expression (1) described in the above embodiment, the resonance time of the first resonance circuit and the recovery times Trr2 of the diodes D5 and D6 in the surge voltage suppression circuit 2 are as follows: It is preferable to set so as to satisfy (2). In such a configuration, in addition to the rectifying diodes 4A and 4B described above, the reverse voltage applied to the diodes D5 and D6 gradually reaches the input voltage according to resonance at 1/4 of the resonance time, and recovers gently during that period. Therefore, the surge voltage rise in the diodes D5 and D6 is also suppressed. Therefore, it is possible to suppress the occurrence of ringing in the reverse voltage applied to the diodes D5 and D6, thereby suppressing the generation of noise.
1/4 × {2π × (L × C) ^1/2 }> Trr2 (2)

  Further, for example, as shown in FIG. 19, in the switching power supply device (FIG. 2) of the above embodiment, the configuration of the inductor Lr, the transformer 3 and its secondary circuit (rectifier circuit 4 and smoothing circuit 5) May be arranged so as to be reversed left and right with respect to the surge voltage suppression circuit 2. Specifically, the inductor Lr may be disposed between the connection points P2 and P3, and the transformer 3 may be disposed between the connection points P1 and P3. Even when configured in this manner, the same effects as those of the above-described embodiment can be obtained.

  For example, as shown in FIG. 20, the center tap type rectifier circuit 4 may be a full bridge type rectifier circuit 41. Specifically, a transformer 33 having a primary side winding 331 and one secondary side winding 332 is provided in place of the transformer 3 of FIG. 1, and four rectifier diodes 41A are provided on the secondary side of the transformer 33. A full bridge type rectifier circuit 41 including ˜41D is provided. When configured in this manner, the maximum value (peak value) of the surge voltage applied to the rectifier diodes 41A to 41D is set to 1 × Vin / n (n: the primary side winding of the transformer 3 by the same operation as the above embodiment. The ratio of the number of turns of the secondary winding and the secondary winding can be reduced, and can be reduced as compared with a conventional full-bridge type whose maximum value is about 2 × Vin / n. The rectifier diodes 41A to 41D can also be composed of MOS-FET parasitic diodes as in the case of the rectifier diodes 4A and 4B.

  Further, in the above embodiment, the case where the transformer 3 and the inductor Lr are provided magnetically independent from each other has been described. However, as shown in FIGS. 21 and 22, for example, the primary side of the transformer 3 The auxiliary winding 31B of the transformer 3 is provided to the auxiliary winding 31B, and the auxiliary winding 31B and the transformer 3 are magnetically coupled to each other as indicated by symbols M1 and M2 in the figure (magnetic flux (magnetic path) ) May be shared). Specifically, the inductor Lr is disposed between the connection points P1 and P2, and the auxiliary winding 31B of the transformer 3 is connected between the connection points P1 and P3 or between the connection points P1 and P2. Even in such a configuration, the configuration shown in FIG. 21 and FIG. 22 is equivalent to the configuration shown in FIG. 1 or FIG. 19, so that the same effect as the above embodiment can be obtained. it can.

  Further, when the transformer 3 and its auxiliary winding 31B are magnetically coupled as described above, for example, as shown in FIGS. 23 and 24, the surge voltage suppression circuit 21 is used instead of the surge voltage suppression circuit 2. , 22 may be provided. Specifically, an element pair consisting of a diode D5 and a capacitor C5 and an element pair consisting of a diode D6 and a capacitor C6 are connected in parallel between the primary high voltage line L1H and the primary low voltage line L1L. The auxiliary windings 31B and 31C of the transformer 3 may have a center tap type configuration (magnetically coupled to the transformer 3 as indicated by reference numerals M3 and M4 in the figure). Even when configured in this manner, the same effects as those of the above-described embodiment can be obtained. 21 to 24 show an example in which the auxiliary windings 31B and 31C and the transformer 3 are magnetically coupled to each other, but the auxiliary windings 31B and 31C and the resonance inductor Lr are separately provided. It may be magnetically coupled and is equally effective.

  In the above-described embodiment, the case where the resonance inductor Lr is arranged on the primary side of the transformer 3 has been described. However, as shown in FIGS. It may be provided on the secondary side of the transformer 3. Specifically, as shown in FIGS. 25 and 27, a pair of inductors LrA and LrB magnetically coupled to each other are connected to the cathodes of the rectifier diodes 4A and 4B in the center tap type rectifier circuit 4, respectively. It may be provided between the secondary side windings 32A and 32B of the transformer 3, and the inductor Lr is rectified in the full bridge type rectifier circuit 41 as shown in FIGS. You may make it arrange | position between the connection point of the anode of the diode 41A and the cathode of the rectifier diode 41B, and the connection point of the anode of the rectifier diode 41C and the cathode of the rectifier diode 41D. Further, as indicated by reference numerals M5 and M6 in FIGS. 25 and 26, respectively, the transformer 3 and its auxiliary windings 31B and 331B may be magnetically coupled, as shown in FIGS. As indicated by reference numerals M7 and M8 in the middle, the resonance inductor Lr or the resonance inductors LrA and LrB and the auxiliary windings 31B and 331B may be magnetically coupled. Even when configured as described above, the same effects as those of the above-described embodiment can be obtained.

  Furthermore, of course, you may make it comprise combining these modifications.

It is a circuit diagram showing the structure of the switching power supply device which concerns on one embodiment of this invention. FIG. 2 is a timing waveform diagram for explaining the operation of the switching power supply device of FIG. 1. It is a circuit diagram for demonstrating operation | movement of the switching power supply device of FIG. It is a circuit diagram for demonstrating operation | movement of the switching power supply device following FIG. FIG. 5 is a circuit diagram for explaining the operation of the switching power supply device following FIG. 4. FIG. 6 is a circuit diagram for explaining the operation of the switching power supply device following FIG. 5. It is a circuit diagram for demonstrating operation | movement of the switching power supply device following FIG. It is a circuit diagram for demonstrating operation | movement of the switching power supply device following FIG. It is a circuit diagram for demonstrating operation | movement of the switching power supply device following FIG. FIG. 10 is a circuit diagram for explaining an operation of the switching power supply device following FIG. 9. It is a circuit diagram for demonstrating operation | movement of the switching power supply device following FIG. FIG. 12 is a circuit diagram for explaining the operation of the switching power supply device following FIG. 11. It is a circuit diagram for demonstrating operation | movement of the switching power supply device following FIG. It is a circuit diagram for demonstrating operation | movement of the switching power supply device following FIG. FIG. 15 is a timing waveform diagram for explaining the operation of the switching power supply device following FIG. 14. It is a timing waveform diagram for comparing the operation of the switching power supply device according to FIG. 1 and Comparative Examples 1 and 2. 6 is a circuit diagram illustrating a configuration of a switching power supply device according to a comparative example 1. FIG. 10 is a circuit diagram illustrating a configuration of a switching power supply device according to a comparative example 2. FIG. It is a circuit diagram showing the structure of the switching power supply device which concerns on the modification of this invention. It is a circuit diagram showing the structure of the rectifier circuit which concerns on the modification of this invention. It is a circuit diagram showing the structure of the switching power supply device which concerns on the other modification of this invention. It is a circuit diagram showing the structure of the switching power supply device which concerns on the other modification of this invention. It is a circuit diagram showing the structure of the switching power supply device which concerns on the other modification of this invention. It is a circuit diagram showing the structure of the switching power supply device which concerns on the other modification of this invention. It is a circuit diagram showing the structure of the switching power supply device which concerns on the other modification of this invention. It is a circuit diagram showing the structure of the switching power supply device which concerns on the other modification of this invention. It is a circuit diagram showing the structure of the switching power supply device which concerns on the other modification of this invention. It is a circuit diagram showing the structure of the switching power supply device which concerns on the other modification of this invention.

Explanation of symbols

DESCRIPTION OF SYMBOLS 10 ... High voltage battery, 1 ... Bridge circuit, 11 ... Input smoothing capacitor, 2, 21, 22 ... Surge voltage suppression circuit, 3, 33 ... Transformer, 31, 31A, 331, 331A ... Primary winding, 32, 332 ... secondary winding, 31B, 31C, 331B ... auxiliary winding, 4, 41 ... rectifier circuit, 4A, 4B, 41A-41D ... rectifier diode, 5 ... smoothing circuit, 51 ... choke coil, 52 ... output smoothing capacitor , 6 ... drive circuit, 7 ... load, S1 to S4 ... switching element, D1 to D6 ... diode, C1 to C6 ... capacitor, Lr, LrA, LrB ... inductor, T1, T2 ... input terminal, T3, T4 ... output terminal , L1H ... Primary side high-pressure line, L1L ... Primary side low-pressure line, LO ... Output line, LG ... Ground line, P1 to P3 ... Connection point, CT ... Center tap, Vin ... Inflow power voltage, Vout ... DC output voltage, VP1~VP3 ... _{potential, V P1-P3, V P3} -P2 ... potential, V4A, V4B ... reverse voltage (surge voltage), Ir, I31, ID5, ID6, IC5, IC6 , I5, I6, I4A, I4B, I51, Ia to Iq, Ixa to Ixb, current, SG1 to SG4, switching signal, t0 to t20, timing, Td, dead time, φ, phase difference.

Claims (12)

A full bridge type bridge circuit configured to include four switching elements and generating an input AC voltage based on a DC input voltage;
A transformer having a primary winding and a secondary winding and transforming the input AC voltage to generate an output AC voltage;
A rectifier circuit that is provided on the secondary side of the transformer and includes a plurality of first rectifier elements, and generates a DC output voltage by rectifying the output AC voltage by the plurality of first rectifier elements;
A surge voltage suppression circuit configured to include two element pairs that are connected in parallel to the bridge circuit and are connected in parallel with a second rectifier element and a first capacitor element connected in reverse direction;
A resonant inductor that forms a first resonant circuit together with the first capacitive element;
A drive circuit for driving the bridge circuit,
The switching power supply device, wherein a resonance time of the first resonance circuit and a recovery time of the first rectifying element are set so as to satisfy the following conditional expression (1).
1/4 × {2π × (L × C) ^1/2 }> Trr1 (1)
However,
{2π × (L × C) ^1/2 }: Resonance time for one cycle in the first resonance circuit L: Inductance of the resonance inductor C: Capacitance value of the first capacitance element Trr1: Recovery time of the first rectification element To do. 2. The switching power supply according to claim 1, wherein a resonance time of the first resonance circuit and a recovery time of the second rectifying element are set so as to satisfy the following conditional expression (2): apparatus.
1/4 × {2π × (L × C) ^1/2 }> Trr2 (2)
However,
Trr2: The recovery time of the second rectifying element. The switching power supply according to claim 1 or 2, wherein in the surge voltage suppression circuit, the two element pairs are connected in series to each other. The primary winding of the transformer is H-bridge connected to one bridge circuit composed of two switching elements connected in series among the four switching elements and the two element pairs,
The resonance inductor is H-bridge connected to the other bridge circuit composed of the other two switching elements connected in series among the four switching elements and the two element pairs. The switching power supply device according to claim 3. 4. The switching power supply device according to claim 1, wherein the resonance inductor is disposed on a secondary side of the transformer. 5. The switching power supply according to any one of claims 1 to 5, wherein the transformer and the resonance inductor are magnetically independent of each other. An auxiliary winding is provided on the primary side of the transformer,
The switching power supply according to any one of claims 1 to 5, wherein the auxiliary winding and the resonance inductor share a magnetic flux with each other. A second capacitive element connected in parallel to each of the four switching elements;
The switching power supply according to any one of claims 1 to 7, wherein the resonance inductor constitutes a second resonance circuit together with the second capacitance element. The switching element is composed of a field effect transistor;
The switching power supply device according to claim 8, wherein the second capacitor element is configured by a parasitic capacitance of the field effect transistor. The switching power supply device according to any one of claims 1 to 9, wherein the first rectifier element is configured by a parasitic diode of a field effect transistor. The switching power supply according to any one of claims 1 to 10, wherein the rectifier circuit is a center tap rectifier circuit including two first rectifier elements. The switching power supply according to any one of claims 1 to 10, wherein the rectifier circuit is a full-bridge rectifier circuit configured to include four first rectifier elements.