JP3920896B2 - Linear position detector - Google Patents

Description

  The present invention relates to a linear position detection apparatus and can be applied to construction machines, automobiles, machine tools, and other fields.

Some conventional linear position detection devices use a potentiometer. However, since there is a sliding contact in the potentiometer, there is a difficulty in durability.
In addition, the conventionally known inductive position detection device includes a differential transformer as a linear position detection device and a resolver as a rotation position detection device. The differential transformer excites one primary winding in one phase, and differentially varies according to the linear position of the iron core that is linked to the detection target position at each of the two secondary windings that are differentially connected. The reluctance which changes continuously is produced, and the voltage amplitude level of the one-phase induction output AC signal obtained as a result indicates the linear position of the iron core. The resolver excites a plurality of primary windings in one phase, extracts an output AC signal indicating the amplitude function characteristics of the sine phase from the secondary winding for extracting the sine phase, and extracts from the secondary winding for extracting the cosine phase. An output AC signal indicating the amplitude function characteristic of the cosine phase is extracted. The two-phase resolver output is processed using a conversion circuit called a known R / D converter, and the phase value corresponding to the detected rotational position can be measured digitally.
Also, an output AC signal obtained by exciting a plurality of primary windings by a plurality of phases of AC signals such as a sine phase and a cosine phase, and electrically shifting the AC signal in accordance with a detection target linear position or rotation position. Is also known, which measures the linear position or rotational position of the detection object digitally by measuring the electrical phase shift amount of the output AC signal (for example, Japanese Patent Laid-Open No. 49-107758, JP-A-53-106065, JP-A-55-13891, JP-A-1-25286, etc.).

However, the potentiometer conventionally known as a linear position detection device has a difficulty in durability because of the sliding contact as described above. Moreover, it was not suitable for use in a poor environment.
In addition, conventionally known inductive position detection devices are generally non-contact in structure, and can be easily and inexpensively manufactured with a simple configuration of a coil and a magnetic body (iron piece, etc.) Although it can withstand use in a poor environment, it is necessary to provide a pair of primary and secondary coils.
The present invention has been made in view of the above-described points, and an object of the present invention is to provide a linear position detection device having a simplified configuration.

Linear position detecting device according to the present invention, comprises a coil portion and the magnetic response members, and one displacement in a non-contact manner relative linear with respect to the other of the before and Symbol coil portion magnetism-responsive member, in this A linear position detecting device for detecting a relative linear position of the magnetic response member with respect to the coil unit by obtaining a corresponding output signal from the coil unit, wherein the coil unit is excited by a predetermined AC signal. 1 and 2, each coil group is provided in a different arrangement along the direction of relative linear displacement of the magnetic response member with respect to the coil portion, whereby the first coil group is , Arranged to produce an output AC signal indicative of the amplitude function characteristics of the sine phase, depending on the relative linear position of the magnetic response member relative to the coil portion; The second coil group is arranged to generate an output AC signal indicating an amplitude function characteristic of a cosine phase according to a relative linear position of the magnetic response member with respect to the coil portion, and the first coil group Is composed of two-pole coils arranged so as to exhibit a magnetoresistive change characteristic of a sine phase and a minus sine phase with respect to the relative linear position of the magnetic response member, and a second coil group is formed of the magnetic response member. The coil is composed of two-pole coils arranged so as to exhibit the magnetoresistive change characteristics of the cosine phase and the minus cosine phase with respect to the relative linear position, and the coils of each pole in each coil group are excited by the AC signal. It is composed of only one coil, and the inductor according to the relative linear position of the magnetic response member with respect to each coil. An output AC voltage signal indicating an amplitude change based on a change in impedance is taken out from each one coil, and based on this, a sine phase and a minus sine phase output outputted from two coils of the first coil group are differentially outputted. And generating an output AC signal indicating the amplitude function characteristic of the sine phase, and differentially synthesizing outputs of the cosine phase and the minus cosine phase output from the two coils of the second coil group. An output AC signal indicating the amplitude function characteristic of the cosine phase is generated.

  According to the present invention, the coil portion includes first and second coil groups excited by a predetermined AC signal, and each coil group is along a direction of relative linear displacement of the magnetic response member with respect to the coil portion. The first coil group is arranged to generate an output AC signal that exhibits an amplitude function characteristic of a sine phase according to a relative linear position of the magnetic response member, Further, since the second coil group is arranged to generate an output AC signal indicating the amplitude function characteristic of the cosine phase, a linear position detection device that outputs a two-phase output AC signal of the sine phase and the cosine phase is provided. Can be provided. Further, each coil in each coil group consists of only one coil (primary coil) excited by the AC signal, and the amplitude based on the inductance change according to the position of the magnetic response member with respect to each one coil. An output AC voltage signal indicating a change is taken out from each one coil, and based on this, an output AC signal indicating an amplitude function characteristic of the sine phase and an amplitude function characteristic of the cosine phase are obtained from the first and second coil groups. Since the output AC signals shown are respectively generated, the linear position detecting device can be miniaturized by simplifying the coil configuration. Further, the output of the sine phase and the minus sine phase output from the two coils of the first coil group is differentially combined to generate an output AC signal indicating the amplitude function characteristic of the sine phase, By combining the output of the cosine phase and the minus cosine phase output from the two coils of the coil group to generate an output AC signal that indicates the amplitude function characteristics of the cosine phase, the temperature compensation characteristics are also excellent. A highly accurate linear position detecting device can be provided.

  According to the present invention, various other embodiments can be taken, the details of which are shown below by way of example.

Hereinafter, some representative examples of embodiments of the present invention will be described with reference to the accompanying drawings. The illustrated examples can be combined with each other, and these combinations are also included in the practice of the present invention.
FIG. 1 is a sectional view in the axial direction of a linear position detecting apparatus 10 according to the present invention. In this embodiment, since the linear position detection device 10 is used as an inclinometer, it is hereinafter referred to as an inclinometer 10. The storage body 1 made of a non-magnetic material such as plastic or stainless steel has a shape like a winding shaft, and a passage 1a bent downward is provided inside the shaft as shown in the figure. . A magnetic response member 3 of an appropriate size or amount is accommodated in the passage 1a so as to be movable according to gravity. In the example of FIG. 1, the magnetic response member 3 is made of a spherical magnetic material such as iron. One or a plurality of coils 11 to 15 and 21 to 24 are sequentially arranged and wound around the passage 1a of the storage body 1, that is, around the winding shaft. The coil part 2 is comprised by these coils 11-15, 21-24. Note that both ends of the passage 1a are closed so that the internal magnetic response member 3 does not pop out.

With the above configuration, the inductive coupling in the coil portion 2 changes according to the linear position of the magnetic response member 3 in the passage 1a, that is, the relative linear position of the magnetic response member 3 with respect to the coil portion 2, and the output corresponding to this. A signal can be obtained from the coil section 2. Accordingly, a detection output signal corresponding to the linear position of the magnetic response member 3 in the passage 1a can be obtained from the coil section 2.
Here, since the passage 1a of the storage body 1 is bent downward, when the storage body 1 is in a horizontal position, the magnetic response member 3 in the passage 1a always has a predetermined weight due to its own weight. It is located at the position (the lowest position corresponding to inclination 0). When the storage body 1 is tilted, the magnetic response member 3 is linearly displaced along the passage 1a accordingly, and a detection output signal corresponding to the linear position of the magnetic response member 3 in the passage 1a is obtained from the coil section 2. It is done. Therefore, the output signal of the coil part 2 responds to the inclination of the storage body 1, and can be appropriately used as a detection signal of the inclination.

  An arbitrary detection principle can be used as the detection principle by the coil unit 2. As a simple example, there can be a pickup coil system. That is, the linear position of the magnetic response member 3 in the passage 1a can be detected by specifying the coil closest to the magnetic response member 3 based on the output signal levels of the plurality of coils arranged along the passage 1a. Accordingly, the degree of inclination of the storage body 1 can be detected / detected. As an example for detecting the linear position of the magnetic response member 3 more finely, the coil unit 2 can be configured according to a linear differential transformer principle. That is, one or a plurality of linear differential transformers are configured as the coil unit 2, and the pickup is obtained by combining the output voltage value of the linear differential transformer and data indicating which linear differential transformer the output is obtained from. The linear position of the magnetic response member 3 can be detected with finer accuracy than the coil method.

In order to be able to detect the linear position of the magnetic response member 3 with a finer accuracy and more accurately, the coil unit 2 may be configured according to the resolver principle.
When the coil unit 2 is configured in accordance with the resolver principle, it includes primary coils 11 to 15 that are excited by a one-phase AC signal and a plurality of secondary coils 21 to 24. Each of the secondary coils 21 to 24 is arranged at a predetermined interval along the passage 1a. On the other hand, since it is commonly excited by a one-phase AC signal, the number of primary coils 11 to 15 may be one or an appropriate plural number, and the arrangement thereof may be appropriate. However, the primary coils 11 to 15 are appropriately separated and arranged so that the secondary coils 21 to 24 are sandwiched between the primary coils 11 to 24 as shown in FIG. This is preferable because the generated magnetic field can be effectively applied to the individual secondary coils 21 to 24 and the magnetic response member 3 can be effectively affected by the magnetic field.

Depending on the linear position of the magnetic response member 3 in the passage 1a, the corresponding position of the magnetic response member 3 with respect to the coil portion 2 changes, so that the magnetic coupling between the primary coils 11-15 and the secondary coils 21-24 is achieved. The induction output AC signal that is changed according to the linear position and is amplitude-modulated according to the linear position has different amplitude function characteristics according to the displacement of the arrangement of the secondary coils 21 to 24, and each 2 It is induced in the secondary coils 21-24. The induction output AC signals induced in the secondary coils 21 to 24 have the same electrical phase and the same amplitude because the primary coils 11 to 15 are commonly excited by the one-phase AC signal. The function has a phase shifted according to the displacement of the arrangement of the secondary coils 21 to 24.
That is, the amplitude function of the inductive output AC signals generated in the four secondary coils 21 to 24 can be set so as to exhibit desired characteristics, and when configured as a resolver type position detection device, each 2 The amplitude function of the inductive output AC signal generated in the secondary coils 21 to 24 can be set so as to correspond to a sine function, a cosine function, a minus sine function, and a minus cosine function, respectively. Depending on various conditions, the arrangement of each coil can change slightly, so the arrangement and number of turns of each coil can be adjusted as appropriate to obtain the desired functional characteristics, or the secondary output level can be adjusted by electrical amplification. The desired amplitude function characteristic is finally obtained.

  For example, if the output of the secondary coil 21 corresponds to a sine function (indicated by s in the figure), the output of the secondary coil 23 arranged with a predetermined distance (for example, p / 2) with respect to this. Is set to correspond to a minus sine function (/ s (s bar) is added in the figure), and a first output having an amplitude function of a sine function is obtained by differentially synthesizing both outputs. An AC signal can be obtained. In addition, the output of the secondary coil 22 that is shifted from the secondary coil 21 corresponding to the sine function output by a half of the predetermined distance (for example, p / 4) is a cosine function (c is added in the figure). Correspondingly, the output of the secondary coil 24 arranged so as to be shifted by p / 2 is set so as to correspond to a minus cosine function (/ c (c bar) is added in the figure). The second output AC signal having an amplitude function of a cosine function can be obtained by differentially synthesizing the outputs. In the specification, for convenience of description, a bar symbol indicating inversion is described as “/ (slash)”, which corresponds to the bar symbol in the figure.

  FIG. 2 is a circuit diagram of the primary and secondary coils of the coil unit 2, and a common excitation AC signal (indicated by sin ωt for convenience of description) is applied to the primary coils 11 to 15. In response to the excitation of the primary coil, an AC signal having an amplitude value corresponding to the linear position of the magnetic response member 3 that changes corresponding to the inclination angle α of the housing 1 is induced in each of the secondary coils 21 to 24. The Each induced voltage level indicates two-phase function characteristics sinθ and cosθ having a phase angle θ correlated to the tilt angle α and opposite-phase function characteristics −sinθ and −cosθ. That is, the inductive output signals of the secondary coils 21 to 24 have two-phase function characteristics sinθ and cosθ and opposite phase function characteristics −sinθ and −cosθ corresponding to the phase angle θ correlated with the inclination angle α. It can be set so as to be output in an amplitude-modulated state. For convenience of explanation, coefficients according to other conditions such as the number of turns of the coil are omitted, the secondary coil 21 is set as a sine phase, the output signal is indicated as “sin θ · sin ωt”, and the output is output as the secondary coil 22 is set as a cosine phase. The signal is indicated by “cos θ · sin ωt”. The output signal is indicated by “−sin θ · sin ωt” with the secondary coil 23 as a negative sine phase, and the output signal is indicated with “−cos θ · sin ωt” with the secondary coil 24 as a negative cosine phase. A first output AC signal A (= 2sinθ · sinωt) having an amplitude function of a sine function is obtained by differentially combining the induction outputs of the sine phase and the minus sine phase. Further, a second output AC signal B (= 2 cos θ · sin ωt) having an amplitude function of a cosine function is obtained by differentially combining the induction outputs of the cosine phase and the minus cosine phase. In order to simplify the expression, the coefficient “2” is omitted, and in the following, the first output AC signal A is represented by “sinθ · sinωt”, and the second output AC signal B is represented by “cosθ · sinωt”. ".

Thus, the first output AC signal A = sinθ · sinωt having the first function value sinθ having the phase angle θ correlated with the inclination angle α as the amplitude value, and the second function value cosθ corresponding to the same phase angle θ. Is output as a second output AC signal B = cos θ · sin ωt. According to such a coil configuration, two output AC signals A and B (signatures) having an in-phase AC and a two-phase amplitude function similar to those obtained in a resolver conventionally known as a rotary position detecting device. It can be understood that the output and the cosine output can be obtained from the coil section 2.
The two-phase output AC signals (A = sin θ · sin ωt and B = cos θ · sin ωt) output from the coil unit 2 can be used in the same manner as the output of a conventionally known resolver. For example, as shown in FIG. 2, the output AC signals A and B of the coil unit 2 are input to an appropriate digital phase detection circuit 40, and the phase value θ of the sine function sin θ and the cosine function cos θ is detected by a digital phase detection method. The digital data Dθ of the phase angle θ can be obtained. Therefore, the digital data Dθ can be used as detection data for the inclination angle α. As a digital phase detection method employed in the digital phase detection circuit 40, a known RD (resolver-digital) converter may be applied, or a new method developed by the present inventors may be employed. Good.

The shape of the magnetic response member 3 is not limited to a sphere, and may be a cylindrical shape or other appropriate shape. Further, the magnetic response member 3 is not limited to a solid member, and may be made of a non-fixed object such as a magnetic fluid or a magnetic powder. The material of the magnetic response member 3 is not limited to a magnetic material, and may be a good conductor such as copper.
FIG. 3 shows some modified examples of the magnetic response member 3. FIG. 3A shows a cylindrical solid magnetic response member 3a. (B) shows an example in which an appropriate amount of the magnetic fluid 3 b is used as the magnetic response member 3. (C) shows an example in which an appropriate amount of the magnetic powder 3 c is used as the magnetic response member 3. The magnetic powder 3c is not limited to a fine powder, and may be a granular material such as iron sand. Although not shown in particular, depending on the purpose of use, when a solid magnetic response member 3 as shown in FIG. 1 is used, a nonmagnetic viscous fluid is sealed in the passage 1a, and the magnetic response member 3 corresponding to the inclination is used. An appropriate amount of dumping action may be exerted on the movement of the.

In FIG. 1, the appearance of the storage body 1 is like a bobbin or a winding shaft, but is not limited to this, and it may have a bent shape as shown in FIG. 4. In that case, the coil portion 2 is fitted around the bent tube (housing 1).
In each of the above embodiments, the number and arrangement of the primary and secondary coils in the coil unit 2 can be variously modified and changed in design. Further, the number of phases of the secondary coil output signal is not limited to the two phases of sine and cosine, but may be another form, for example, a three-phase type shifted by 120 degrees.

The inclinometer 10 as described above can detect an inclination in only one direction (one axis) along the direction of the passage 1a. For example, when the target inclination direction is determined in a predetermined direction as in the case of detecting the inclination of the work arm of the construction machine, one inclinometer 10 may be provided.
However, if it is desired to detect the inclination in at least two directions as in the case of detecting the front-rear inclination and the horizontal inclination in the vehicle body or other cases, at least two inclinometers 10 are different from each other. It may be arranged in the direction of. For example, FIG. 5 schematically shows an example, and two inclinometers 10X and 10Y are combined so as to cross each other at an angle of 90 degrees. Each inclinometer 10X, 10Y has the same configuration as the inclinometer 10 described above. Thus, the tilt (tilt component) in the X-axis direction of the detection target can be detected by the inclinometer 10X, and the tilt (tilt component) in the Y-axis direction of the detection target can be detected by the inclinometer 10Y.

FIG. 6 shows a modification of FIG. 5 in which the curvature of the passage 1a in the housing 1 of each of the inclinometers 10X ′ and 10Y ′ corresponding to the X and Y axes is greatly increased. Thus, as the curvature of the passage 1a is increased, the displacement amount of the magnetic response member 3 with respect to the actual inclination angle α becomes relatively small (it becomes difficult to move in the passage 1a). Therefore, by adjusting the curvature of the passage 1a in the storage body 1, it is possible to adjust the sensitivity of the inclination angle α to be detected or the detectable angle range.
As can be easily understood, having the detection sensitivity or the range adjustment function by adjusting the curvature of the passage 1a in this way, according to the present invention, whether or not there is an inclination in response to an inclination of a predetermined range or a predetermined range. It means that an inclinometer capable of detecting can also be provided. Of course, even in the uniaxial inclinometer 10, detection sensitivity or range adjustment by adjusting the curvature of the passage 1a is possible.

FIG. 7 shows an example in which a known RD (resolver-digital) converter is applied as the digital phase detection circuit 40. Resolver type two-phase output AC signals A = sin θ · sin ωt and B = cos θ · sin ωt output from the secondary coils 21 to 24 of the coil unit 2 are input to the analog multipliers 30 and 31, respectively. The sequential phase generation circuit 32 generates digital data having a phase angle φ, and the sine / cosine generation circuit 33 generates analog signals having a sine value sinφ and a cosine value cosφ corresponding to the phase angle φ. The multiplier 30 multiplies the sine-phase output AC signal A = sinθ · sinωt by the cosine value cosφ from the sine / cosine generation circuit 33 to obtain “cosφ · sinθ · sinωt”. The other multiplier 31 multiplies the output AC signal B = cosθ · sinωt of the cosine phase by the sine value sinφ from the sine / cosine generation circuit 33 to obtain “sinφ · cosθ · sinωt”. The subtractor 34 obtains the difference between the output signals of the multipliers 30 and 31 and sequentially controls the phase generation operation of the phase generation circuit 32 by the output of the subtractor 34 as follows. That is, the generated phase angle φ of the sequential phase generation circuit 32 is first reset to 0, and then increases sequentially, and stops increasing when the output of the subtractor 34 becomes 0. The output of the subtractor 34 becomes zero when “cosφ · sinθ · sinωt” = “sinφ · cosθ · sinωt” is satisfied, that is, φ = θ is satisfied, and the phase generation circuit 32 sequentially The digital data of the phase angle φ coincides with the digital value of the phase angle θ of the amplitude function of the output AC signals A and B. Accordingly, a reset trigger is periodically applied at an arbitrary timing to sequentially reset the generated phase angle φ of the phase generation circuit 32 to 0, and the increment of the phase angle φ is started, and the output of the subtractor 34 is set to 0. Then, the increment is stopped and digital data of the phase angle θ is obtained.
It is known that the sequential phase generating circuit 32 includes an up / down counter and a VCO, and the VCO is driven by the output of the subtractor 34 to control the up / down counting operation of the up / down counter. In that case, a periodic reset trigger is not necessary.

An error occurs in the electrical AC phase ωt in the secondary output AC signal due to changes in the impedance of the primary and secondary coils of the coil unit 2 due to temperature change or the like. In the phase detection circuit as described above, sinωt Convenient because the phase error is automatically canceled out. On the other hand, in a system in which an electrical phase shift is generated in a one-phase output AC signal by exciting with a conventionally known two-phase AC signal (for example, sinωt and cosωt), such a temperature change is caused. The output phase error based on it cannot be removed.
By the way, since the phase detection circuit composed of the conventional RD converter as described above is a follow-up comparison method, there is a problem that a clock delay occurs when φ is followed up and the response is poor.
Therefore, the present inventors have developed a novel phase detection circuit as described below, and it is convenient to use it.

FIG. 8 shows an embodiment of a novel digital phase detection circuit 40 applied to the tilt detection apparatus according to the present invention.
In FIG. 8, in the detection circuit unit 41, a counter 42 counts a predetermined high-speed clock pulse CK, and based on the count value, an excitation AC signal (for example, sin ωt) is generated from the excitation signal generation circuit 43, and the coil unit 2 Primary coils 11 to 15 are provided. The modulo number of the counter 42 corresponds to one cycle of the excitation AC signal. For convenience of explanation, it is assumed that 0 of the count value corresponds to 0 phase of the reference sine signal sinωt. Two-phase output AC signals A = sin θ · sin ωt and B = cos θ · sin ωt output from the secondary coils 21 to 24 of the coil unit 2 are input to the detection circuit unit 41.

  In the detection circuit unit 41, the first AC output signal A = sin θ · sin ωt is input to the phase shift circuit 44, and its electrical phase is phase-shifted by a predetermined amount, for example, advanced by 90 degrees, and phase-shifted AC signal A ′ = sin θ · cos ωt is obtained. In addition, the detection circuit unit 41 is provided with an addition circuit 45 and a subtraction circuit 46. In the addition circuit 45, the phase-shifted AC signal A ′ = sinθ · cosωt output from the phase shift circuit 44 and the coil The second AC output signal B = cos θ · sin ωt output from the secondary coils 21 to 24 of the unit 10 is added, and the added output is expressed by an abbreviated expression B + A ′ = cos θ · sin ωt + sin θ · cos ωt = sin (ωt + θ). A first electrical AC signal Y1 is obtained. In the subtracting circuit 46, the phase-shifted AC signal A ′ = sin θ · cos ωt and the second AC output signal B = cos θ · sin ωt are subtracted, and as a subtraction output, B−A ′ = cos θ · sin ωt− A second electrical AC signal Y2 that can be expressed by the following equation is obtained: sinθ · cosωt = sin (ωt−θ). In this way, the first electrical AC signal Y1 = sin (ωt + θ) having the electrical phase angle (+ θ) shifted in the positive direction corresponding to the linear position (x) of the magnetic response member 3 in the passage 1a. ) And a second electrical AC signal Y2 = sin (ωt−θ) having an electrical phase angle (−θ) shifted in the negative direction corresponding to the same linear position (x). Each is obtained by processing.

  The output signals Y1 and Y2 of the adder circuit 45 and the subtractor circuit 46 are input to zero cross detection circuits 47 and 48, respectively, and the respective zero crosses are detected. As a method of detecting the zero cross, for example, a zero cross in which the amplitude values of the signals Y1 and Y2 change from negative to positive, that is, zero phase is detected. Zero-cross detection pulses detected by the circuits 47 and 48, that is, zero phase detection pulses are input to the latch circuits 49 and 50 as latch pulses LP1 and LP2. In the latch circuits 49 and 50, the count value of the counter 42 is latched at the timing of the respective latch pulses LP1 and LP2. As described above, the modulo number of the counter 42 corresponds to one cycle of the excitation AC signal, and the count value 0 corresponds to the 0 phase of the reference sine signal sinωt. The data D1 and D2 latched in the latch circuits 49 and 50 respectively correspond to the phase shifts of the output signals Y1 and Y2 with respect to the reference sine signal sinωt. The outputs of the latch circuits 49 and 50 are input to the error calculation circuit 51 to calculate “(D1 + D2) / 2”. This calculation may actually be performed by shifting the addition result of the binary data “D1 + D2” to the lower one bit.

Here, in consideration of the influence of the length of the wiring cable between the coil unit 2 and the detection circuit unit 41 and the impedance change caused by the temperature change or the like in each primary and secondary coil of the coil unit 2, When the phase fluctuation error of the output signal is indicated by “± d”, the above signals in the detection circuit unit 41 are expressed as follows.
A = sin θ · sin (ωt ± d)
A ′ = sin θ · cos (ωt ± d)
B = cos θ · sin (ωt ± d)
Y1 = sin (ωt ± d + θ)
Y2 = sin (ωt ± d−θ)
D1 = ± d + θ
D2 = ± d−θ

That is, since each phase shift measurement data D1, D2 performs phase shift count using the reference sine signal sinωt as a reference phase, a value including the phase variation error “± d” is obtained as described above. . Therefore, by calculating “(D1 + D2) / 2” in the error calculation circuit 51,
(D1 + D2) / 2 = {(± d + θ) + (± d−θ)} / 2
= ± 2d / 2 = ± d
Thus, the phase variation error “± d” can be calculated.

The data of the phase fluctuation error “± d” obtained by the error calculation circuit 51 is given to the subtraction circuit 52, and is subtracted from one phase shift measurement data D1. That is, in the subtraction circuit 52, “D1− (± d)” is subtracted.
D1− (± d) = ± d + θ− (± d) = θ
Thus, digital data indicating the correct detected phase difference θ from which the phase fluctuation error “± d” has been removed is obtained. Thus, according to the present invention, it can be understood that the phase variation error “± d” is canceled out and only the correct phase difference θ is extracted.

This point will be further described with reference to FIG. FIG. 9 shows a waveform near the zero phase of the sine signal sinωt that is a reference for phase measurement and the first and second AC signals Y1 and Y2, and FIG. 9A shows a positive phase fluctuation error ( In the case of + d), (b) shows the case of minus (-d). In the case of FIG. 5A, the zero phase of the first signal Y1 advances by “θ + d” with respect to the zero phase of the reference sine signal sinωt, and the corresponding phase difference detection data D1 becomes “θ + d”. The corresponding phase difference is shown. Further, the zero phase of the second signal Y2 is delayed by “−θ + d” with respect to the zero phase of the reference sine signal sinωt, and the corresponding phase difference detection data D2 is a phase difference corresponding to “−θ + d”. Indicates. In this case, the error calculation circuit 51
(D1 + D2) / 2 = {(+ d + θ) + (+ d−θ)} / 2
= + 2d / 2 = + d
Thus, the phase fluctuation error “+ d” is calculated. Then, by the subtraction circuit 52,
D1 − (+ d) = + d + θ − (+ d) = θ
Is calculated, and the correct phase difference θ is extracted.

In the case of FIG. 9B, the zero phase of the first signal Y1 is advanced by “θ-d” with respect to the zero phase of the reference sine signal sinωt, and the corresponding phase difference detection data D1 is “θ -D "represents the phase difference. Further, the 0 phase of the second signal Y2 is delayed by “−θ−d” with respect to the 0 phase of the reference sine signal sinωt, and the corresponding phase difference detection data D2 becomes “−θ−d”. The corresponding phase difference is shown. In this case, the error calculation circuit 51
(D1 + D2) / 2 = {(− d + θ) + (− d−θ)} / 2
= -2d / 2 = -d
Thus, the phase fluctuation error “−d” is calculated. Then, by the subtraction circuit 52,
D1 − (− d) = − d + θ − (− d) = θ
Is calculated, and the correct phase difference θ is extracted.
In the subtracting circuit 52. It will be understood that “D2− (± d)” may be subtracted, and in principle, data (−θ) reflecting the correct phase difference θ can be obtained in the same manner as described above.

  Further, as can be understood from FIG. 9, the electrical phase difference between the first signal Y1 and the second signal Y2 is 2θ, and it is always accurate to cancel the phase fluctuation error “± d” between the two. This indicates a double value of the phase difference θ. Accordingly, the configuration of the circuit portion including the latch circuits 49 and 50, the error calculation circuit 51, the subtraction circuit 52, and the like in FIG. 8 is appropriately changed to a configuration for directly obtaining the electrical phase difference 2θ of the signals Y1 and Y2. It may be. For example, from the generation time point of the pulse LP1 corresponding to the 0 phase of the first signal Y1 output from the zero cross detection circuit 47, the pulse LP2 corresponding to the 0 phase of the second signal Y2 output from the zero cross detection circuit 48 is generated. Digital data corresponding to the electrical phase difference (2θ) that offsets the phase fluctuation error “± d” can be obtained by gating the period up to the point of occurrence by appropriate means and counting the gate period. If this is shifted down by 1 bit, data corresponding to θ can be obtained.

  In the above embodiment, the latch circuit 49 for latching + θ and the latch circuit 50 for latching −θ latch the output of the same counter 42, and the sign of the latched data is positive or negative. Is not specifically mentioned. However, an appropriate design process may be applied to the positive and negative signs of the data in accordance with the spirit of the present invention. For example, assuming that the modulo number of the counter 42 is 4096 (decimal number display), the digital counts 0 to 4095 may be appropriately processed according to the phase angle of 0 degrees to 360 degrees. In the simplest design example, the most significant bit of the count output of the counter 42 is a sign bit, the digital counts 0 to 2047 correspond to +0 degrees to +180 degrees, and the digital counts 2048 to 4095 are set to −180 degrees to −0 degrees. Correspondingly, arithmetic processing may be performed. Alternatively, as another example, by converting the input data or output data of the latch circuit 50 into a two's complement, the digital count 4095-0 can correspond to a negative angle data expression of -360 degrees to -0 degrees. May be.

By the way, there is no particular problem when the tilt angle α is stationary, but when the detection target tilt angle α changes with time, the corresponding linear position (x), that is, the phase angle θ also changes with time. It will be. In that case, the phase shift amount θ of each of the output signals Y1 and Y2 of the adder circuit 45 and the subtractor circuit 46 is not a constant value but shows a dynamic characteristic that changes with time according to the moving speed, and this is expressed as θ ( t), each output signal Y1, Y2 is
Y1 = sin {ωt ± d + θ (t)}
Y2 = sin {ωt ± d−θ (t)}
It becomes. That is, with respect to the frequency of the reference signal sinωt, the fast-phase output signal Y1 transitions in a frequency increasing direction according to + θ (t), and the slow-phase output signal Y2 according to −θ (t). The frequency transitions in the direction of decreasing frequency. Under such dynamic characteristics, the period of each signal Y1, Y2 transitions in the opposite direction one after another for each period of the reference signal sin ωt, so that each latch data D1, The measurement time reference for D2 is different, and an accurate phase variation error “± d” cannot be obtained by simply calculating both data D1 and D2 by the circuits 51 and 52.

The simplest method for avoiding such a problem is that in the configuration of FIG. 8, the output when the tilt angle α is moving in time is ignored, and only the output in the stationary state is used, and the stationary To limit the function of the device to measure the tilt angle α when the condition is obtained. That is, the present invention may be implemented for such a limited purpose.
However, it is desirable to be able to accurately detect the phase difference θ corresponding to the detection target inclination angle α every moment even when the detection target inclination angle α is changing with time. Therefore, in order to solve the above problems, the phase difference θ corresponding to the detection target inclination angle α is detected every moment even when the detection target inclination angle α is changing over time. The improvement measures made possible will be described with reference to FIG.

FIG. 10 shows an example of modification of the error calculation circuit 51 and the subtraction circuit 52 in the detection circuit unit 41 of FIG. 8, and the configuration of other parts not shown is the same as FIG. Good. When the phase difference θ corresponding to the inclination angle θ when the detection target inclination angle θ changes with time is expressed by + θ (t) and −θ (t), the output signals Y1 and Y2 It can be expressed as follows. The phase shift measured value data D1 and D2 obtained by the latch circuits 49 and 50 corresponding to the
D1 = ± d + θ (t)
D2 = ± d−θ (t)
It becomes.
In this case, ± d + θ (t) repeatedly changes in time in the plus direction in the range of 0 ° to 360 ° in accordance with the time change of θ. Further, ± d−θ (t) repeatedly changes in time in the minus direction in the range of 360 degrees to 0 degrees in accordance with the time change of θ. Therefore, there are cases where ± d + θ (t) ≠ ± d−θ (t), but there are also cases where the changes of both intersect, and in this case, ± d + θ (t) = ± d−θ (t) holds. . As described above, when ± d + θ (t) = ± d−θ (t) is satisfied, the electrical phases of the output signals Y1 and Y2 coincide with each other, and the latch corresponding to the respective zero-cross detection timings. The generation timings of the pulses LP1 and LP2 are the same.

In FIG. 10, the coincidence detection circuit 53 detects that the generation timings of the latch pulses LP1 and LP2 corresponding to the zero cross detection timings of the output signals Y1 and Y2 are coincident, and in response to this detection, coincidence detection pulses. Generate an EQP. On the other hand, in the time variation determination circuit 54, the detection target inclination angle α changes with time by an appropriate means (for example, by detecting presence or absence of temporal change in the value of one phase difference measurement data D1). It is determined that the mode is selected, and the time variation mode signal TM is output in accordance with this determination.
A selector 55 is provided between the error calculation circuit 51 and the subtraction circuit 52, and when the time variation mode signal TM is not generated, that is, TM = “0”, that is, the detection target inclination angle θ changes with time. If not, the output of the error calculation circuit 51 applied to the selector input B is selected and input to the subtraction circuit 52. Thus, the circuit of FIG. 10 when the input B of the selector 55 is selected operates equivalently to the circuit of FIG. That is, when the detection target is stationary, the output data of the error calculation circuit 51 is directly given to the subtraction circuit 52 via the input B of the selector 55, and operates in the same manner as the circuit of FIG.

  On the other hand, when the time variation mode signal TM is generated, that is, when TM = “1”, that is, the detection target is temporally changing, the output of the latch circuit 56 applied to the input A of the selector 55 is selected. To the subtracting circuit 52. When the time variation mode signal TM is “1” and the coincidence detection pulse EQP is generated, the condition of the AND gate 57 is satisfied, and a pulse responding to the coincidence detection pulse EQP is output from the AND gate 57. A latch instruction is given to the latch circuit 56. The latch circuit 56 latches the output count data of the counter 42 in response to the latch instruction. Here, when the coincidence detection pulse EQP is generated, the output of the counter 42 is simultaneously latched in the latch circuits 49 and 50, so that D1 = D2, and the data latched in the latch circuit 56 is D1 or D2 ( However, this corresponds to D1 = D2).

The coincidence detection pulse EQP is generated when the zero cross detection timings of the output signals Y1 and Y2 coincide, that is, when “± d + θ (t) = ± d−θ (t)” is established. Since the data latched in the latch circuit 56 in response to D1 corresponds to D1 or D2 (where D1 = D2),
(D1 + D2) / 2
Is equivalent to This means
(D1 + D2) / 2 = [{± d + θ (t)} + {± d−θ (t)}] / 2
= 2 (± d) / 2 = ± d
This means that the data latched by the latch circuit 56 accurately indicates the phase fluctuation error “± d”.

Thus, when the detection target fluctuates in time, data accurately indicating the phase fluctuation error “± d” is latched by the latch circuit 56 in accordance with the coincidence detection pulse EQP, and the output data of the latch circuit 56 is The signal is supplied to the subtraction circuit 52 via the input A of the selector 55. Therefore, the subtracting circuit 52 can obtain data θ (or θ (t) in the case of temporal variation) that accurately responds only to the detection target inclination angle θ from which the phase variation error “± d” has been removed.
In FIG. 10, the AND gate 57 may be omitted, and the coincidence detection pulse EQP may be directly applied to the latch control input of the latch circuit 56.
Further, not only the output count data of the counter 42 but also the output data “± d” of the error calculation circuit 51 may be latched in the latch circuit 56 as shown by a broken line in FIG. In this case, the output timing of the output data of the error calculation circuit 51 corresponding to the generation timing of the coincidence detection pulse EQP is somewhat delayed because of the circuit operation delay of the latch circuits 49 and 50 and the error calculation circuit 51. Therefore, it is preferable to latch the output of the error calculation circuit 51 in the latch circuit 56 after performing an appropriate time delay adjustment.
Further, it can be understood that when the detection circuit unit 41 is configured in consideration of only dynamic characteristics, the circuit 51 and the selector 55 in FIG. 10 and the one latch circuit 49 or 50 in FIG. 1 may be omitted. I will.

FIG. 11 shows another embodiment of the phase difference detection calculation method that can cancel the phase fluctuation error “± d”.
The first and second AC output signals A and B of the resolver type output from the secondary coils 21 to 24 of the coil unit 2 are input to the detection circuit unit 60, and the first and second AC output signals A and B are input in the same manner as in the example of FIG. The AC output signal A = sin θ · sin ωt is input to the phase shift circuit 44, and the electrical phase is phase-shifted by a predetermined amount to obtain the phase-shifted AC signal A ′ = sin θ · cos ωt. The subtracting circuit 46 subtracts the phase-shifted AC signal A ′ = sin θ · cos ωt and the second AC output signal B = cos θ · sin ωt, and outputs B−A ′ = cos θ · An electrical AC signal Y2 can be obtained that can be expressed by the following equation: sinωt−sinθ · cosωt = sin (ωt−θ). The output signal Y2 of the subtraction circuit 46 is input to the zero cross detection circuit 48, and the latch pulse LP2 is output in response to the zero cross detection and input to the latch circuit 50.

  The embodiment of FIG. 11 differs from the embodiment of FIG. 8 in that a reference for measuring the phase shift amount θ from the AC signal Y2 = sin (ωt−θ) including the electrical phase shift corresponding to the detection target. The phase is different. In the example of FIG. 8, the reference phase when measuring the phase shift amount θ is the zero phase of the reference sine signal sinωt, which is not input to the coil unit 2 of the inclinometer 10. This does not include a phase variation error “± d” based on a change in coil impedance due to a change or the like or other various factors. Therefore, in the example of FIG. 8, two AC signals Y1 = sin (ωt + θ) and Y2 = sin (ωt−θ) are formed, and the phase difference error “± d” is obtained by obtaining the electrical phase difference. I try to offset it. On the other hand, in the embodiment of FIG. 11, a reference phase for measuring the phase shift amount θ is formed based on the first and second AC output signals A and B output from the coil unit 2. The reference phase itself includes the phase fluctuation error “± d” to eliminate the phase fluctuation error “± d”.

  That is, in the detection circuit unit 60, the first and second AC output signals A and B output from the coil unit 2 are input to the zero cross detection circuits 61 and 62, respectively, and each zero cross is detected. Note that the zero cross detection circuits 61 and 62 respond to both the zero cross (so-called 0 phase) in which the amplitude values of the input signals A and B change from negative to positive and the zero cross (so-called 180 degree phase) in which the amplitude changes from positive to negative. A zero cross detection pulse is output. This is because sin θ and cos θ that determine the positive / negative polarity of the amplitudes of the signals A and B are arbitrarily positive or negative depending on the value of θ, and in order to detect a zero cross every 360 degrees based on the combination of both, This is because it is necessary to detect a zero cross every 180 degrees. The zero-cross detection pulses output from both the zero-cross detection circuits 61 and 62 are OR-combined by an OR circuit 63, and the output of the OR circuit 63 is an appropriate ½ frequency-dividing pulse circuit 64 (for example, 1 such as a T-flip-flop). / 2 frequency dividing circuit and pulse output AND gate), and every other zero cross detection pulse is taken out, and zero cross every 360 degrees, that is, zero cross detection pulse corresponding to only 0 phase is a reference phase signal. Output as a pulse RP. This reference phase signal pulse RP is given to the reset input of the counter 65. The counter 65 continuously counts a predetermined clock pulse CK, and the count value is repeatedly reset to 0 according to the reference phase signal pulse RP. The output of the counter 65 is input to the latch circuit 50, and the count value is latched in the latch circuit 50 at the generation timing of the latch pulse LP2. Data D latched in the latch circuit 50 is output as measurement data of the phase difference θ corresponding to the detection target position x.

  The first and second AC output signals A and B output from the coil unit 2 are A = sin θ · sin ωt and B = cos θ · sin ωt, respectively, and the electrical phases are the same. Therefore, the zero crossing should be detected at the same timing, but since the amplitude coefficient varies with the sine function sinθ and the cosine function cosθ, either amplitude level may be 0 or close to 0. If on the other hand, virtually no zero cross can be detected. Therefore, in this embodiment, zero cross detection processing is performed for each of the two AC output signals A = sin θ · sin ωt and B = cos θ · sin ωt, and one of the two zero cross detection outputs is OR-synthesized so that one of the amplitude levels is amplitude level. Even if the zero cross detection is impossible due to the small size, the zero cross detection output signal having the larger amplitude level on the other side can be used.

  In the case of the example of FIG. 11, assuming that the phase fluctuation error due to the coil impedance change or the like of the coil unit 2 is “−d”, for example, the AC signal Y2 output from the subtraction circuit 46 is shown in FIG. As shown, Y2 = sin (ωt−d−θ). In this case, the output signals A and B of the coil unit 2 have amplitude values sin θ and cos θ corresponding to the angle θ, respectively, and A = sin θ · sin (ωt−d) as illustrated in FIG. , B = cos θ · sin (ωt−d), and the like includes a phase fluctuation error. Accordingly, the reference phase signal pulse RP obtained at the timing as shown in FIG. 12C based on this zero cross detection is shifted from the 0 phase of the original reference sine signal sinωt by the phase variation error −d. . Accordingly, if the phase shift amount of the output AC signal Y2 = sin (ωt−d−θ) of the subtracting circuit 46 is measured using the reference phase signal pulse RP as a reference, an accurate value θ with the phase fluctuation error −d removed. Will be obtained.

  In addition, if apparatus conditions, such as the wiring length of the coil part 2, are determined, the impedance change will depend mainly on temperature. Then, the phase fluctuation error ± d corresponds to data indicating the temperature of the surrounding environment where the tilt detection device is provided. Therefore, in the circuit having the circuit 51 for calculating the phase fluctuation error ± d as in the embodiment of FIG. 9, the data of the phase fluctuation error ± d obtained there can be appropriately output as temperature detection data. Therefore, according to such a configuration of the present invention, not only the inclination of the detection target can be detected by one inclination detection device, but also data indicating the temperature of the surrounding environment of the inclination detection device can be obtained. It has an excellent effect of being able to. Of course, there is also an excellent effect that high-precision detection is possible in response to the inclination of the detection target without being affected by the impedance change on the sensor side due to temperature change or the length of the wiring cable. . Further, since the examples of FIGS. 8 and 10 are methods for measuring the phase difference in the AC signal, it is possible to perform detection with excellent high-speed response compared to the detection method as shown in FIG. Excellent effect.

In each of the above embodiments, the detection principle by the coil unit 2 and the magnetic response member 3 may be configured by a known phase shift type position detection principle. For example, in the coil unit 2 shown in FIG. 2, the relationship between the primary coil and the secondary coil is reversed, and the sine phase coil 21 and the minus sine phase coil 23 are connected to the sine signals sin ωt, − Excited by sin ωt, the cosine phase coil 22 and the minus cosine phase coil 24 are excited by cosine signals cos ωt and −cos ωt having opposite phases, and an electrical phase shift θ corresponding to the detection target inclination is generated from the coils 11 to 14. An output signal sin (ωt−θ) including this may be obtained.
As described above, the detection principle by the coil unit 2 and the magnetic response member 3 may be configured to obtain an analog detection output based on a known differential transformer type position detection principle.

Alternatively, in each of the above embodiments, the configuration of the coil unit 2 is not configured to include a pair of a primary coil and a secondary coil, but is configured by only one coil, and the one coil is a predetermined AC signal. The inclination detection data may be obtained by driving at a constant voltage and measuring a current change based on an inductance change that occurs according to the amount of penetration of the magnetic body (magnetic response member 3) into the coil. In that case, the required measurement is performed by a method of measuring the amplitude change of the output signal in response to the current change or a method of measuring the phase change between the output signals at each end of the coil in response to the current change. Can do.
In each of the above embodiments, the output format of the detection data is not limited to digital absolute data or analog voltage data, but may be an appropriate format according to the purpose of use, such as incremental pulse data or a repetitive pulse signal obtained by frequency conversion of an absolute value. As good as

Sectional drawing which shows one Example of the linear position detection apparatus which concerns on this invention. The circuit diagram which shows the structural example of the coil part in FIG. The figure which shows the example of a change of the magnetic response member in FIG. The external appearance schematic diagram which shows another Example of the linear position detection apparatus which concerns on this invention. 2 is a schematic diagram showing an example in which two linear position detection devices shown in FIG. 1 are combined in an orthogonal relationship to detect a tilt in two axes. The schematic diagram which shows another example which detects the inclination of a biaxial direction by combining two linear position detection apparatuses which concern on this invention by orthogonal relationship. The block diagram which shows an example of the measurement circuit of a phase detection type applicable to the linear position detection apparatus which concerns on this invention. The block diagram which shows another example of the measurement circuit of a phase detection type applicable to the linear position detection apparatus which concerns on this invention. Operation | movement explanatory drawing of FIG. The block diagram which shows the example of a change added to the circuit of FIG. The block diagram which shows another example of the measurement circuit of a phase detection type applicable to the linear position detection apparatus which concerns on this invention. Operation | movement explanatory drawing of FIG.

Explanation of symbols

10, 10X, 10Y, 10X ', 10Y' Linear position detector (inclinometer)
DESCRIPTION OF SYMBOLS 1 Storage body 1a Passage 2 Coil part 11-15 Primary coil 21-24 Secondary coil 3 Magnetic response member 40 Digital phase detection circuit

Claims (1)

Comprising a coil portion and the magnetic response members, displaced in a non-contact manner relative linear one of the before and Symbol coil unit magnetic response member relative to the other, an output signal corresponding thereto is obtained from the coil portion A linear position detecting device for detecting a relative linear position of the magnetic response member with respect to the coil portion ,
The coil unit includes first and second coil groups excited by a predetermined AC signal, and each coil group is provided in a different arrangement along a direction of relative linear displacement of the magnetic response member with respect to the coil unit. Thereby, the first coil group is arranged to generate an output AC signal indicating an amplitude function characteristic of a sine phase according to a relative linear position of the magnetic response member with respect to the coil portion, and The second coil group is arranged to generate an output AC signal indicating an amplitude function characteristic of a cosine phase according to a relative linear position of the magnetic response member with respect to the coil portion.
The first coil group is composed of two-pole coils arranged so as to exhibit magnetoresistive change characteristics of the sine phase and the minus sine phase with respect to the relative linear position of the magnetic response member, and the second coil group Is composed of two-pole coils arranged to exhibit magnetoresistive change characteristics of a cosine phase and a minus cosine phase, respectively, with respect to the relative linear position of the magnetic response member,
Furthermore, the coil of each pole in each coil group consists of only one coil excited by the AC signal, and an output showing an amplitude change based on an inductance change according to a relative linear position of the magnetic response member with respect to each coil. An AC voltage signal is extracted from each one coil, and based on this, the sine phase amplitude function is obtained by differentially synthesizing the outputs of the sine phase and the minus sine phase output from the two coils of the first coil group. An output AC signal indicating the characteristic is generated, and an output indicating the amplitude function characteristic of the cosine phase by differentially combining the outputs of the cosine phase and the minus cosine phase output from the two coils of the second coil group. A linear position detecting device that generates an AC signal.

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