JP3347705B2 - Phase shifters, attenuators and nonlinear signal generators - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a small phase shifter, an attenuator and a nonlinear signal generator whose input and output impedances are matched.

[0002]

2. Description of the Related Art With the rapid development of wireless multimedia communication in recent years, demands for further miniaturization and economy of wireless devices have been increasing. Monolithic Microwave Integrated Circ
uit: MMIC) is attracting attention as a fundamental technology for promoting the miniaturization and economicalization of wireless devices. The reason is, of course, that the MMIC itself is small,
This is because a chip having high uniformity can be manufactured without adjustment by a semiconductor process, so that mass productivity is high, and further, high integration and high precision reproducibility can reduce mounting cost and improve reliability.

As high-frequency functional circuits expected to be downsized by MMIC, there are known an amplifier for amplifying a high-frequency signal, an oscillator for generating a local oscillation signal, a frequency converter for performing frequency conversion, and the like. Furthermore, phase shifters that control the phase of high-frequency signals, attenuators that control the amplitude of high-frequency signals, and nonlinear circuits that generate nonlinear signals are used for application to antenna directivity control circuits and power amplifier distortion compensation circuits. Expectations are also being raised for miniaturization of MMICs for signal generators and the like.

[0004] A conventional phase shifter and attenuator will be described. FIG. 62 is a circuit diagram of a conventional phase shifter and attenuator. The phase shifter and the attenuator are reflection type phase shifters and attenuators using a 90-degree branch line hybrid. For example, "Miyauchi, Yamamoto:" Microwave circuit for communication ", IEICE, pp. 312-320 (Showa 5
6 / October) shows the basic operation principle of the phase shifter and the attenuator.

As shown in FIG. 62, the 90-degree branch line hybrid is composed of four high-frequency transmission lines 3a, 3b, 3c, and 3d having an electric length of 90 degrees at the frequency f ₀ . Each connection point of the high-frequency transmission lines 3a to 3d is connected to an input / output terminal 4a of a 90-degree branch line hybrid,
4b, 4c, and 4d. The input terminal 1 is connected to the input / output terminal 4a of the 90-degree branch line hybrid, and the output terminal 2 is connected to the input / output terminal 4b of the 90-degree branch line hybrid. Also, the input / output terminals 4c and 4d of the 90-degree branch line hybrid have:
The variable impedance elements 5a and 5b are respectively connected. Here, Z ₀ the input impedance of the input and output terminals 1 and 2, the high-frequency transmission line 3a, Z ₀ characteristic impedance of 3b, the high-frequency transmission line 3c, the characteristic impedance of 3d Z _0/2 ^1/2, variable impedance Element 5
a, the impedance of the 5b and Z _1.

Next, the operation of the conventional configuration shown in FIG. 62 will be described. The signal input from the input terminal 1 is distributed by the 90-degree branch line hybrid constituted by the high-frequency transmission lines 3a to 3d, and is output from the input / output terminals 4c and 4d of the 90-degree branch line hybrid. These input / output terminals 4c, 4d are connected to the variable impedance elements 5a,
5b. Therefore, part of the signal power is absorbed by the resistance component R ₁ of the impedance Z _1, the remaining signal is given a phase change by the reactance component X ₁ of the impedance Z _1, reflecting the input terminal 1 and output terminal 2 Is done.

In this case, the variable impedance elements 5a,
The impedance Z ₁ of 5b are identical, the variable impedance element 5a, signals reflected in the input terminal 1 from each 5b becomes equal amplitude and opposite phases is canceled. On the other hand, the signals reflected from the variable impedance elements 5a and 5b to the output terminal 2 are combined with the same amplitude and the same phase. Therefore, by changing the variable impedance element 5a, the impedance Z ₁ of 5b,
The configuration shown in FIG. 62 can be operated as a phase shifter or an attenuator while ensuring the matching of the input / output impedance at the frequency f ₀ .

The structure shown in FIG. 62 operates as a phase shifter.
If so, the impedance Z ₁Is virtually riaq
Tance component X₁Variable impedance to consist of
Elements 5a and 5b, and their reactance component X₁
May be changed continuously. X is the reactance component₁
From (X₁+ ΔX₁) When the phase shifter is changed
The phase change amount θ is given by the following equation. θ = -2tan^-1[(X₁+ ΔX₁) / Z₀] +2 tan^-1(X₁/ Z₀) [Rad] (1) The configuration shown in FIG. 62 is operated as an attenuator.
In the case, the impedance Z₁Is substantially the resistance component R₁
Variable impedance elements 5a, 5
b and its resistance component R₁If you change continuously
good. The attenuation L of the attenuator in this case is given by the following equation.
You. L = 20log_Ten| (Z₀+ R₁) / (Z₀-R₁) | [DB] (2)

FIG. 63 shows a conventional phase shifter shown in FIG.
FIG. 3 is a circuit diagram showing a specific example of the embodiment. In this figure, FIG.
The same parts as those shown in FIG.
Abbreviate. In the phase shifter shown in FIG.
The variable capacitors 11a, 1
1b are used respectively. High-frequency transmission lines 3a-
3d is assumed to be lossless, and the input / output impedance Z
₀= 50Ω, frequency f ₀= 5 GHz.

FIG. 64 shows an amplitude characteristic (forward transmission coefficient S _21).
FIG. ₁₁ is a diagram showing simulation results of the input reflection coefficient and the input reflection coefficient S ₁₁ ). The horizontal axis is frequency [GHz], the left vertical axis is forward transmission coefficient S ₂₁ [dB], and the right vertical axis is input reflection coefficient S _11.
[DB]. FIG. 65 is a diagram showing a simulation result of the phase characteristic (forward transmission coefficient S ₂₁ ).
The horizontal axis is frequency [GHz], and the vertical axis is forward transmission coefficient S.
₂₁ [deg. ]. Figure 64, In Figure 65, the variable capacitor 11a, the capacitance C ₁ of 11b 0.05 pF, 0.
The values were changed to 1 pF, 0.3 pF, 0.5 pF, and 0.7 pF. As can be seen from the figure, the frequency f = 4.5G
Hz to 5.4 GHz, an amplitude fluctuation of 0.5 d
B, the input reflection amount is −10 dB or less (FIG. 64),
A characteristic having a phase change amount of 60 degrees or more (FIG. 65) is obtained.

FIG. 66 is a circuit diagram showing a specific example of the conventional attenuator shown in FIG. In this figure, FIG.
The same parts as those shown in FIG. In the attenuator shown in FIG. 66, variable resistors 21a and 21b are used as variable impedance elements 5a and 5b, respectively. It is assumed that the high-frequency transmission line has no loss, and the input / output impedance Z ₀ = 50Ω and the frequency f ₀ = 5 GHz.

FIG. 67 is a diagram showing a simulation result of the amplitude characteristic (forward transmission coefficient S ₂₁ ). The horizontal axis is frequency [GHz], and the vertical axis is forward transfer coefficient S ₂₁ [dB].
It is. FIG. 68 is a diagram showing a simulation result of the amplitude characteristic (input reflection coefficient S ₁₁ ). The horizontal axis is frequency [G
Hz], and the vertical axis represents the input reflection coefficient S ₁₁ [dB]. FIG.
7, FIG. 68, the variable resistor 21a, the resistor R ₁ of 21b 0Ω, 10Ω, 20Ω, 30Ω , is varied with 50 [Omega. As can be seen from the figure, the frequency f = 4.5 GHz-
At 5.5 GHz, a characteristic having an attenuation of 14 dB or more (FIG. 67) and an input reflection of -14 dB or less (FIG. 68) are obtained.

Next, a conventional nonlinear signal generator will be described. FIG. 69 is a circuit diagram of a conventional nonlinear signal generator. This nonlinear signal generator is a nonlinear signal generator using a 90-degree branch line hybrid. For example, Japanese Patent Application Laid-Open No. 63-189004 discloses the basic operation principle of this nonlinear signal generator. In this figure, the same parts as those in FIG. The nonlinear signal generator shown in FIG. 69 has an electrical length of 90 at frequency f ₀ , as in FIG.
It has a 90-degree branch line hybrid composed of four high-frequency transmission lines 3a to 3d.

A non-linear element composed of diodes 31a and 32a, a terminating resistor 33a, DC blocking capacitors 34a and 35a, and a bias terminal 36 is connected to the input / output terminal 4c of the 90-degree branch line hybrid. Specifically, the input / output terminal 4c of the 90-degree branch line hybrid is connected to the anode of the diode 31a,
The cathode of the diode 32a and one end of the terminating resistor 33a are connected. The anode of the diode 32a and the other end of the terminating resistor 33a are grounded at high frequencies by DC blocking capacitors 35a and 34a, respectively. The cathode of the diode 31a is directly grounded. Then, the diode 32a and the capacitor 35a
Is connected to the bias terminal 36. Thus, the bias current from the bias terminal 36 flows through the diodes 31a and 32a.

Similarly, the input / output terminal 4d of the 90-degree branch line hybrid includes diodes 31b and 32b, a terminating resistor 33b, and DC blocking capacitors 34b and 35b.
And a non-linear element composed of the bias terminal 36 and the bias terminal 36 are connected. Specifically, the input / output terminal 4c of the 90-degree branch line hybrid is connected to the anode of the diode 31b, the cathode of the diode 32b,
b is connected to one end. The anode of the diode 32b and the other end of the terminating resistor 33b are grounded at high frequencies by DC blocking capacitors 35b and 34b, respectively. The cathode of the diode 31b is directly grounded. Then, the diode 32b and the capacitor 3
The bias terminal 36 is connected to the connection point 5b. Thus, the bias current from the bias terminal 36 flows through the diodes 31b and 32b.

Next, the operation of the conventional configuration shown in FIG. 69 will be described. The signal input from the input terminal 1 is distributed by the 90-degree branch line hybrid constituted by the high-frequency transmission lines 3a to 3d, and is output from the input / output terminals 4c and 4d of the 90-degree branch line hybrid. The signal output from the input / output terminal 4c is
The signal input to the terminal a and the terminal resistor 33a and output from the input / output terminal 4d are input to the diodes 31b and 32b and the terminal resistor 33b.

At this time, the value of the combined impedance of the diodes 31a and 32a and the terminating resistor 33a is made equal to the characteristic impedance Z _0, and
b, the value of the combined impedance between 32b and the terminating resistor 33b is to be equal to the characteristic impedance Z _0, it is assumed that the bias current from the bias terminal 36 is properly set. In this case, the linear signal component of the input signal is suppressed by the combined impedance and the diode 31
a, 32a and diodes 31b, 32b only output a non-linear signal corresponding to the input signal power at output terminal 2.
Output.

[0018]

However, in the conventional phase shifter, attenuator and non-linear signal generator using such a 90-degree branch-line hybrid, the frequency f _{0 is} required to form the 90-degree branch-line hybrid.
In this case, as many as four high-frequency transmission lines 3a to 3d having an electric length of 90 degrees are required, and the size becomes large. Therefore, when the conventional phase shifter, attenuator, and nonlinear signal generator are applied to, for example, an array antenna that requires a small number of elements and a nonlinear distortion compensation circuit of a power amplifier that requires a small size and weight, However, there is a problem that the size of the becomes large.

Therefore, an object of the present invention is to reduce the size of a phase shifter whose input and output impedances are matched. Another object of the present invention is to reduce the size of an attenuator whose input and output impedances are matched. Another object of the present invention is to
An object of the present invention is to reduce the size of a nonlinear signal generator whose input and output impedances are matched.

[0020]

In order to achieve such an object, a phase shifter according to the present invention has an impedance connected between an input terminal and an output terminal and substantially constituted by reactance. A first high-frequency impedance element, a first high-frequency phase shift element having one end connected to the input terminal, having a phase change amount of 90 degrees at frequency f ₀ and having an impedance conversion function, an output terminal, and a first high-frequency impedance element. A second high-frequency phase shift element connected between the other end of the phase-shift element and having an amount of phase change at a frequency f ₀ of 90 degrees and having an impedance conversion function;
A second high-frequency impedance element having one end connected to a common connection point with the high-frequency phase shift element and the other end grounded, and having an impedance substantially constituted by reactance. And the impedance of the second high-frequency impedance element are set such that the input / output reflection coefficient at the frequency f ₀ is substantially zero. In this case, the reactance of each of the first and second high-frequency impedance elements may be variable.

Each of the first and second high-frequency phase shift elements
The first configuration example has a frequency f₀Electrical length at 90 degrees high
It is a frequency transmission line. Also, the first and second high-frequency transfer
The second configuration example of each of the phase elements has a frequency f ₀Electricity in
A high-frequency transmission line whose length is smaller than 90 degrees and a high-frequency transmission line
One end is connected to both ends of the road and
Composed of two capacitors whose other ends are grounded.
It is a π-type circuit. Also, first and second high-frequency phase shifters
The third configuration example of each of the inductors is an inductor and a capacitor.
And a lumped constant circuit composed of

In the phase shifter using the first configuration example of the first and second high-frequency phase shift elements, the input impedance of the input terminal and the output impedance of the output terminal are Z ₀ , respectively, and the reactance of the first high-frequency impedance element. Is X ₁ , the characteristic impedance of the high-frequency transmission line used as the first and second high-frequency phase shift elements is Z ₂ ,
When the reactance of the high-frequency impedance element is X ₃ , the reactance X ₃ is set by the relational expression of X ₃ = Z ₂ ² X ₁ / (4Z ₀ ² ).

In the phase shifter using the second configuration example of the first and second high-frequency phase shift elements, the input impedance of the input terminal and the output impedance of the output terminal are Z ₀ , respectively, and the reactance of the first high-frequency impedance element. Is X ₁ , the electrical length and characteristic impedance of the high-frequency transmission line included in the first and second high-frequency phase shift elements are θ and Z, respectively, and the capacitance of the capacitor included in the first and second high-frequency phase shift elements is C. , When the reactance of the second high-frequency impedance element is X ₃ , the capacitance C and the reactance X ₃ are represented by the following relationship: C = 1 / (2πf ₀ Ztan θ) X ₃ = (Z sin θ) ² X ₁ / (4Z ₀ ² ) It is set by the formula.

A third configuration example of the first and second high-frequency phase shift elements is a T-type composed of a capacitor having one end grounded and two inductors each having one end connected to the other end of the capacitor. In the case of a circuit, in a phase shifter using the circuit, the input impedance of the input terminal and the output impedance of the output terminal are Z ₀ , the reactance of the first high-frequency impedance element is X ₁ , the capacitance of the capacitor is C, and the inductance of the inductor is C. When the inductance is L and the reactance of the second high-frequency impedance element is X ₃ , the capacitance C and the reactance X ₃ are C = 1 / [(2πf ₀ ) ² L] X ₃ = (2πf ₀ L) ² X ₁ / (4Z ₀ ² ) is set.

The third structure of the first and second high-frequency phase shift elements
An example is the inductor and the two ends of the inductor.
One end is connected and the other end is grounded.
In the case of a π-type circuit composed of two capacitors,
In a phase shifter using this, the input impedance of the input terminal is
Output impedance of the
Z ₀, The reactance of the first high-frequency impedance element
Is X₁, The capacitance of the capacitor is C, the inductance of the inductor
L is the reactance of the second high-frequency impedance element.
Close is X_Three, The capacitance C and the reactance X
_ThreeIs C = 1 / [(2πf₀)^TwoL] X_Three = (2πf₀L)^TwoX₁/ (4Z₀ ^Two) Is set by the relational expression.

A third configuration example of the first and second high-frequency phase shift elements is a T-type composed of an inductor having one end grounded and two capacitors each having one end connected to the other end of the inductor. In the case of a circuit, in a phase shifter using the circuit, the input impedance of the input terminal and the output impedance of the output terminal are Z ₀ , the reactance of the first high-frequency impedance element is X ₁ , the capacitance of the capacitor is C, and the inductance of the inductor is C. When the inductance is L and the reactance of the second high-frequency impedance element is X ₃ , the capacitance C and the reactance X ₃ are C = 1 / [(2πf ₀ ) ² L] X ₃ = (2πf ₀ L) ² X ₁ / (4Z ₀ ² ) is set.

The third structure of the first and second high-frequency phase shift elements
An example is a capacitor and both ends of the capacitor.
One end is connected and the other end is grounded.
In the case of a π-type circuit composed of two inductors,
In a phase shifter using this, the input impedance of the input terminal is
Output impedance of the
Z ₀, The reactance of the first high-frequency impedance element
Is X₁, The capacitance of the capacitor is C, the inductance of the inductor
L is the reactance of the second high-frequency impedance element.
Close is X_Three, The capacitance C and the reactance X
_ThreeIs C = 1 / [(2πf₀)^TwoL] X_Three = (2πf₀L)^TwoX₁/ (4Z₀ ^Two) Is set by the relational expression.

One configuration example of each of the first and second high-frequency impedance elements is a variable capacitor. Another configuration example of each of the first and second high-frequency impedance elements is a resonance circuit. The first configuration example of the resonance circuit is a series resonance circuit in which an inductor and a capacitor are connected in series. The second configuration example of the resonance circuit is a parallel resonance circuit in which an inductor and a capacitor are connected in parallel. The third configuration example of the resonance circuit is a composite resonance circuit in which a second capacitor is connected in parallel to a series resonance circuit in which an inductor and a first capacitor are connected in series. A fourth configuration example of the resonance circuit is a composite resonance circuit in which two series resonance circuits in which an inductor and a capacitor are connected in series are connected in parallel.

[0029]

[0030]

[0031]

In order to achieve the above object, the present invention
Is the light attenuator connected between the input and output terminals?
Have an impedance composed substantially of resistors
One end is connected to the first high-frequency impedance element and the input terminal.
Connected and the phase change at frequency f ₀ is 90 degrees
First high-frequency phase shift element having impedance conversion function
Between the output terminal and the other end of the first high-frequency phase shifter.
And the phase change at frequency f ₀ is 90 degrees,
Second high-frequency phase shifter having impedance conversion function
Between the first high-frequency phase shift element and the second high-frequency phase shift element
Whether one end is connected to the common connection point and the other end is grounded
Have an impedance composed substantially of resistors
A second high-frequency impedance element;
Each of the second high-frequency phase-shifting elements has a voltage at the frequency f _0.
High-frequency transmission line with airiness less than 90 degrees and high-frequency transmission
Each end is connected to both ends of the line and each
And two other capacitors grounded at the other end.
That a π-type circuit, each output impedance of the input impedance and the output terminal of the input terminal Z _0, first
Resistance of the high-frequency impedance element R _1, included in the first and second electrical length and a characteristic impedance of each θ and Z of the high-frequency transmission line included in the high-frequency phase shifting elements, the first and second high-frequency phase shifting elements of the The capacitance of the capacitor to be used is C, and the resistance of the second
_When it is ₃ , the capacitance C and the resistance R ₃ are set by the relational expression of C = 1 / (2πf ₀ Ztan θ) R ₃ = (Z sin θ) ² R ₁ / (4Z ₀ ² ).

Also, the attenuator of the present invention has an input terminal and an output terminal.
Connected between the terminals and consisting essentially of a resistor
First high frequency impedance element having impedance
Child and one end connected to the input terminal and position of the frequency f ₀
Phase change is 90 degrees and has impedance conversion function
A first high-frequency phase shifter, an output terminal and a first high-frequency shifter.
It is connected between the other end of the phase element and the phase at the frequency f ₀
The amount of change is 90 degrees and has an impedance conversion function
A second high-frequency phase shift element, a first high-frequency phase shift element, and a second
When one end is connected to the common connection point with the high-frequency phase shift element
The other end is grounded at its other end and substantially comprises a resistor.
Second high-frequency impedance element having impedance
And each of the first and second high-frequency phase shift elements
Includes a capacitor whose one end is grounded, Ri T-type circuit der composed of two inductors each one end connected to the other end of the capacitor, the output impedance of the input impedance and the output terminal of the input terminals, respectively Z
₀ , when the resistance of the first high-frequency impedance element is R ₁ , the capacitance of the capacitor is C, the inductance of the inductor is L, and the resistance of the second high-frequency impedance element is R ₃ , the capacitance C and the resistance R ₃ are C = 1 / [(2πf ₀ ) ² L] R ₃ = (2πf ₀ L) ² R ₁ / (4Z ₀ ² )

The attenuator of the present invention has an input terminal and an output terminal.
Connected between the terminals and consisting essentially of a resistor
First high frequency impedance element having impedance
Child and one end connected to the input terminal and position of the frequency f ₀
Phase change is 90 degrees and has impedance conversion function
A first high-frequency phase shifter, an output terminal and a first high-frequency shifter.
It is connected between the other end of the phase element and the phase at the frequency f ₀
The amount of change is 90 degrees and has an impedance conversion function
A second high-frequency phase shift element, a first high-frequency phase shift element, and a second
When one end is connected to the common connection point with the high-frequency phase shift element
The other end is grounded at its other end and substantially comprises a resistor.
Second high-frequency impedance element having impedance
And each of the first and second high-frequency phase shift elements
The inductor and, Ri π type circuit der consists respective other end with two capacitors which are grounded with each one at both ends of the inductor is connected, the output of the input impedance and the output terminal of the input terminal When the impedance is Z ₀ , the resistance of the first high-frequency impedance element is R ₁ , the capacitance of the capacitor is C, the inductance of the inductor is L, and the resistance of the second high-frequency impedance element is R ₃ , the capacitance C and the resistance R ₃ is set by the relational expression of C = 1 / [(2πf ₀ ) ² L] R ₃ = (2πf ₀ L) ² R ₁ / (4Z ₀ ² ).

The attenuator of the present invention has an input terminal and an output terminal.
Connected between the terminals and consisting essentially of a resistor
First high frequency impedance element having impedance
Child and one end connected to the input terminal and position of the frequency f ₀
Phase change is 90 degrees and has impedance conversion function
A first high-frequency phase shifter, an output terminal and a first high-frequency shifter.
It is connected between the other end of the phase element and the phase at the frequency f ₀
The amount of change is 90 degrees and has an impedance conversion function
A second high-frequency phase shift element, a first high-frequency phase shift element, and a second
When one end is connected to the common connection point with the high-frequency phase shift element
The other end is grounded at its other end and substantially comprises a resistor.
Second high-frequency impedance element having impedance
And each of the first and second high-frequency phase shift elements
Includes an inductor whose one end is grounded, one end of each of the T-type circuit der composed of two capacitors connected to the other end of the inductor is, the output impedance of the input impedance and the output terminal of the input terminals, respectively Z
₀ , when the resistance of the first high-frequency impedance element is R ₁ , the capacitance of the capacitor is C, the inductance of the inductor is L, and the resistance of the second high-frequency impedance element is R ₃ , the capacitance C and the resistance R ₃ are C = 1 / [(2πf ₀ ) ² L] R ₃ = (2πf ₀ L) ² R ₁ / (4Z ₀ ² )

The attenuator of the present invention has an input terminal and an output terminal.
Connected between the terminals and consisting essentially of a resistor
First high frequency impedance element having impedance
Child and one end connected to the input terminal and position of the frequency f ₀
Phase change is 90 degrees and has impedance conversion function
A first high-frequency phase shifter, an output terminal and a first high-frequency shifter.
It is connected between the other end of the phase element and the phase at the frequency f ₀
The amount of change is 90 degrees and has an impedance conversion function
A second high-frequency phase shift element, a first high-frequency phase shift element, and a second
When one end is connected to the common connection point with the high-frequency phase shift element
The other end is grounded at its other end and substantially comprises a resistor.
Second high-frequency impedance element having impedance
And each of the first and second high-frequency phase shift elements
A capacitor and, Ri π type circuit der consists respective other end with two inductors is grounded with each one at both ends of the capacitor is connected, the output of the input impedance and the output terminal of the input terminal When the impedance is Z ₀ , the resistance of the first high-frequency impedance element is R ₁ , the capacitance of the capacitor is C, the inductance of the inductor is L, and the resistance of the second high-frequency impedance element is R ₃ , the capacitance C and the resistance R ₃ is set by the relational expression of C = 1 / [(2πf ₀ ) ² L] R ₃ = (2πf ₀ L) ² R ₁ / (4Z ₀ ² ). Also, the attenuator described above
, The first and second high-frequency impedance elements
May be variable.

In order to achieve the above object, a non-linear signal generator according to the present invention is connected between an input terminal and an output terminal and has an impedance including a resistance component. Generating a nonlinear signal in response to the first
A first high-frequency phase-shift element having one end connected to the input terminal, a phase change amount at a frequency f ₀ of 90 degrees, and an impedance conversion function,
And the other end of the high-frequency phase-shifting element of
One end is connected to a second high-frequency phase shift element having a phase change amount of 90 degrees at ₀ and having an impedance conversion function, and a common connection point between the first high-frequency phase shift element and the second high-frequency phase shift element. The other end is grounded, has an impedance including a resistance component, and has a first
And a second non-linear element that generates a non-linear signal similar to the non-linear element described above. The resistance component of the impedance of the first non-linear element and the resistance component of the impedance of the second non-linear element are input and output at the frequency f _0. The reflection coefficient is set to be approximately zero.

A first configuration example of each of the first and second high-frequency phase shift elements is a high-frequency transmission line having an electrical length of 90 degrees at the frequency f ₀ . Each of the second configuration example of the first and second high-frequency phase shifting elements, the electrical length at the frequency f ₀ 9
This is a π-type circuit composed of a high-frequency transmission line smaller than 0 degrees and two capacitors each having one end connected to both ends of the high-frequency transmission line and the other end grounded. A third configuration example of each of the first and second high-frequency phase shift elements is a lumped constant circuit including an inductor and a capacitor.

In the nonlinear signal generator using the first configuration example of the first and second high-frequency phase shift elements, the input impedance of the input terminal and the output impedance of the output terminal are Z ₀ , respectively, and the resistance of the first nonlinear element is The component is R ₁ , the first
When the characteristic impedance of the high-frequency transmission line used as the second high-frequency phase shift element is Z ₂ , and the resistance component of the second nonlinear element is R ₃ , the resistance components R ₁ and R ₃ are R ₁ = 2Z ₀ R ₃ = Z ₂ ² R ₁ / (4Z ₀ ² )

In the nonlinear signal generator using the second configuration example of the first and second high-frequency phase shifting elements, the input impedance of the input terminal and the output impedance of the output terminal are Z ₀ , respectively, and the resistance of the first nonlinear element is The component is R ₁ , the first
The electrical length and characteristic impedance of the high-frequency transmission line included in the first and second high-frequency phase shift elements are θ and Z, the capacitance of the capacitor included in the first and second high-frequency phase shift elements is C, and the second nonlinear element. Is R ₃ , the capacitance C and the resistance components R ₁ and R ₃ are C = 1 / (2πf ₀ Ztan θ) R ₁ = 2Z ₀ R ₃ = (Z sin θ) ² R ₁ / (4Z ₀ ² ).

A third configuration example of the first and second high-frequency phase shift elements is a T-type composed of a capacitor having one end grounded and two inductors each having one end connected to the other end of the capacitor. In the case of a circuit, in a nonlinear signal generator using this, the input impedance of the input terminal and the output impedance of the output terminal are Z ₀ and the first impedance, respectively.
When the resistance component of the non-linear element is R ₁ , the capacitance of the capacitor is C, the inductance of the inductor is L, and the resistance component of the second non-linear element is R ₃ , the capacitance C and the resistance components R ₁ and R ₃ are C = 1 / [(2πf ₀ ) ² L] R ₁ = 2Z ₀ R ₃ = (2πf ₀ L) ² R ₁ / (4Z ₀ ² )

A third configuration example of the first and second high-frequency phase-shifting elements comprises an inductor and one end connected to both ends of the inductor and the other end grounded.
In the case of a π-type circuit composed of two capacitors,
In the nonlinear signal generator using this, the input impedance of the input terminal and the output impedance of the output terminal are Z ₀ , the resistance component of the first nonlinear element is R ₁ , the capacitance of the capacitor is C, the inductance of the inductor is L,
When the resistance component of the second nonlinear element is R ₃ , the capacitance C
And the resistance components R ₁ and R ₃ are set by the relational expression of C = 1 / [(2πf ₀ ) ² L] R ₁ = 2Z ₀ R ₃ = (2πf ₀ L) ² R ₁ / (4Z ₀ ² ) ing.

A third configuration example of the first and second high-frequency phase shift elements is a T-type composed of an inductor having one end grounded and two capacitors each having one end connected to the other end of the inductor. In the case of a circuit, in a nonlinear signal generator using this, the input impedance of the input terminal and the output impedance of the output terminal are Z ₀ and the first impedance, respectively.
When the resistance component of the non-linear element is R ₁ , the capacitance of the capacitor is C, the inductance of the inductor is L, and the resistance component of the second non-linear element is R ₃ , the capacitance C and the resistance components R ₁ and R ₃ are C = 1 / [(2πf ₀ ) ² L] R ₁ = 2Z ₀ R ₃ = (2πf ₀ L) ² R ₁ / (4Z ₀ ² )

A third configuration example of the first and second high-frequency phase shift elements is a capacitor having a capacitor and one end connected to both ends of the capacitor and the other end grounded.
In the case of a π-type circuit composed of two inductors,
In the nonlinear signal generator using this, the input impedance of the input terminal and the output impedance of the output terminal are Z ₀ , the resistance component of the first nonlinear element is R ₁ , the capacitance of the capacitor is C, the inductance of the inductor is L,
When the resistance component of the second nonlinear element is R ₃ , the capacitance C
And the resistance components R ₁ and R ₃ are set by the relational expression of C = 1 / [(2πf ₀ ) ² L] R ₁ = 2Z ₀ R ₃ = (2πf ₀ L) ² R ₁ / (4Z ₀ ² ) ing.

One configuration example of each of the first and second nonlinear elements has two diodes connected in parallel with opposite polarities and a resistor connected in parallel with these diodes, and each of the diodes has a bias. It is configured to allow a current to flow.

[0046]

DETAILED DESCRIPTION OF THE INVENTION The present invention uses two high-frequency phase-shifting element having an impedance conversion function is a phase change amount 90 degrees at the frequency f _0, the high-frequency circuit which attained matching of input and output impedance Is the most important feature. For example, when a high-frequency transmission line having an electrical length of 90 degrees at the frequency f ₀ is used as the high-frequency phase shift element, a high-frequency circuit using a conventional 90-degree branch line hybrid requiring four similar high-frequency transmission lines is used. , The number of required high-frequency transmission lines is halved. Therefore, by using the present invention, a phase shifter,
The attenuator and the nonlinear signal generator can be downsized. Less than,
Embodiments of the present invention will be described in detail with reference to the drawings.

[First Embodiment: Phase Shifter] Embodiment using a variable reactance element as a high-frequency impedance element: FIG. 1 is a circuit diagram showing a configuration of a phase shifter according to the present invention. Input terminal 101 and output terminal 102
Is connected to a variable reactance element (first high-frequency impedance element) 170a. The impedance of the variable reactance element 170a is substantially constituted by reactance. This reactance and X _1. Reactance X ₁ is variable. The input impedance of the input / output terminal 101 and the output impedance of the output terminal 102 are _denoted by Z0.

The input terminal 101 has one end of the first high-frequency phase shift element 103a (the input / output terminal 104 of the high-frequency phase shift element).
a) is connected. One end of a second high-frequency phase shift element 103b (input / output terminal 104b of the high-frequency phase shift element) is connected to the output terminal 102. High frequency phase shift element 103
The other end of each of a and 103b (input / output terminal 104c of the high-frequency phase shift element) is commonly connected. Both the high-frequency phase shift elements 103a and 103b have a phase change amount of 90 degrees at the frequency f ₀ and have an impedance conversion function. Frequency phase shifting elements 103a, an equivalent characteristic impedance when replacing the 103b in the high-frequency transmission line and Z _2.

One end of a variable reactance element (second high-frequency impedance element) 170b is connected to the input / output terminal 104c of the high-frequency phase shift element. The other end of variable reactance element 170b is grounded. The impedance of the variable reactance element 170b is substantially constituted by reactance. This reactance and X _3. Reactance X ₃ is variable. The above-described impedance conversion function of the high-frequency phase shift elements 103a and 103b means that when the impedance of the variable reactance element 170b is converted and the converted impedance of the variable reactance element 170b is combined with the impedance of the variable reactance element 170a, This is a function to make the input / output reflection coefficient viewed from the input / output terminals 104a and 104b of the phase element substantially zero to achieve matching.

Next, the operation of the phase shifter shown in FIG. 1 will be described. The signal input from the input terminal 101 is divided into a first path passing through the variable reactance element 170a and a second path passing through the high-frequency phase shift element 103a, the variable reactance element 170b, and the high-frequency phase shift element 103b. You. Signal passing through the first journey, depending reactance X ₁ of the variable reactance elements 170a, predetermined phase variation is given. When the frequency of the signal passing through the second path is f ₀ , a phase change of 90 degrees is given to each of the signals by the high-frequency phase shift elements 103a and 103b, and the reactance X ₃ of the variable reactance element 170b causes A predetermined phase change is provided.

At this time, the reactances of the variable reactance elements 170a and 170b are combined so that the signals passing through the respective paths are combined at the input / output terminal 104b of the high-frequency phase shift element and output from the output terminal 102 while maintaining the same amplitude. X ₁ and X ₃ are set. By changing the reactances X ₁ and X ₃ of the variable reactance elements 170a and 170b set in this way simultaneously and continuously, the phase change amount of the phase shifter shown in FIG. 1 is continuously changed. Can be. Here, the input reflection coefficient S ₁₁ and the output reflection coefficient S ₂₂ of the phase shifter shown in Figure 1,

[0052]

(Equation 1)

Can be expressed by Therefore, the reactance X
_{3 ^{X 3 = Z 2 2 X 1}} / (4Z 0 2) By setting the relation of (4), since the input and output reflection coefficients S _11, S ₂₂ at the frequency f ₀ becomes zero , The input / output impedance at the frequency f ₀ can be matched. In addition,
When actually producing a phase shifter, the input / output reflection coefficients S ₁₁ and S ₂₂ at the frequency f ₀ need not be strictly zero, but a sufficient effect can be obtained if they are substantially zero.

[0054] At this time, the forward transmission coefficient S ₂₁ and the reverse transmission coefficient S ₁₂ of the phase shifter shown in FIG. _{1, S 21 = S 12 = (} 2Z 0 -X 1) / (2Z 0 + X ₁ ) ... (5) Also, the variable reactance element 170a,
The phase shift amount θ of the phase shifter when the reactances X ₁ and X ₃ of 170b are changed from X ₁ to (X ₁ + ΔX ₁ ) while maintaining the relationship of Expression (4) is given by the following expression. θ = −2 tan ⁻¹ [(X ₁ + ΔX ₁ ) / (2Z ₀ )] + 2 tan ⁻¹ [X ₁ / (2Z ₀ )] [rad] (6)

High-frequency phase shift element 103 having a phase change amount of 90 degrees at frequency f ₀ and having an impedance conversion function
a, 103b is, (a) the electrical length at the frequency f ₀ is 90 degrees of the high-frequency transmission line (see FIG. 2), a small high-frequency transmission line and its ends than the electrical length of 90 degrees at (b) frequency f ₀ A π-type circuit composed of two capacitors having one end connected and the other end grounded (see FIG. 3), (c) a lumped constant circuit composed of an inductor and a capacitor (see FIGS. 4 to 7), etc. It is configured using In the above configuration example,
The phase shifter can be miniaturized in the order of (a)>(b)> (c).
Hereinafter, a configuration example of a phase shifter using various high-frequency phase shift elements 103a and 103b will be described.

(First Configuration Example) FIG. 2 is a circuit diagram showing a first configuration example of the phase shifter shown in FIG. In this figure, the same parts as those in FIG. In the first configuration example, high-frequency transmission lines 113a and 113b whose electric length at a frequency f ₀ is 90 degrees are used as high-frequency phase shift elements 103a and 103b having an impedance conversion function, respectively. Note that the input / output terminals 114a, 114b,
Reference numeral 114c denotes input / output terminals 104a to 104a of the high-frequency phase shift element.
04c.

Characteristics of high-frequency transmission lines 113a and 113b
Impedance is Z_TwoThen, the input reflection of this phase shifter
Coefficient S₁₁And output reflection coefficient S_{twenty two}Is similar to equation (3)
Can be expressed. Therefore, the variable reactance element 170
a, the reactance X of 170b ₁, X_ThreeIs shown in equation (4).
The frequency f₀Enter at
Since the output reflection coefficient becomes zero, the frequency f₀Input / output
BR> Impedance can be matched. This and
And the forward transfer coefficient S of this phase shifter_{twenty one}And reverse transmission
Coefficient S₁₂Can be expressed in the same way as Expression (5).

(Second Configuration Example) FIG. 3 is a circuit diagram showing a second configuration example of the phase shifter shown in FIG. In this figure, the same parts as those in FIG. In the second configuration example, π-type circuits in which both ends of a high-frequency transmission line are grounded via capacitors are used as high-frequency phase shift elements 103a and 103b having an impedance conversion function.

The high-frequency transmission lines 123a and 123b have an electrical length θ at a frequency f ₀ of less than 90 degrees. One end of a high-frequency transmission line 123a is connected to one end of a capacitor 126a, and the other end of the high-frequency transmission line 123a is connected to one end of a capacitor 126b. Similarly, one end of a capacitor 126d is connected to one end of the high-frequency transmission line 123b, and one end of a capacitor 126c is connected to the other end of the high-frequency transmission line 123b. Capacitor 126a
To 126d are grounded. The above high-frequency transmission line 123a and capacitors 126a, 126
b constitutes one π-type circuit, and the high-frequency transmission line 1
23b and the capacitors 126c and 126d from the other π
A pattern circuit is configured. The input / output terminal 12 of the π-type circuit
Reference numerals 4a, 124b, and 124c correspond to the input / output terminals 104a to 104c of the high-frequency phase shift element.

The characteristic impedance of the high-frequency transmission lines 123a and 123b is Z, and the capacitance of the capacitors 126a to 126d is C. When this capacitance C is set to C = 1 / (2πf ₀ Ztanθ) (7), the input reflection coefficient S ₁₁ and the output reflection coefficient S ₂₂ of this phase shifter are

[0061]

(Equation 2)

Can be expressed by Therefore, by setting the reactance X ₃ of the variable reactance element 170b by the relational expression of X ₃ = (Z sin θ) ² X ₁ / (4Z ₀ ² ) (9), the input / output reflection at the frequency f ₀ is obtained. Since the coefficient becomes zero, it is possible to match the input / output impedance at the frequency f ₀ . At this time, the forward transfer coefficient S ₂₁ and the backward transfer coefficient S
₁₂ can be expressed in the same way as Expression (5). It should be noted that this second configuration example includes a capacitor 126b and a capacitor 126c.
Are provided individually. However, the capacitors 126b, 1
26c is connected to the input / output terminal 124c, so that it is possible to collectively replace the capacitors 26c with one capacitor having a capacity of 2C.

(Third Configuration Example) FIG. 4 is a circuit diagram showing a third configuration example of the phase shifter shown in FIG. In this figure, the same parts as those in FIG. In the third configuration example, a T-type circuit in which a connection point between two inductors is grounded via a capacitor is used as each of the high-frequency phase shift elements 103a and 103b having an impedance conversion function. One end of the capacitor 136a is grounded, and the other end is connected to a connection point between the inductors 133a and 133b. Also,
One end of the capacitor 136b is grounded, and the other end is connected to a connection point between the inductors 133c and 133d. The above capacitor 136a and inductors 133a, 133
b to form one T-type circuit, and the capacitor 136
The other T-type circuit is constituted by b and the inductors 133c and 133d. The input / output terminal 134 of the T-type circuit
a, 134b and 134c correspond to the input / output terminals 104a to 104c of the high-frequency phase shift element.

The inductance of the inductors 133a to 133d is L, and the capacitance of the capacitors 136a and 136b is C.
And When this capacitance C is set to C = 1 / [(2πf ₀ ) ² L] (10), the input reflection coefficient S ₁₁ and the output reflection coefficient S ₂₂ of this phase shifter are

[0065]

(Equation 3)

Can be expressed by Therefore, by setting the reactance X ₃ of the variable reactance element 170b by the relational expression of X ₃ = (2πf ₀ L) ² X ₁ / (4Z ₀ ² ) (12), the input at the frequency f ₀ is obtained. Since the output reflection coefficient becomes zero, the input / output impedance at the frequency f ₀ can be matched. At this time, the forward transfer coefficient S ₂₁ and the backward transfer coefficient S
₁₂ can be expressed in the same way as Expression (5).

(Fourth Configuration Example) FIG. 5 is a circuit diagram showing a fourth configuration example of the phase shifter shown in FIG. In this figure, the same parts as those in FIG. In the fourth configuration example, π-type circuits in which both ends of an inductor are grounded via capacitors are used as the high-frequency phase shift elements 103a and 103b having an impedance conversion function.

One end of inductor 143a is connected to one end of capacitor 146a, and the other end of inductor 143a is connected to one end of capacitor 146b. Similarly, a capacitor 146d is connected to one end of the inductor 143b.
Is connected to one end of the inductor 143b, and one end of the capacitor 146c is connected to the other end of the inductor 143b. Capacitor 1
The other ends of the respective 46a to 146d are grounded.
The above inductor 143a and capacitors 146a, 14
6b constitutes one π-type circuit, and the inductor 14
3b and the capacitors 146c and 146d form the other π-type circuit. The input / output terminal 144 of the π-type circuit
a, 144b, and 144c correspond to the input / output terminals 104a to 104c of the high-frequency phase shift element.

The inductance of inductors 143a and 143b is L, and the capacitance of capacitors 146a to 146d is C.
And When this capacitance C is set as in equation (10),
Input reflection coefficient of the phase shifter S ₁₁ and the output reflection coefficient S ₂₂
Can be expressed in the same manner as Expression (11). For this reason, the reactance X _{1 of} the variable reactance elements 170a and 170b,
The X ₃ by setting the relationship as shown in equation (12), since the input and output reflection coefficient at the frequency f ₀ becomes zero, it can be matched input and output impedances at a frequency f _0. At this time, the forward transmission coefficient S ₂₁ and the reverse transmission coefficient S ₁₂ of the phase shifter may be as for formula (5) representation. Note that the fourth configuration example includes the capacitors 146b and 146c individually. However, both the capacitors 146b and 146c
4C, it is possible to collectively replace the capacitors with one capacitor having a capacity of 2C.

(Fifth Configuration Example) FIG. 6 is a circuit diagram showing a fifth configuration example of the phase shifter shown in FIG. In this figure, the same parts as those in FIG. The fifth configuration example uses a T-type circuit in which a connection point between two capacitors is grounded via an inductor as the high-frequency phase shift elements 103a and 103b having an impedance conversion function. One end of the inductor 153a is grounded, and the other end is connected to a connection point between the capacitors 156a and 156b. Also,
One end of the inductor 153b is grounded, and the other end is connected to a connection point between the capacitors 156c and 156d. The above inductor 153a and capacitors 156a, 156
b to form one T-type circuit, and the inductor 153
b and capacitors 156c and 156d form the other T-type circuit. The input / output terminal 154 of the T-type circuit
a, 154b, and 154c correspond to the input / output terminals 104a to 104c of the high-frequency phase shift element.

The inductance of inductors 153a and 153b is L, and the capacitance of capacitors 156a to 156d is C.
And When this capacitance C is set as in equation (10),
Input reflection coefficient of the phase shifter S ₁₁ and the output reflection coefficient S ₂₂
Can be expressed in the same manner as Expression (11). For this reason, the reactance X _{1 of} the variable reactance elements 170a and 170b,
The X ₃ by setting the relationship as shown in equation (12), since the input and output reflection coefficient at the frequency f ₀ becomes zero, it can be matched input and output impedances at a frequency f _0. At this time, the forward transmission coefficient S ₂₁ and the reverse transmission coefficient S ₁₂ of the phase shifter may be as for formula (5) representation.

(Sixth Configuration Example) FIG. 7 is a circuit diagram showing a sixth configuration example of the phase shifter shown in FIG. In this figure, the same parts as those in FIG. In the sixth configuration example, π-type circuits in which both ends of a capacitor are grounded via an inductor are used as the high-frequency phase shift elements 103a and 103b having an impedance conversion function.

One end of the capacitor 166a is connected to one end of the inductor 163a, and the other end of the capacitor 166a is connected to one end of the inductor 163b. Similarly, one end of the capacitor 166b is connected to the inductor 163d.
Is connected to one end of the capacitor 166b, and one end of the inductor 163c is connected to the other end of the capacitor 166b. Inductor 1
The other end of each of 63a to 163d is grounded.
The above inductors 163a, 163b and capacitor 16
6a constitutes one π-type circuit, and the inductor 16
The other π-type circuit is composed of 3c and 163d and the capacitor 166b. The input / output terminal 164 of the π-type circuit
a, 164b and 164c correspond to the input / output terminals 104a to 104c of the high-frequency phase shift element.

The inductance of the inductors 163a to 163d is L, and the capacitance of the capacitors 166a and 166b is C.
And When this capacitance C is set as in equation (10),
Input reflection coefficient of the phase shifter S ₁₁ and the output reflection coefficient S ₂₂
Can be expressed in the same manner as Expression (11). For this reason, the reactance X _{1 of} the variable reactance elements 170a and 170b,
The X ₃ by setting the relationship as shown in equation (12), since the input and output reflection coefficient at the frequency f ₀ becomes zero, it can be matched input and output impedances at a frequency f _0. At this time, the forward transmission coefficient S ₂₁ and the reverse transmission coefficient S ₁₂ of the phase shifter may be as for formula (5) representation. The sixth configuration example includes the inductor 163b and the inductor 163c individually. However, both the inductors 163b and 163c
4c, it is possible to collectively replace the inductor with one inductor having an inductance of L / 2.

(Specific Example of Phase Shifter and Its Characteristics) Next, FIG.
2 shows a specific example of the phase shifter shown in FIG. 1, and shows a simulation result of amplitude characteristics and phase characteristics of the specific example. FIG. 8 is a diagram in which the first configuration example of the phase shifter shown in FIG. 2 is applied to an actual circuit. In this figure, the same parts as those in FIGS. 1 and 2 are denoted by the same reference numerals, and the description thereof will be appropriately omitted. In the phase shifter shown in FIG. 8, variable capacitors 171a, 171 are used as variable reactance elements 170a, 170b.
71b are each used. High frequency transmission line 11
The electrical length at frequency f ₀ = 5 GHz of 3a, 113b is 9
0 degrees. Further, the high-frequency transmission lines 113a and 113b
Is assumed to be lossless, and the input / output impedance Z ₀ =
It is assumed to be 50Ω.

FIG. 9 shows high-frequency transmission lines 113a and 113
b is a diagram showing simulation results of the amplitude characteristic in the case of the characteristic impedance Z ₂ = 70.7Ω the (forward transfer coefficient S ₂₁ and input reflection coefficient S _11). The horizontal axis is frequency [GH
z], the left vertical axis represents the forward transfer coefficient S ₂₁ [dB], and the right vertical axis represents the input reflection coefficient S ₁₁ [dB]. In addition, FIGS. 11, 14, 17, 20, 23, 26, 30, 3 to be described later.
2, 35, 37, 40, 42, 45, and 47 are also amplitude characteristic diagrams, and the horizontal axis and the vertical axis are the same as those in FIG. FIG. 10 is a diagram showing a simulation result of a phase characteristic (forward transfer coefficient S ₂₁ ) when the characteristic impedance Z _{2 of the} high-frequency transmission lines 113a and 113b is 70.7Ω. The horizontal axis is frequency [GHz], and the vertical axis is forward transfer coefficient S ₂₁ [deg. ]. In addition, FIG.
15, 18, 21, 24, 27, 31, 33, 36, 3
8, 41, 43, 46, and 48 are also phase characteristic diagrams, and the horizontal axis and the vertical axis are the same as those in FIG.

[0077] Figure 9, 10, equation (4), the variable capacitance C ₃ of the variable capacitor 171b capacitor 171a
Of to twice the volume C _1, 0.05 pF capacitance C ₁ of the variable capacitor 171a, 0.1pF, 0.15pF,
The values were changed to 0.2 pF, 0.3 pF, 0.4 pF, and 0.5 pF. As can be seen from the figure, the frequency f = 4.
In the range of 0 GHz to 6.0 GHz, the amplitude fluctuation is set to 0.1 GHz.
The characteristics are obtained within 5 dB, the input reflection amount is -12 dB or less (FIG. 9), and the phase change amount is 80 degrees or more (FIG. 10).

Similarly, FIGS. 11 and 12 show high-frequency transmission lines.
Characteristic impedance Z of roads 113a and 113b_Two= 50
Ω amplitude characteristics (forward transmission coefficient S_{twenty one}And input reflector
Number S ₁₁) And phase characteristics (forward transmission coefficient S)_{twenty one}) Stain
FIG. 9 is a diagram showing a simulation result. In FIG. 11 and FIG.
From the equation (4), the capacitance C of the variable capacitor 171b is_Three
Is the capacitance C of the variable capacitor 171a.₁Set to 4 times
The capacitance C of the variable capacitor 171a₁Is 0.05 pF,
0.1 pF, 0.15 pF, 0.2 pF, 0.3 pF,
The values are changed to 0.4 pF and 0.5 pF. Minute from the figure
Thus, the frequency f = 2.4 GHz to 5.7 GHz
Here, the amplitude fluctuation is within 0.5 dB, and the input reflection amount
-15 dB or less (FIG. 11), and the phase change amount is 60
A characteristic higher than that (FIG. 12) is obtained.

FIG. 13 is a diagram in which the second configuration example of the phase shifter shown in FIG. 3 is applied to an actual circuit. In this figure, the same parts as those in FIGS. 1 and 3 are denoted by the same reference numerals, and the description thereof will be appropriately omitted. In the phase shifter shown in FIG.
Variable capacitors 171a and 171b are used as variable reactance elements 170a and 170b, respectively. The frequency f _{0 of the} high-frequency transmission lines 123a, 123b =
45 degree electrical length θ at 5 GHz, characteristic impedance Z
= 70.7Ω. High-frequency transmission lines 123a, 123
Assume b is lossless. At this time, from the equation (7), the capacitance C of the capacitors 126a to 126d is 0.45
Set to pF. Further, the high-frequency transmission lines 123a, 1
Assume each of the equivalent characteristic impedance Z ₂ and Z ₂ = 50 [Omega each π-type circuit composed of a 23b and a capacitor 126a-126d. Further, it is assumed that the input / output impedance Z ₀ = 50Ω.

FIG. 14 shows a simulation result of amplitude characteristics (forward transfer coefficient S ₂₁ and input reflection coefficient S ₁₁ ) of the π-type circuit shown in FIG. 13 when characteristic impedance Z ₂ = 50Ω. . FIG. 15 is a diagram showing a simulation result of the phase characteristic (forward transfer coefficient S ₂₁ ) when the characteristic impedance of the π-type circuit is Z ₂ = 50Ω. 14, 15, equation (4), the variable capacitance C ₃ of the variable capacitor 171b capacitor 171a
The set to four times the capacity C _1, 0.05 pF capacitance C ₁ of the variable capacitor 171a, 0.1pF, 0.15pF,
The values were changed to 0.2 pF, 0.3 pF, 0.4 pF, and 0.5 pF. As can be seen from the figure, the frequency f = 2.
In the range of 9 GHz to 5.6 GHz, an amplitude variation of 0.1 GHz is used.
Within 5 dB, the input reflection amount is -11 dB or less (FIG. 1
4) A characteristic of a phase change amount of 60 degrees or more (FIG. 15) is obtained.

The phase shifter shown in FIG. 13 includes capacitors 126b and 126c separately. However, since the capacitors 126b and 126c are both connected to the input / output terminal 124c, the capacitance is 2
It is also possible to collectively replace with one capacitor C.

FIG. 16 shows the third structure of the phase shifter shown in FIG.
It is the figure which applied the example of a composition to an actual circuit. Smell this figure
Therefore, the same parts as those in FIGS.
The description will be omitted. In the phase shifter shown in FIG.
Variable keys are used as variable reactance elements 170a and 170b.
The capacitors 171a and 171b are used respectively.
You. Inductance L of inductors 133a to 133d
Is set to L = 1.6 nH. In addition, inductors 133a to
133d and capacitors 136a and 136b.
Frequency f of each T-type circuit₀At 5 GHz
Equivalent characteristic impedance Z_TwoTo Z_Two= 50Ω assumed
I do. At this time, from the equation (10), the capacitor 136 is obtained.
The capacitance C of a, 136b is set to 0.64 pF. Sa
In addition, the input / output impedance Z ₀= 50Ω.

FIG. 17 is a diagram showing a simulation result of amplitude characteristics (forward transfer coefficient S ₂₁ and input reflection coefficient S ₁₁ ) of the T-type circuit shown in FIG. 16 when characteristic impedance Z ₂ = 50Ω. . FIG. 18 is a diagram showing a simulation result of a phase characteristic (forward transfer coefficient S ₂₁ ) of the T-type circuit when the characteristic impedance is Z ₂ = 50Ω. 17, 18, equation (4), the variable capacitance C ₃ of the variable capacitor 171b capacitor 171a
The set to four times the capacity C _1, 0.05 pF capacitance C ₁ of the variable capacitor 171a, 0.1pF, 0.15pF,
0.2 pF, 0.3 pF, 0.4 pF, 0.5 pF,
It is changed to 0.6 pF, 0.7 pF, and 0.8 pF. As can be seen from the figure, the frequency f = 1.5 GHz-
At 5.8 GHz, the characteristics are obtained in which the amplitude fluctuation is within 0.5 dB, the input reflection amount is -12 dB or less (FIG. 17), and the phase change amount is 60 degrees or more (FIG. 18).

FIG. 19 shows the fourth structure of the phase shifter shown in FIG.
It is the figure which applied the example of a composition to an actual circuit. Smell this figure
1 and 5 are denoted by the same reference numerals, and
The description will be omitted. In the phase shifter shown in FIG.
Variable keys are used as variable reactance elements 170a and 170b.
The capacitors 171a and 171b are used respectively.
You. Inductance L of inductors 143a and 143b
Is set to L = 1.6 nH. Also, the inductors 143a,
143b and capacitors 146a to 146d.
Frequency f of each π-type circuit₀At 5 GHz
Equivalent characteristic impedance Z_TwoTo Z_Two= 50Ω assumed
I do. At this time, from equation (10), the capacitor 146a
The capacitance C of １４146d is set to 0.64 pF. Further
And the input / output impedance Z ₀= 50Ω.

FIG. 20 is a diagram showing a simulation result of amplitude characteristics (forward transmission coefficient S ₂₁ and input reflection coefficient S ₁₁ ) of the π-type circuit shown in FIG. 19 when the characteristic impedance Z ₂ = 50Ω. . FIG. 21 is a diagram showing a simulation result of a phase characteristic (forward transfer coefficient S ₂₁ ) when the characteristic impedance Z ₂ of the π-type circuit is 50Ω. 20, 21, equation (4), the variable capacitance C ₃ of the variable capacitor 171b capacitor 171a
The set to four times the capacity C _1, 0.05 pF capacitance C ₁ of the variable capacitor 171a, 0.1pF, 0.15pF,
The values were changed to 0.2 pF, 0.3 pF, 0.4 pF, and 0.5 pF. As can be seen from FIG.
In the range of 0 GHz to 5.5 GHz, the amplitude fluctuation is set to 0.1 GHz.
Within 5 dB, the input reflection amount is -11 dB or less (FIG. 2).
0), a characteristic having a phase change amount of 60 degrees or more (FIG. 18) is obtained.

The phase shifter shown in FIG. 19 includes capacitors 146b and 146c separately. However, since the capacitors 146b and 146c are both connected to the input / output terminal 144c, the capacitance is 2
It is also possible to collectively replace with one capacitor C.

FIG. 22 shows the fifth structure of the phase shifter shown in FIG.
It is the figure which applied the example of a composition to an actual circuit. Smell this figure
Therefore, the same parts as those in FIGS.
The description will be omitted. In the phase shifter shown in FIG.
Variable keys are used as variable reactance elements 170a and 170b.
The capacitors 171a and 171b are used respectively.
You. Inductance L of inductors 153a and 153b
Is set to L = 1.6 nH. In addition, inductors 153a,
153b and capacitors 156a to 156d.
Frequency f of each T-type circuit₀At 5 GHz
Equivalent characteristic impedance Z_TwoTo Z_Two= 50Ω assumed
I do. At this time, from equation (10), the capacitor 156a
The capacitance C of １５156d is set to 0.64 pF. Further
And the input / output impedance Z ₀= 50Ω.

FIG. 23 is a diagram showing a simulation result of amplitude characteristics (forward transmission coefficient S ₂₁ and input reflection coefficient S ₁₁ ) of the T-type circuit shown in FIG. 22 when the characteristic impedance Z ₂ = 50Ω. . FIG. 24 is a diagram showing a simulation result of a phase characteristic (forward transfer coefficient S ₂₁ ) of the T-type circuit when the characteristic impedance Z ₂ = 50Ω. 23, FIG. 24, the formula (4), the variable capacitance C ₃ of the variable capacitor 171b capacitor 171a
The set to four times the capacity C _1, 0.05 pF capacitance C ₁ of the variable capacitor 171a, 0.1pF, 0.15pF,
The values were changed to 0.2 pF, 0.3 pF, 0.4 pF, and 0.5 pF. As can be seen from the figure, the frequency f = 4.
In the range of 8 GHz to 5.2 GHz, the amplitude variation is set to 0.1 GHz.
Within 5 dB, the input reflection amount is -20 dB or less (FIG. 2).
3) A characteristic having a phase change amount of 90 degrees or more (FIG. 24) is obtained.

FIG. 25 shows the sixth structure of the phase shifter shown in FIG.
It is the figure which applied the example of a composition to an actual circuit. Smell this figure
1 and 7 are denoted by the same reference numerals, and
The description will be omitted. In the phase shifter shown in FIG.
Variable keys are used as variable reactance elements 170a and 170b.
The capacitors 171a and 171b are used respectively.
You. Inductance L of inductors 163a to 163d
Is set to L = 1.6 nH. In addition, the inductors 163a to
163d and capacitors 166a and 166b.
Frequency f of each π-type circuit₀At 5 GHz
Equivalent characteristic impedance Z_TwoTo Z_Two= 50Ω assumed
I do. At this time, from the equation (10), the capacitor 166 is obtained.
The capacitance C of a, 166b is set to 0.64 pF. Sa
In addition, the input / output impedance Z ₀= 50Ω.

FIG. 26 is a diagram showing a simulation result of amplitude characteristics (forward transfer coefficient S ₂₁ and input reflection coefficient S ₁₁ ) of the π-type circuit shown in FIG. 25 when the characteristic impedance Z ₂ = 50Ω. . FIG. 27 is a diagram showing a simulation result of a phase characteristic (forward transfer coefficient S ₂₁ ) when the characteristic impedance Z ₂ of the π-type circuit is 50Ω. 26, 27, equation (4), the variable capacitance C ₃ of the variable capacitor 171b capacitor 171a
The set to four times the capacity C _1, 0.05 pF capacitance C ₁ of the variable capacitor 171a, 0.1pF, 0.15pF,
The values were changed to 0.2 pF, 0.3 pF, 0.4 pF, and 0.5 pF. As can be seen from the figure, the frequency f = 4.
In the range of 3 GHz to 5.7 GHz, an amplitude variation of 0.
Within 5 dB, the input reflection amount is -20 dB or less (FIG. 2).
6) A characteristic of a phase change amount of 80 degrees or more (FIG. 27) is obtained.

The phase shifter shown in FIG. 25 includes an inductor 163b and an inductor 163c individually. However, since both of the inductors 163b and 163c are connected to the input / output terminal 164c, it is possible to replace the inductors with one inductor having an inductance of L / 2.

[0092] 2. Embodiment using resonance circuit as high-frequency impedance element: FIG. 28 is a circuit diagram showing another configuration of the phase shifter according to the present invention. The phase shifter shown in FIG. 1 uses variable reactance elements 170a and 170b as the first and second high-frequency impedance elements, respectively. As shown in FIG. Resonant circuit 17 as an element
A phase shifter can also be configured using 2a and 172b. The resonance circuits 172a and 172b are configured using an inductor, a capacitor, an inductance component realized by a transmission line, a capacitance component realized by a transmission line, and the like. The impedance of the resonance circuits 172a and 172b is substantially constituted by reactance. FIG.
8 differs from the phase shifter shown in FIG. 1 only in the configuration of the first and second high-frequency impedance elements, and the phase shifter shown in FIG. The same operation as the phase shifter is performed.

Here, the input impedance of the input terminal 101
And the output impedance of the output terminal 102.
Z₀, Frequency f of high-frequency phase shift elements 103a and 103b₀
Is 90 degrees, and the high-frequency phase shifting elements 103a, 103a
03b is replaced by a high-frequency transmission line.
Z_TwoOf the resonance circuit 172a
X₁, The reactance of the resonance circuit 172b is X_ThreeWhen
I do. In this case, the input reflection coefficient of the phase shifter shown in FIG.
Number S₁₁And output reflection coefficient S _{twenty two}Is

[0094]

(Equation 4)

Can be expressed by Therefore, the reactance X
_{3 ^{X 3 = Z 2 2 X 1}} / (4Z 0 2) By setting the relation of (14), since the input and output reflection coefficients S _11, S ₂₂ at the frequency f ₀ becomes zero , The input / output impedance at the frequency f ₀ can be matched. In addition,
When actually producing a phase shifter, the input / output reflection coefficients S ₁₁ and S ₂₂ at the frequency f ₀ need not be strictly zero, but a sufficient effect can be obtained if they are substantially zero.

At this time, the phase shifter shown in FIG.
Forward transfer coefficient S_{twenty one}And reverse transfer coefficient S₁₂Is S_{twenty one}= S₁₂= (2Z₀-X₁) / (2Z₀+ X₁) (15) To operate as a phase shifter, a resonant circuit
Reactance X of 172a and 172b ₁, X_ThreeAt the same time
It may be changed continuously. X₁From (X₁+ ΔX
₁), The phase change amount θ of the phase shifter is
Given by the formula. θ = -2tan^-1[(X₁+ ΔX₁) / (2Z₀)] + 2 tan^-1[X₁/ (2Z₀)] [Rad] (16)

By using the resonance circuits 172a and 172b as the first and second high-frequency impedance elements, the amount of phase change can be increased. Further, similarly to the phase shifter shown in FIG.
a, 103b is, (a) the electrical length at the frequency f ₀ is 90 degrees of the high-frequency transmission line (see FIG. 2), a small high-frequency transmission line and its ends than the electrical length of 90 degrees at (b) frequency f ₀ A π-type circuit composed of two capacitors having one end connected and the other end grounded (see FIG. 3), (c) a lumped constant circuit composed of an inductor and a capacitor (see FIGS. 4 to 7), etc. It is configured using In the above configuration example,
The phase shifter can be downsized in the order of (a)>(b)> (c).

(Specific Example and Characteristics of Phase Shifter) Next, FIG.
8 shows a specific example of the phase shifter shown in FIG. 8, and shows a simulation result of amplitude characteristics and phase characteristics of the specific example.
FIG. 29 is a circuit diagram showing a specific example of the phase shifter shown in FIG. 28, the same parts as those in FIGS. 2 and 28 are denoted by the same reference numerals, and the description thereof will be appropriately omitted. The phase shifter shown in FIG. 29 corresponds to the resonance circuit 1 shown in FIG.
As 72a and 172b, a series resonance circuit in which an inductor and a capacitor are connected in series is used. Specifically, the resonance circuit 172a includes a series resonance circuit in which an inductor 191a and a variable capacitor 181a are connected in series, and the resonance circuit 172b includes an inductor 191a.
b and a variable capacitor 181b are connected in series to form a series resonance circuit.

In this phase shifter, high-frequency transmission lines 113a and 113b having an electrical length of 90 degrees at a frequency f ₀ = 5 GHz are used as high-frequency phase shift elements 103a and 103b having an impedance conversion function, respectively. Assuming that the high-frequency transmission lines 113a and 113b are lossless, the input / output impedance Z ₀ = 50Ω. FIG.
FIG. 8 is a diagram showing a simulation result of amplitude characteristics (forward transmission coefficient S ₂₁ and input reflection coefficient S ₁₁ ) when the characteristic impedance Z ₂ = 70.7Ω of the high-frequency transmission lines 113a and 113b. FIG. 31 also shows the characteristic impedance Z ₂ = 7 of the high-frequency transmission lines 113a and 113b.
FIG. 14 is a diagram illustrating a simulation result of a phase characteristic (forward transfer coefficient S ₂₁ ) in the case of 0.7Ω.

In FIGS. 30 and 31, the inductor 191a
Let L ₁ = 4 nH. At this time, from the equation (14), the inductance L of the inductor 191b is obtained.
₃ is _１ ／ of the inductance L ₁ of the inductor 191a.
Set to double. Similarly, from the expression (14), the capacitance C ₃ of the variable capacitor 181b is set to twice the capacitance C ₁ of the variable capacitor 181a, and the capacitance C ₃ of the variable capacitor 181a is set.
₁ is 0.05 pF, 0.1 pF, 0.15 pF, 0.2
pF, 0.3 pF, 0.4 pF, and 0.5 pF. As can be seen from the figure, the frequency f = 4.0 GH
0.5 dB as amplitude fluctuation at z to 6.0 GHz
Within this range, characteristics of an input reflection amount of −12 dB or less (FIG. 30) and a phase change amount of 210 degrees or more (FIG. 31) are obtained.

Similarly, FIG.
Amplitude characteristics when the characteristic impedance Z ₂ = 50Ω of the a and 113b (forward transmission coefficient S ₂₁ and input reflection coefficient S ₁₁ )
It is a figure showing the simulation result of. FIG. 33
FIG. 9 is a diagram showing a simulation result of a phase characteristic (forward transfer coefficient S ₂₁ ) in the case where the characteristic impedance Z ₂ = 50Ω of the high-frequency transmission lines 113a and 113b.

32 and 33, the inductor 191a
Let L ₁ = 4 nH. At this time, from the equation (14), the inductance L of the inductor 191b is obtained.
₃ 1/4 of the inductor 191a of the inductance L ₁
Set to double. Similarly, from the equation (14), the capacity C ₃ of the variable capacitor 181b is set to be four times the capacity C ₁ of the variable capacitor 181a, and the capacity C ₃ of the variable capacitor 181a is set.
₁ is 0.05 pF, 0.1 pF, 0.15 pF, 0.2
pF, 0.3 pF, 0.4 pF, and 0.5 pF. As can be seen from the figure, the frequency f = 4.0 GH
0.5 dB as amplitude fluctuation at z to 6.0 GHz
Within this range, the input reflection amount is -10 dB or less (FIG. 32), and the phase change amount is 200 degrees or more (FIG. 33).

FIG. 34 is a circuit diagram showing another specific example of the phase shifter shown in FIG. 28, the same parts as those in FIGS. 2 and 28 are denoted by the same reference numerals, and the description thereof will be appropriately omitted. The phase shifter shown in FIG. 34 uses a parallel resonance circuit in which an inductor and a capacitor are connected in parallel as the resonance circuits 172a and 172b shown in FIG. Specifically, the resonance circuit 172a includes a parallel resonance circuit in which an inductor 192a and a variable capacitor 182a are connected in parallel, and the resonance circuit 172b includes a parallel resonance circuit in which an inductor 192b and a variable capacitor 182b are connected in parallel. Be composed.

In this phase shifter, high-frequency transmission lines 113a and 113b having an electrical length of 90 degrees at a frequency f ₀ = 5 GHz are used as high-frequency phase shift elements 103a and 103b having an impedance conversion function, respectively. Assuming that the high-frequency transmission lines 113a and 113b are lossless, the input / output impedance Z ₀ = 50Ω. FIG.
FIG. 8 is a diagram showing a simulation result of amplitude characteristics (forward transmission coefficient S ₂₁ and input reflection coefficient S ₁₁ ) when the characteristic impedance Z ₂ = 70.7Ω of the high-frequency transmission lines 113a and 113b. FIG. 36 also shows the characteristic impedance Z ₂ = 7 of the high-frequency transmission lines 113a and 113b.
FIG. 14 is a diagram illustrating a simulation result of a phase characteristic (forward transfer coefficient S ₂₁ ) in the case of 0.7Ω.

In FIGS. 35 and 36, the inductor 192a
Let L ₁ = 4 nH. At this time, from the equation (14), the inductance L of the inductor 192b is obtained.
₃ is の of the inductance L ₁ of the inductor 192a.
Set to double. Similarly, from the equation (14), the capacitance C ₃ of the variable capacitor 182b is set to twice the capacitance C ₁ of the variable capacitor 182a, and the capacitance C ₃ of the variable capacitor 182a is set.
₁ is 0.05 pF, 0.1 pF, 0.15 pF, 0.2
pF, 0.3 pF, 0.4 pF, and 0.5 pF. As can be seen from the figure, the frequency f = 4.0 GH
0.5 dB as amplitude fluctuation at z to 6.0 GHz
Within this range, a characteristic having an input reflection amount of −12 dB or less (FIG. 35) and a phase change amount of 90 degrees or more (FIG. 36) are obtained.

Similarly, FIG. 37 shows the high-frequency transmission line 113.
Amplitude characteristics when the characteristic impedance Z ₂ = 50Ω of the a and 113b (forward transmission coefficient S ₂₁ and input reflection coefficient S ₁₁ )
It is a figure showing the simulation result of. FIG. 38
FIG. 9 is a diagram showing a simulation result of a phase characteristic (forward transfer coefficient S ₂₁ ) in the case where the characteristic impedance Z ₂ = 50Ω of the high-frequency transmission lines 113a and 113b.

In FIGS. 37 and 38, the inductor 192a
Let L ₁ = 4 nH. At this time, from the equation (14), the inductance L of the inductor 192b is obtained.
₃ 1/4 of the inductor 192a of the inductance L ₁
Set to double. Similarly, from the equation (14), the capacity C ₃ of the variable capacitor 182b is set to be four times the capacity C ₁ of the variable capacitor 182a, and the capacity C ₃ of the variable capacitor 182a is set.
₁ is 0.05 pF, 0.1 pF, 0.15 pF, 0.2
pF, 0.3 pF, 0.4 pF, and 0.5 pF. As can be seen from the figure, the frequency f = 4.0 GH
In the range of z to 6.0 GHz, the amplitude variation is within 0.5 dB, the input reflection amount is -13 dB or less (FIG. 37), and the phase change amount is 100 degrees or more (FIG. 38).

FIG. 39 is a circuit diagram showing another specific example of the phase shifter shown in FIG. 28, the same parts as those in FIGS. 2 and 28 are denoted by the same reference numerals, and the description thereof will be appropriately omitted. In the phase shifter shown in FIG. 39, as the resonance circuits 172a and 172b shown in FIG. 28, a second capacitor is connected in parallel to a series resonance circuit in which an inductor and a first capacitor are connected in series. This uses a complex resonance circuit. Specifically, the inductor 193a and the first variable capacitor 183a are connected in series to form a series resonance circuit, and the series resonance circuit and the second variable capacitor 183b are connected in parallel to form a composite resonance circuit. You. This composite resonance circuit is used as the resonance circuit 172a. Further, the inductor 193b and the first variable capacitor 183c are connected in series to form a series resonance circuit, and this series resonance circuit and the second variable capacitor
3d are connected in parallel to form a composite resonance circuit. This composite resonance circuit is used as the resonance circuit 172b.

In this phase shifter, high-frequency transmission lines 113a and 113b having a 90-degree electrical length at a frequency f ₀ = 5 GHz are used as high-frequency phase shift elements 103a and 103b having an impedance conversion function, respectively. Assuming that the high-frequency transmission lines 113a and 113b are lossless, the input / output impedance Z ₀ = 50Ω. FIG.
FIG. 8 is a diagram showing a simulation result of amplitude characteristics (forward transmission coefficient S ₂₁ and input reflection coefficient S ₁₁ ) when the characteristic impedance Z ₂ = 70.7Ω of the high-frequency transmission lines 113a and 113b. FIG. 41 also shows the characteristic impedance Z ₂ = 7 of the high-frequency transmission lines 113a and 113b.
FIG. 14 is a diagram illustrating a simulation result of a phase characteristic (forward transfer coefficient S ₂₁ ) in the case of 0.7Ω.

In FIGS. 40 and 41, the inductor 193a
And the variable capacitor 1 has an inductance L ₁ = 4 nH.
83a, and equally C ₁ capacity 183b, variable capacitor 183c, and equally C ₃ capacity 183d. At this time, from the equation (14), the inductance L _{3 of the} inductor 193b is changed to the inductance L of the inductor 193a.
Set to 1/2 of ₁ . Similarly, from equation (14), the capacitance C ₃ of the variable capacitors 183c and 183d is set to twice the capacitance C ₁ of the variable capacitors 183a and 183b,
Variable capacitors 183a, the capacitance C ₁ of 183b 0.0
5 pF, 0.1 pF, 0.15 pF, 0.2 pF, 0.
The values are changed to 3 pF and 0.4 pF. As shown in the figure, at a frequency f = 4.0 GHz to 6.0 GHz, the amplitude fluctuation is within 0.5 dB, the input reflection amount is -12 dB or less (FIG. 40), and the phase change amount is 170 degrees or more (FIG. 4).
The characteristic of 1) is obtained.

Similarly, FIG.
Amplitude characteristics when the characteristic impedance Z ₂ = 50Ω of the a and 113b (forward transmission coefficient S ₂₁ and input reflection coefficient S ₁₁ )
It is a figure showing the simulation result of. FIG.
FIG. 9 is a diagram showing a simulation result of a phase characteristic (forward transfer coefficient S ₂₁ ) in the case where the characteristic impedance Z ₂ = 50Ω of the high-frequency transmission lines 113a and 113b.

42 and 43, the inductor 193a
And the variable capacitor 1 has an inductance L ₁ = 4 nH.
83a, and equally C ₁ capacity 183b, variable capacitor 183c, and equally C ₃ capacity 183d. At this time, from the equation (14), the inductance L _{3 of the} inductor 193b is changed to the inductance L of the inductor 193a.
Set to 1/4 times ₁ . Similarly, from equation (14), the capacitance C ₃ of the variable capacitors 183c and 183d is set to be four times the capacitance C ₁ of the variable capacitors 183a and 183b,
Variable capacitors 183a, the capacitance C ₁ of 183b 0.0
5 pF, 0.1 pF, 0.15 pF, 0.2 pF, 0.
The values are changed to 3 pF and 0.4 pF. As can be seen from the figure, at a frequency f = 4.0 GHz to 6.0 GHz, the amplitude fluctuation is within 0.5 dB, the input reflection amount is -10 dB or less (FIG. 42), and the phase change amount is 160 degrees or more (FIG. 43). Characteristics are obtained.

FIG. 44 is a circuit diagram showing another specific example of the phase shifter shown in FIG. 28, the same parts as those in FIGS. 2 and 28 are denoted by the same reference numerals, and the description thereof will be appropriately omitted. The phase shifter shown in FIG. 44 uses, as the resonance circuits 172a and 172b shown in FIG. 28, a composite resonance circuit in which two series resonance circuits in which an inductor and a capacitor are connected in series are connected in parallel. It is. Specifically, one series resonance circuit is formed by connecting inductor 194a and variable capacitor 184a in series, and the other series resonance circuit is formed by connecting inductor 194b and variable capacitor 184b in series. A composite resonance circuit in which these two series resonance circuits are connected in parallel is used as the resonance circuit 172a. In addition, the inductor 194c
And the variable capacitor 184c are connected in series to form one series resonance circuit, and the inductor 194d and the variable capacitor 184d are connected in series to form the other series resonance circuit. A composite resonance circuit in which these two series resonance circuits are connected in parallel is used as the resonance circuit 172b.

In this phase shifter, high-frequency transmission lines 113a and 113b having an electrical length of 90 degrees at a frequency f ₀ = 5 GHz are used as high-frequency phase shift elements 103a and 103b having an impedance conversion function, respectively. Assuming that the high-frequency transmission lines 113a and 113b are lossless, the input / output impedance Z ₀ = 50Ω. FIG.
FIG. 8 is a diagram showing a simulation result of amplitude characteristics (forward transmission coefficient S ₂₁ and input reflection coefficient S ₁₁ ) when the characteristic impedance Z ₂ = 70.7Ω of the high-frequency transmission lines 113a and 113b. FIG. 46 also shows the characteristic impedance Z ₂ = 7 of the high-frequency transmission lines 113a and 113b.
FIG. 14 is a diagram illustrating a simulation result of a phase characteristic (forward transfer coefficient S ₂₁ ) in the case of 0.7Ω.

45 and 46, the inductor 194a
, The inductance L ₁ = 4 nH, and the inductor 194
set b of the inductance L ₂ to 1/2 of the inductance L ₁ of the inductor 194a. The variable capacitor 184a, and equal C ₁ capacity 184b, variable capacitor 184 c, and equally C ₃ capacity 184d. In this case, setting the equation (14), the inductance L ₃ of the inductor 194c is set to 1/2 of the inductance L ₁ of the inductor 194a, the inductance L4 of the inductor 194d to 1/2 of the inductance L ₂ of the inductor 194b I do. Similarly, from equation (14), variable capacitor 184 c, capacitor C ₃ to the variable capacitor 184a of 184d, is set to 2 times the capacity C ₁ of 184b, variable capacitors 184a, 184b 0.3 pF capacitance C ₁ of 0. The values were changed to 35 pF and 0.4 pF. As can be seen from the figure, the frequency f = 4.6 GHz-
At 5.4 GHz, a characteristic having an amplitude fluctuation of 0.5 dB or less, an input reflection amount of −20 dB or less (FIG. 45), and a phase change amount of 100 degrees or more (FIG. 46) are obtained.

Similarly, FIG. 47 shows the high-frequency transmission line 113.
Amplitude characteristics when the characteristic impedance Z ₂ = 50Ω of the a and 113b (forward transmission coefficient S ₂₁ and input reflection coefficient S ₁₁ )
It is a figure showing the simulation result of. FIG.
FIG. 9 is a diagram showing a simulation result of a phase characteristic (forward transfer coefficient S ₂₁ ) in the case where the characteristic impedance Z ₂ = 50Ω of the high-frequency transmission lines 113a and 113b.

47 and 48, the inductor 194a
, The inductance L ₁ = 4 nH, and the inductor 194
set b of the inductance L ₂ to 1/2 of the inductance L ₁ of the inductor 194a. The variable capacitor 184a, and equal C ₁ capacity 184b, variable capacitor 184 c, and equally C ₃ capacity 184d. In this case, setting the equation (14), the inductance L ₃ of the inductor 194c is set to 1/4 of the inductance L ₁ of the inductor 194a, the inductance L4 of the inductor 194d to 1/4 of the inductance L ₂ of the inductor 194b I do. Similarly, from equation (14), variable capacitor 184 c, capacitor C ₃ to the variable capacitor 184a of 184d, is set to 4 times the capacity C ₁ of 184b, variable capacitors 184a, 184b 0.3 pF capacitance C ₁ of 0. The values were changed to 35 pF and 0.4 pF. As can be seen from the figure, the frequency f = 4.7 GHz-
At 5.3 GHz, characteristics of amplitude fluctuation within 0.5 dB, input reflection amount of -20 dB or less (FIG. 47), and phase change amount of 160 degrees or more (FIG. 48) are obtained.

(Prototype Production and Experimental Results of MMIC Phase Shifter)
The phase shifter according to the present invention described above is suitable for being formed by an MMIC. FIG. 49 is a circuit diagram showing a specific prototype example of the phase shifter shown in FIG. FIG. 1 shows a circuit configuration of an MMIC phase shifter using a coplanar line. 29, the same parts as those of FIG. 29 are denoted by the same reference numerals, and the description thereof will be appropriately omitted. This MMI
In the C process, an Au conductor coplanar line (characteristic impedance Z ₂ = 50Ω) 113aa, 113bb,
Inductors 191a1, 191a2, 191b1, 19
1b2, resistor 185, capacitor 186, GaAsME
SFETs 181a1, 181a2, 181b1, 181
b2 is formed. GaAs MESFET 181a1,
The gate lengths of 181a2, 181b1, and 181b2 are 0.
3 μm, transconductance is g _m = 200 mS /
mm or more, and the cutoff frequency is f _T = 20 GHz or more.

In this phase shifter, the GaAs MESFET 1
81a1, 181a2, 181b1, and 181b2 are connected to their respective drain terminals and source terminals to form GaAs.
sMESFETs 181a1, 181a2, 181b1,
The Schottky gate capacitance of 181b2 is used as the capacitance of the varactor diode FETc. GaAsME
SFETs 181a1, 181a2, 181b1, 181
The gate width of b2 (ie, varactor diode FETc) is 80 μm.

Further, in order to secure the symmetry of the pattern layout and suppress variations in electrical characteristics, a series circuit of an inductor and a GaAs MESFET (ie, varactor diode FETc) having the same size is arranged in series and parallel, respectively. ing. Specifically, the inductance of the inductors 191a1 and 191a2, GaAs
The capacities of the sMESFETs 181a1 and 181a2 are the same, and the inductor 191a1 and the GaAsM
ESFET 181a1 series circuit and inductor 191a
2 and a series circuit of GaAs MESFET 181a2 are connected in series. In addition, the inductor 191b
1,191b2 inductance, GaAsMESFE
The capacitances of T181b1 and 181b2 are the same, and the inductor 191b1 and the GaAs MESFET
181b1 and the inductor 191b2 and G
The series circuit of aAsMESFET 181b2 is connected in parallel.

GaAs MESFETs 181a1, 181
a2 is connected in common to a voltage terminal 181a3 via a resistor 185,
GaAsME according to the voltage Vg1 applied from 1a3
SFET (ie, varactor diode FETc) 1
The capacitances of 81a1 and 181a2 change. Similarly, Ga
The gate terminals of the AsMESFETs 181b1 and 181b2 are commonly connected to the voltage terminal 181b3, and the capacitance of the GaAs MESFETs (that is, the varactor diode FETc) 181b1 and 181b2 changes according to the voltage Vg2 applied from the voltage terminal 181b3. In addition, GaAs MESFETs 181b1 and 181b
Each of the gate terminals 2 is grounded at a high frequency via a capacitor 186.

FIG. 50 is a plan view of the prototype example shown in FIG. The chip size of this prototype is 0.91mm ×
It is as small as 0.78 mm (= 0.71 mm ² ). FIG.
FIG. 1 is a diagram showing a measurement result of an amplitude characteristic (input reflection coefficient S ₁₁ ) when the characteristic impedance Z ₂ = 50Ω of the coplanar lines 113aa and 113bb. The horizontal axis represents the frequency [GHz], and the vertical axis represents the input reflection coefficient S ₁₁ [dB].
FIG. 52 also shows a coplanar line 113aa, 1
If the characteristic impedance Z ₂ = 50 [Omega of 13bb is a diagram showing a measurement result of the amplitude characteristics of (forward transfer factor S _21). The horizontal axis is frequency [GHz], and the vertical axis is forward transmission coefficient S.
₂₁ [dB]. FIG. 53 also shows the characteristic impedance Z ₂ of the coplanar lines 113aa and 113bb.
FIG. 9 is a diagram showing a measurement result of a phase characteristic (forward transfer coefficient S ₂₁ ) when = 50Ω. The horizontal axis is frequency [GHz], and the vertical axis is forward transfer coefficient S ₂₁ [deg. ].

In FIG. 51 to FIG. 53, the voltages Vg1 and Vg2 are changed from −5.0 V to +0.4 V while 0 V is applied to the input terminal 101 and the output terminal 102 from the bias terminal of the network analyzer. As can be seen from the figure, at a frequency f = 19 GHz to 24 GHz, the input reflection amount is -10 dB or less (FIG. 51).
Good characteristics are obtained in which the amplitude fluctuation is within 0.8 dB (FIG. 52) and the phase change amount is 100 degrees or more (FIG. 53). Although a GaAs substrate was used in this prototype, it goes without saying that good characteristics can also be obtained by an MMIC process using a semiconductor substrate such as Si or InP. Furthermore, although a coplanar line is used as the transmission line, it goes without saying that good characteristics can be obtained similarly by using a microstrip line or the like.

As described above, the phase shifter according to the present invention has the MM
Suitable for being formed in an IC. By forming the phase shifter with the MMIC, a small phase shifter can be created. In addition, since a highly uniform chip can be manufactured without adjustment by a semiconductor process, mass productivity can be improved. Furthermore, high integration and high reproducibility make it possible to reduce mounting cost and improve reliability.

(Comparison between the prior art and the present invention) Next, the phase shifter according to the present invention and the conventional phase shifter will be compared. The conventional phase shifter shown in FIG. 62, a first configuration example of a phase shifter of the present invention shown in FIG. 2, the electrical length at the frequency f ₀ by using a 90-degree frequency transmission lines It is common in that it is configured.
Therefore, here, the phase shifter shown in FIG. 62 and the phase shifter shown in FIG. 2 are compared. In the conventional phase shifter shown in FIG. 62, as many as four high-frequency transmission lines 3a to 3d were required to form a 90-degree branch line hybrid. On the other hand, the phase shifter according to the present invention shown in FIG. 2 can be composed of two similar high-frequency transmission lines 113a and 113b. Therefore, the number of required high-frequency transmission lines can be reduced to half, so that a compact phase shifter having a quarter of the conventional size can be realized. In addition, by adopting various forms shown in FIGS. 3 to 7 as the high-frequency phase shift elements 103a and 103b, the phase shifter can be further reduced in size.

Further, a wider band can be achieved by the present invention. In the case of a phase shifter, the allowable range of the input reflection amount is -10 dB or less. For applications requiring high gain, the input reflection amount is -2.
Desirably, it is 0 dB or less. In the case of the conventional phase shifter shown in FIG. 62, as shown in FIG.
The frequency f is 4.9 GHz to 5.1 GHz in a band where the input reflection amount is −20 dB or less. On the other hand, in the case of the phase shifter according to the present invention shown in FIG. 2, the input reflection amount is -10 dB as shown in FIG.
The following band has a frequency f = 1.6 GHz to 6.0 GHz or more, and the band having an input reflection amount of −20 dB or less has a frequency f = 4.6 GHz to 5.4 GHz. in this way,
The phase shifter shown in FIG. 2 has a wider band. Even if the various forms shown in FIGS. 3 to 7 are adopted as the high-frequency phase shift elements 103a and 103b, a wide band can be realized similarly.

[Embodiment 2: Attenuator] FIG. 54 is a circuit diagram showing a configuration of an attenuator according to the present invention. Input terminal 20
A variable resistance element (first high-frequency impedance element) 270a is connected between the output terminal 1 and the output terminal 202. The impedance of the variable resistance element 270a is substantially composed of a resistance component. This resistance to R _1. Resistor R ₁ is variable. Incidentally, the output impedance of the input impedance and the output terminal 202 of the input terminal 201 and Z _0.

The input terminal 201 is connected to one end of the first high-frequency phase shift element 203a (the input / output terminal 204 of the high-frequency phase shift element).
a) is connected. One end of a second high-frequency phase shift element 203b (input / output terminal 204b of the high-frequency phase shift element) is connected to the output terminal 202. High frequency phase shift element 203
The other end of each of a and 203b (input / output terminal 204c of the high-frequency phase shift element) is commonly connected. Frequency phase shifting elements 203a, 203b are both phase change amount of the frequency f ₀ is 90 degrees, and has an impedance conversion function. Frequency phase shifting elements 203a, an equivalent characteristic impedance when replacing the 203b in the high-frequency transmission line and Z _2.

A variable resistance element (second high-frequency impedance element) 27 is connected to the input / output terminal 204c of the high-frequency phase shift element.
0b is connected to one end. The other end of the variable resistance element 270b is grounded. The impedance of the variable resistance element 270b is substantially composed of a resistance component. This resistance to R _3. Resistor R ₃ is variable. The above-described impedance conversion function of the high-frequency phase shift elements 203a and 203b means that the impedance of the variable resistance element 270b is converted, and when the converted impedance of the variable resistance element 270b and the impedance of the Phase element input / output terminals 204a,
This is a function to make the input / output reflection coefficient viewed from the point 204b substantially zero and to achieve matching.

Next, the operation of the attenuator shown in FIG. 54 will be described. The signal input from the input terminal 201 is transmitted through the first path through the variable resistance element 270a, the high-frequency phase shift element 203a, the variable resistance element 270b, and the high-frequency phase shift element 2a.
It is distributed to the second journey passing through 03b. When the frequency of the input signal is f ₀ , the signals distributed in the second path are given a phase change of 90 degrees by the high-frequency phase shift elements 203a and 203b.

In each process, the variable resistance element 270
a, a part of the signal power is absorbed by the resistors R ₁ and R _{3 of} the 270b. Signals that have not been absorbed in each process are combined at the input / output terminal 204b of the high-frequency phase shifter and output from the output terminal 202. And the variable resistance element 27
By simultaneously and continuously changing the resistances R ₁ and R ₃ of Oa and 270b, the attenuation of the attenuator shown in FIG. 54 can be changed continuously. Here, FIG.
The input reflection coefficient S ₁₁ and output reflection coefficient S ₂₂ of the attenuator shown in FIG.

[0132]

(Equation 5)

Can be expressed by Therefore, by setting the resistance R ₃ by the relational expression of R ₃ = Z ₂ ² R ₁ / (4Z ₀ ² ) (18), the input / output reflection coefficients S ₁₁ and S ₂₂ at the frequency f ₀ are obtained. Becomes zero, so that the input / output impedance at the frequency f ₀ can be matched. In addition,
When an attenuator is actually created, the input / output reflection coefficients S ₁₁ and S ₂₂ at the frequency f ₀ do not need to be strictly zero.

At this time, the forward transfer coefficient S ₂₁ and the backward transfer coefficient S ₁₂ of the attenuator shown in FIG. 54 are as follows: S ₂₁ = S ₁₂ = (2Z ₀ −R ₁ ) / (2Z ₀ + R ₁ ) (19) Also, the variable resistance elements 270a, 270b
When the resistances R ₁ and R ₃ are changed while maintaining the relationship of Expression (18), the attenuation L of the attenuator is given by the following expression. L = 20 log ₁₀ | (2Z ₀ + R ₁ ) / (2Z ₀ −R ₁ ) | [dB] (20)

High frequency phase shift element 203 having a phase change amount of 90 degrees at frequency f ₀ and having an impedance conversion function
a, 203b is, (a) the electrical length at the frequency f ₀ is 90 degrees of the high-frequency transmission line, the electrical length at the (b) frequency f ₀ 90
Using a π-type circuit composed of a high-frequency transmission line having a smaller frequency and two capacitors having one end connected to both ends and the other end grounded, and (c) a lumped constant circuit composed of an inductor and a capacitor. Be composed. In the above configuration example, the size of the attenuator can be reduced in the order of (a)>(b)> (c). Matching conditions of attenuator using a high-frequency phase shifting elements having such a structure, Figures 2-7 reactance X ₁ in the matching conditions of the phase shifter shown in, X ₃ each resistor R _1, R _By replacing with ₃ , it can be easily derived.

(A) As the high frequency phase shift elements 203a and 203b having the impedance conversion function, the frequency f ₀
-Frequency transmission lines 213a and 213 having an electrical length of 90 degrees
b (see FIG. 55): this high-frequency transmission line 2
Assuming that the characteristic impedance of the variable resistors 13a and 213b is Z ₂ , the resistances R ₁ and R _{3 of the} variable resistance elements 271a and 271b are
Is set to a relationship as shown in Expression (18),
The input / output impedance at the frequency f ₀ can be matched.

(B) As the high-frequency phase shift elements 203a and 203b having the impedance conversion function, the frequency f ₀
-Frequency transmission line 123 whose electrical length θ is smaller than 90 degrees
a and 123b, and 2 connected between both ends of the high-frequency transmission lines 123a and 123b and the ground, respectively.
Capacitors 126a, 126b and 126c, 1
26d (see FIGS. 3 and 54): high-frequency transmission lines 123a and 123b
Is the characteristic impedance of Z, and capacitors 126a-12
Let C be the capacity of 6d. This capacitance C is set to C = 1 / (2πf ₀ Ztanθ) (21). In this case, the resistance R of the variable resistance element 270b is
₃ by setting the relationship of ^{_{R 3 = (Zsinθ) 2 R}} 1 / (4Z 0 2) ··· (22), can be matched input and output impedances at a frequency f _0.

(C-1) As high-frequency phase shift elements 203a and 203b having an impedance conversion function, capacitors 136a and 136b each having one end grounded, and two terminals each having one end connected to the other end of each of the capacitors 136a and 136b. Inductors 133a, 133b
In the case of using a T-type circuit composed of the inductors 133c and 133d (see FIGS. 4 and 54):
The inductance of 3a to 133d is L, the capacitor 13
Let C be the capacity of 6a, 136b. This capacitance C is set to C = 1 / [(2πf ₀ ) ² L] (23). In this case, the resistance R of the variable resistance element 270b is
By setting ₃ as R ₃ = (2πf ₀ L) ² R ₁ / (4Z ₀ ² ) (24), the input / output impedance at the frequency f ₀ can be matched.

(C-2) Inductors 143a and 143b and high-frequency phase shifting elements 203a and 203b having an impedance conversion function
a and 146b and two capacitors 146a, 146b and 146 respectively connected between both ends of the capacitors 146a and 143b and the ground.
In the case of using a π-type circuit composed of c and 146d (see FIGS. 5 and 54): inductors 143a and 143d
3b is L, capacitors 146a-14
Let C be the capacity of 6d. This capacity C is set as in equation (23). In this case, the variable resistance elements 270a, 270
By setting the resistances R ₁ and R ₃ of b in such a relationship as shown in Expression (24), it is possible to match the input / output impedance at the frequency f ₀ .

(C-3) As high-frequency phase shift elements 203a and 203b having an impedance conversion function, inductors 153a and 153b having one end grounded, and two ends each having one end connected to the other end of each of inductors 153a and 153b. Capacitors 156a, 156b
Using a T-type circuit composed of the inductors 156c and 156d (see FIGS. 6 and 54):
3a and 153b have an inductance of L and a capacitor 15 has
Let C be the capacity of 6a to 156d. This capacitance C is calculated by the equation (2)
Set as in 3). In this case, the variable resistance element 270
By setting the resistances R ₁ and R ₃ of a and 270b to have the relationship shown in Expression (24), the input / output impedance at the frequency f ₀ can be matched.

(C-4) Capacitors 166a and 166b as high-frequency phase shift elements 203a and 203b having an impedance conversion function
a and 163b and two inductors 163a, 163b and 163 respectively connected between ground and ground.
In the case of using a π-type circuit composed of c and 163d (see FIGS. 7 and 54): inductors 163a to 163d
The inductance of the 3d is L, and the capacitors 166a, 16
Let C be the capacity of 6b. This capacity C is set as in equation (23). In this case, the variable resistance elements 270a, 270
By setting the resistances R ₁ and R ₃ of b in such a relationship as shown in Expression (24), it is possible to match the input / output impedance at the frequency f ₀ .

(Specific Example of Attenuator and Its Characteristics) Next, FIG.
4 shows a specific example of the attenuator shown in FIG. 4, and shows a simulation result of amplitude characteristics and phase characteristics of the specific example.
FIG. 55 is a circuit diagram showing a specific example of the attenuator shown in FIG. In this figure, the same parts as those of FIG. 54 are denoted by the same reference numerals, and the description thereof will be appropriately omitted. In the attenuator shown in FIG. 55, variable resistances 271a and 271b are used as variable resistance elements 270a and 270b, respectively. Further, as the high-frequency phase shift elements 203a and 203b having the impedance conversion function, the frequency f ₀ = 5GH.
high-frequency transmission lines 213a, 213 having an electrical length of 90 degrees at z.
3b are used. High frequency transmission line 213
a, 213b are assumed to be lossless and the input / output impedance Z ₀ = 50Ω. The input / output terminals 214a, 214b, 214c of the high-frequency transmission line correspond to the input / output terminals 204a to 204c of the high-frequency phase shift element.

FIG. 56 shows high-frequency transmission lines 213a and 213a.
3b is a diagram showing a simulation result of the forward transmission coefficient S ₂₁ when the characteristic impedance Z ₂ = 70.7Ω the. The horizontal axis is frequency [GHz], and the vertical axis is forward transmission coefficient S.
₂₁ [dB]. FIG. 57 also shows the characteristic impedance Z ₂ = 7 of the high-frequency transmission lines 213a and 213b.
Is a diagram illustrating a simulation result of the input reflection coefficient S ₁₁ when the 0.7Omu. The horizontal axis represents the frequency [GHz], and the vertical axis represents the input reflection coefficient S ₁₁ [dB]. Figure 56, in Figure 57, from the equation (18), the resistance R ₃ of the variable resistor 271b is set to 1/2 of the resistance R ₁ of the variable resistor 271a, 0 .OMEGA the resistance R ₁ of the variable resistor 271a, 60Ω, 85Ω , 95
Ω and 100Ω. As can be seen from the figure, at a frequency f = 4.5 GHz to 5.5 GHz,
The characteristics of attenuation of 24 dB or more (FIG. 56) and input reflection of -18 dB or less (FIG. 57) are obtained.

Similarly, FIG. 58 shows the high-frequency transmission line 213
FIG. 14A is a diagram showing a simulation result of the forward transfer coefficient S ₂₁ when the characteristic impedance Z ₂ = 50Ω of a and 213b. The horizontal axis represents frequency [GHz], and the vertical axis represents forward transfer coefficient S ₂₁ [dB]. FIG. 59 shows the characteristic impedance Z ₂ = 50 of the high-frequency transmission lines 213a and 213b.
Is a diagram illustrating a simulation result of the input reflection coefficient S ₁₁ when the Omega. The horizontal axis represents the frequency [GHz], and the vertical axis represents the input reflection coefficient S ₁₁ [dB]. 58 and 59, from equation (18), the resistance R ₃ of the variable resistor 271b is set to _１ ／ times the resistance R ₁ of the variable resistor 271a,
The resistance R ₁ of 1a 0Ω, 60Ω, 85Ω, 95Ω , 10
It is changed to 0Ω. As can be seen from the figure, at the frequency f = 4.5 GHz to 5.5 GHz, the attenuation is 28 dB or more (FIG. 58), and the input reflection is −16 dB.
B and below (FIG. 59) are obtained. It should be noted that the attenuator according to the present invention described above has an MMI like the phase shifter.
Suitable for forming with C.

[Embodiment 3: Non-linear signal generator] FIG. 60
FIG. 1 is a circuit diagram showing a configuration of a nonlinear signal generator according to the present invention. The first between the input terminal 301 and the output terminal 302
Are connected. Non-linear element 3
70a generates a non-linear signal according to the input signal power. The impedance of the nonlinear element 370a at the time of the small signal operation is Z _1, and the resistance component of the impedance Z ₁ is R ₁
And Incidentally, the output impedance of the input impedance and the output terminal 302 of the input and terminal 301 and Z _0.

The input terminal 301 is connected to one end of the first high-frequency phase shift element 303a (the input / output terminal 304 of the high-frequency phase shift element).
a) is connected. One end of a second high-frequency phase shift element 303b (input / output terminal 304b of the high-frequency phase shift element) is connected to the output terminal 302. High frequency phase shift element 303
The other end of each of a and 303b (input / output terminal 304c of the high-frequency phase shift element) is commonly connected. Both the high-frequency phase shift elements 303a and 303b have a phase change amount of 90 degrees at the frequency f ₀ and have an impedance conversion function. Frequency phase shifting elements 303a, an equivalent characteristic impedance when replaced by the high-frequency transmission line 303b and Z _2.

One end of a second nonlinear element 370b is connected to the input / output terminal 304c of the high-frequency phase shift element. The other end of the nonlinear element 370b is grounded. The nonlinear element 370b generates the same nonlinear signal as the nonlinear element 370a according to the input signal power. Non-linear element 3
The impedance at the time of the small signal operation of 70b is Z _3, and the resistance component of the impedance Z ₃ is R ₃ . The above-described impedance conversion function of the high-frequency phase shift elements 303a and 303b means that the impedance of the nonlinear element 370b is converted, and when the converted impedance of the nonlinear element 370b is combined with the impedance of the nonlinear element 370a, I / O terminals 304a, 304b
This is a function that makes the input / output reflection coefficient substantially zero when viewed from the point of view to achieve matching. Here, the input reflection coefficient S ₁₁ and the output reflection coefficient S ₂₂ of the nonlinear signal generator illustrated in FIG. 60,

[0148]

(Equation 6)

Can be expressed by Therefore, by setting the resistance R ₃ by the relational expression of R ₃ = Z ₂ ² R ₁ / (4Z ₀ ² ) (26), the input / output reflection coefficients S ₁₁ and S ₂₂ at the frequency f ₀ are obtained. Becomes zero, so that the input / output impedance at the frequency f ₀ can be matched.

At this time, the forward transfer coefficient S ₂₁ and the backward transfer coefficient S ₂₁ of the nonlinear signal generator shown in FIG.
_{12, S 21 = S 12 = (} 2Z 0 -R 1) / (2Z 0 + R 1) can be expressed by ... (27). Therefore, by setting the resistance R ₁ by the relational expression of R ₁ = 2Z ₀ (28), the forward transfer coefficient S ₂₁ and the backward transfer coefficient S ₁₂ at the frequency f ₀ become zero. Input / output reflection coefficients S ₁₁ , S ₂₂ = 0 and forward / backward transfer coefficient S
₂₁ , S ₁₂ = 0 means that the linear signal component of the input signal is completely absorbed. Therefore, no linear signal component is output from the nonlinear signal generator. When a nonlinear signal generator is actually created, the input / output reflection coefficients S ₁₁ and S ₂₂ and the forward / backward transmission coefficients S ₂₁ and S ₁₂ at the frequency f ₀ need not be exactly strictly zero. If it is zero, a sufficient effect can be obtained.

Next, the operation of the nonlinear signal generator shown in FIG. 60 will be described. The signal input from the input terminal 301 is distributed to a first path that passes through the nonlinear element 370a and a second path that passes through the high-frequency phase shift element 303a, the nonlinear element 370b, and the high-frequency phase shift element 303b. In each process, the linear signal components of the input signal are absorbed by the resistance components R ₁ and R ₃ of the impedances Z ₁ and Z ₃ of the nonlinear elements 370a and 370b. On the other hand, the same nonlinear signal is generated in the nonlinear elements 370a and 370b according to the power of the input signal. At this time, by setting the resistances R ₁ and R _{3 according} to the relations of the equations (26) and (28),
The linear signal component of the input signal is completely absorbed. Therefore, only the nonlinear signals generated in the nonlinear elements 370a and 370b are combined at the input / output terminal 304b of the high-frequency phase shift element and output from the output terminal 302.

High frequency phase shift element 303 having a phase change amount of 90 degrees at frequency f ₀ and having an impedance conversion function
a, 303b is, (a) the electrical length at the frequency f ₀ is 90 degrees of the high-frequency transmission line, the electrical length at the (b) frequency f ₀ 90
Using a π-type circuit composed of a high-frequency transmission line having a smaller frequency and two capacitors having one end connected to both ends and the other end grounded, and (c) a lumped constant circuit composed of an inductor and a capacitor. Be composed. In the above configuration example, the nonlinear signal generator can be miniaturized in the order of (a)>(b)> (c). Matching conditions of the nonlinear signal generator using a high-frequency phase shifting elements having such a structure, Figures 2-7 reactance X ₁ in the matching conditions of the phase shifter shown in, resistance X ₃ each R ₁ , by replacing the R _3, easily derivable. The matching conditions and the like of the nonlinear signal generator thus derived are exactly the same as the matching conditions and the like of the attenuator.

FIG. 61 is a circuit diagram showing a configuration example of the nonlinear signal generator shown in FIG. In this figure,
The same parts as those in FIG. 60 are denoted by the same reference numerals, and description thereof will be omitted as appropriate. In the nonlinear signal generator shown in FIG. 61, the high-frequency phase shifter 3 having an impedance conversion function
03a and 303b, the high-frequency transmission lines 313a and 313
13b are used. The input / output terminals 314a, 314b, 314c of the high-frequency transmission line correspond to the input / output terminals 304a to 304c of the high-frequency phase shift element.

The input / output terminals 314a, 314 of the high-frequency transmission line
Between the terminals 14b, diodes 371a and 372a, terminating resistors 373a, and DC blocking capacitors 374a, 375a, 376a, and 376 serve as first nonlinear elements 370a.
b, high frequency blocking inductor 377 and bias terminal 37
8 are connected. Specifically, the two diodes 371a and 372a are connected in parallel with opposite polarities, and the diodes 371a and 372a are further connected.
, A terminating resistor 373a is connected in parallel. A bias terminal 378 for supplying a bias current is connected to the anode of the diode 372a, and the high-frequency blocking inductor 377 is connected between the cathode of the diode 371a and the ground. And the diode 371
a, 372a and the DC blocking capacitor 37 such that a bias current flows through the inductor 377 for blocking high frequency.
4a, 375a, 376a, and 376b are connected. At this time, the diodes 371a and 372a and the terminating resistor 373a are connected to the DC blocking capacitors 374a and 374a.
75a, 376a and 376b are connected to input / output terminals 314a and 314b of the high-frequency transmission line in a high-frequency manner.

Also, the input / output terminal 314 of the high-frequency transmission line
A non-linear element composed of diodes 371b and 372b, a terminating resistor 373b, and DC blocking capacitors 374b and 375b is connected to c as the second non-linear element 370b. Specifically, two diodes 3
71b and 372b are connected in parallel with opposite polarities, and a terminating resistor 373b is connected in parallel with the diodes 371b and 372b. The anode of the diode 371b,
Cathode of diode 372b and termination resistor 373b
Is connected to the input / output terminal 314c. Also,
Anode of diode 372b and termination resistor 373b
Are the DC blocking capacitors 375b and 37, respectively.
4b is grounded at a high frequency, and the cathode of the diode 371b is directly grounded. And diode 3
A bias terminal 378 is connected to a connection between the capacitor 72b and the capacitor 375b. As described above, the bias current from the bias terminal 378 is applied to the diodes 371b and 37b.
2b.

Here, the input impedance of the input terminal 301
Output and output impedance of output terminal 302.
Z₀, The frequency f of the high-frequency transmission lines 313a and 313b₀
The electrical length at 90 °, the high-frequency transmission lines 313a, 313
b characteristic impedance Z_TwoTo Z ₀And Also die
Of the resistors 371a and 372a and the terminating resistor 373a.
Sei inbee dance Z₁, This synthetic imbi dance Z₁
The resistance component of R₁And diodes 371b, 372b
And the combined impedance of the terminating resistor 373b to Z _Three ,
This synthetic imbi dance Z_Three The resistance component of R_ThreeAnd

Next, the operation of the nonlinear signal generator shown in FIG. 61 will be described. The signal input from the input terminal 301 is composed of diodes 371 a and 372 a and a terminating resistor 37.
3a, and diodes 371b and 37
2b and a nonlinear element having a terminating resistor 373b. In this _{case, R 1 = 2Z 0, R} 3 = properly setting the bias current from the bias terminal 378 so that the Z _0/2. Accordingly, Expression (26) and Expression (2)
Since the relationship of 8) is satisfied, the linear signal component (that is, the fundamental wave) of the input signal is completely absorbed.

On the other hand, diodes 371a, 371
At b, 372a, and 372b, a nonlinear signal that is a harmonic of the input signal is generated. The nonlinear signal generated by the diodes 371a and 372a and the nonlinear signal generated by the diodes 371b and 372b are combined at the input / output terminal 314b of the high-frequency transmission line and output from the output terminal 302. Therefore, the linear signal component of the input signal is suppressed, and only the nonlinear signal is output from the output terminal 302. It should be noted that the nonlinear signal generator according to the present invention described above is suitable for being formed by an MMIC, like a phase shifter.

[Other Embodiments] All the embodiments described above are illustrative of the embodiments of the present invention and are not intended to limit the present invention. The present invention is not limited to the above-described various modifications and changes. It can be implemented in embodiments. Therefore,
The scope of the present invention is defined only by the claims and the equivalents thereof. Further, the phase shifter, the attenuator, and the nonlinear signal generator according to the present invention can be widely applied to a directivity control circuit of a wireless communication array antenna, a distortion compensation circuit of a power amplifier, and the like. Furthermore, for the phase shifter, a CDR (Clock and Data Recovery) for optical communication is used.
Circuit) can also be used as a variable clock delay circuit used in the circuit.

[0160]

As described above, the phase shifter and the attenuator according to the present invention include two high-frequency phase shift elements having a phase change of 90 degrees and two high-frequency impedance elements. The impedance of each high-frequency impedance element is set so that the output reflection coefficient becomes substantially zero. Also, the nonlinear signal generator according to the present invention has two high-frequency phase shift elements and two nonlinear elements, and the resistance component of the impedance of each nonlinear element is set so that the input / output reflection coefficient becomes substantially zero. I was doing it. As a result of this configuration, for example, when a high-frequency transmission line having an electrical length of 90 degrees at the frequency f ₀ is used as the high-frequency phase shift element,
Compared to the case of using a conventional 90-degree branch line hybrid using four high-frequency transmission lines having an electric length of 90 degrees at the frequency f ₀ , the number of required high-frequency transmission lines is half, and a phase shifter and an attenuation are required. And a non-linear signal generator. Therefore, by using the present invention, a compact phase shifter, attenuator and non-linear signal generator with a quarter or less of the conventional one can be realized.

In the phase shifter, attenuator and nonlinear signal generator according to the present invention, (a)
A high-frequency transmission line having an electrical length of 90 degrees at a frequency f ₀ ,
(B) a π-type circuit composed of a high-frequency transmission line having an electrical length of less than 90 degrees at a frequency f ₀ and two capacitors having one end connected to both ends and the other end grounded, and (c) an inductor and a capacitor. And a lumped constant circuit composed of In this case, (a)>(b)> (c)
The phase shifter, the attenuator, and the non-linear signal generator can be miniaturized in this order. In the phase shifter according to the present invention, a resonance circuit is used as the high-frequency impedance element. Thereby, the amount of phase change can be increased.

[Brief description of the drawings]

FIG. 1 is a circuit diagram showing a configuration of a phase shifter according to the present invention.

FIG. 2 is a circuit diagram showing a first configuration example of the phase shifter shown in FIG.

FIG. 3 is a circuit diagram showing a second configuration example of the phase shifter shown in FIG.

FIG. 4 is a circuit diagram showing a third configuration example of the phase shifter shown in FIG. 1;

FIG. 5 is a circuit diagram showing a fourth configuration example of the phase shifter shown in FIG. 1;

FIG. 6 is a circuit diagram showing a fifth configuration example of the phase shifter shown in FIG. 1;

FIG. 7 is a circuit diagram showing a sixth configuration example of the phase shifter shown in FIG. 1;

8 is a diagram in which the first configuration example of the phase shifter shown in FIG. 2 is applied to an actual circuit.

9 is a diagram illustrating an example of an amplitude characteristic of the phase shifter illustrated in FIG.

FIG. 10 is a diagram illustrating an example of a phase characteristic of the phase shifter illustrated in FIG. 8;

FIG. 11 is a diagram illustrating another example of the amplitude characteristic of the phase shifter illustrated in FIG. 8;

FIG. 12 is a diagram illustrating another example of the phase characteristics of the phase shifter illustrated in FIG. 8;

13 is a diagram in which the second configuration example of the phase shifter shown in FIG. 3 is applied to an actual circuit.

FIG. 14 is a diagram showing amplitude characteristics of the phase shifter shown in FIG.

FIG. 15 is a diagram showing phase characteristics of the phase shifter shown in FIG.

16 is a diagram in which the third configuration example of the phase shifter shown in FIG. 4 is applied to an actual circuit.

FIG. 17 is a diagram illustrating amplitude characteristics of the phase shifter illustrated in FIG. 16;

FIG. 18 is a diagram showing phase characteristics of the phase shifter shown in FIG.

19 is a diagram in which the fourth configuration example of the phase shifter shown in FIG. 5 is applied to an actual circuit.

FIG. 20 is a diagram illustrating amplitude characteristics of the phase shifter illustrated in FIG. 19;

FIG. 21 is a diagram showing phase characteristics of the phase shifter shown in FIG.

FIG. 22 is a diagram in which the fifth configuration example of the phase shifter shown in FIG. 6 is applied to an actual circuit.

FIG. 23 is a diagram illustrating amplitude characteristics of the phase shifter illustrated in FIG. 22;

FIG. 24 is a diagram showing phase characteristics of the phase shifter shown in FIG.

FIG. 25 is a diagram in which the sixth configuration example of the phase shifter shown in FIG. 7 is applied to an actual circuit.

FIG. 26 is a diagram showing amplitude characteristics of the phase shifter shown in FIG.

FIG. 27 is a diagram showing phase characteristics of the phase shifter shown in FIG.

FIG. 28 is a circuit diagram showing another configuration of the phase shifter according to the present invention.

FIG. 29 is a circuit diagram showing a specific example of the phase shifter shown in FIG.

FIG. 30 is a diagram illustrating an example of an amplitude characteristic of the phase shifter illustrated in FIG. 29;

FIG. 31 is a diagram illustrating an example of a phase characteristic of the phase shifter illustrated in FIG. 29;

32 is a diagram illustrating another example of the amplitude characteristic of the phase shifter illustrated in FIG. 29.

FIG. 33 is a diagram illustrating another example of the phase characteristics of the phase shifter illustrated in FIG. 29.

FIG. 34 is a circuit diagram showing another specific example of the phase shifter shown in FIG. 28.

FIG. 35 is a diagram illustrating an example of amplitude characteristics of the phase shifter illustrated in FIG. 34;

FIG. 36 is a diagram illustrating an example of a phase characteristic of the phase shifter illustrated in FIG. 34;

FIG. 37 is a diagram illustrating another example of the amplitude characteristic of the phase shifter illustrated in FIG. 34;

FIG. 38 is a diagram illustrating another example of the phase characteristics of the phase shifter illustrated in FIG. 34.

FIG. 39 is a circuit diagram showing another specific example of the phase shifter shown in FIG. 28.

40 is a diagram illustrating an example of an amplitude characteristic of the phase shifter illustrated in FIG. 39.

FIG. 41 is a diagram illustrating an example of a phase characteristic of the phase shifter illustrated in FIG. 39;

42 is a diagram illustrating another example of the amplitude characteristic of the phase shifter illustrated in FIG. 39.

FIG. 43 is a view illustrating another example of the phase characteristics of the phase shifter illustrated in FIG. 39;

FIG. 44 is a circuit diagram showing another specific example of the phase shifter shown in FIG. 28.

FIG. 45 is a diagram illustrating an example of amplitude characteristics of the phase shifter illustrated in FIG. 44;

FIG. 46 is a diagram illustrating an example of a phase characteristic of the phase shifter illustrated in FIG. 44;

FIG. 47 is a diagram illustrating another example of the amplitude characteristic of the phase shifter illustrated in FIG. 44.

FIG. 48 is a diagram illustrating another example of the phase characteristics of the phase shifter illustrated in FIG. 44.

FIG. 49 is a circuit diagram showing a specific prototype example of the phase shifter shown in FIG. 29;

50 is a plan view of the prototype example shown in FIG. 49.

FIG. 51 is a diagram showing the input reflection characteristics of the prototype example shown in FIG.

FIG. 52 is a diagram showing forward transfer characteristics of the prototype example shown in FIG. 49;

FIG. 53 is a diagram showing phase characteristics of the prototype example shown in FIG. 49;

FIG. 54 is a circuit diagram showing a configuration of an attenuator according to the present invention.

FIG. 55 is a circuit diagram showing a specific example of the attenuator shown in FIG. 54.

FIG. 56 is a diagram illustrating an example of a forward transfer characteristic of the attenuator illustrated in FIG. 55.

FIG. 57 is a diagram illustrating an example of an input reflection characteristic of the attenuator illustrated in FIG. 55;

58 is a diagram illustrating another example of the forward transfer characteristic of the attenuator illustrated in FIG. 55.

FIG. 59 is a diagram illustrating another example of the input reflection characteristics of the attenuator illustrated in FIG. 55;

FIG. 60 is a circuit diagram showing a configuration of a nonlinear signal generator according to the present invention.

FIG. 61 is a circuit diagram showing one configuration example of the nonlinear signal generator shown in FIG. 60.

FIG. 62 is a circuit diagram of a conventional phase shifter and attenuator.

FIG. 63 is a circuit diagram showing a specific example of the conventional phase shifter shown in FIG.

FIG. 64 is a diagram showing amplitude characteristics of the conventional phase shifter shown in FIG. 63;

FIG. 65 is a diagram showing phase characteristics of the conventional phase shifter shown in FIG. 63.

FIG. 66 is a circuit diagram showing a specific example of the conventional attenuator shown in FIG. 62.

67 is a diagram illustrating an example of a forward transfer characteristic of the conventional attenuator illustrated in FIG. 66.

FIG. 68 is a diagram showing an example of the input reflection characteristics of the conventional attenuator shown in FIG. 66.

FIG. 69 is a circuit diagram of a conventional nonlinear signal generator.

[Explanation of symbols]

101, 201, 301 ... input terminals, 102, 202,
302 output terminal, 103a, 103b, 203a, 2
03b, 303a, 303b ... high-frequency phase shifter, 104
a to 104c, 204a to 204c, 304a to 304
c: input / output terminals of the high-frequency phase shift element, 113a, 113
b, 123a, 123b, 213a, 213b, 313
a, 313b ... high-frequency transmission lines, 114a to 114c,
214a to 214c, 314a to 314c: input / output terminals of high-frequency transmission lines, 124a to 124c, 144a to 1
44c, 164a to 164c... Π-type circuit input / output terminals,
126a-126d, 136a, 136b, 146a-
146d, 156a to 156d, 166a, 166b ...
Capacitors 133a to 133d, 143a, 143
b, 153a, 153b, 163a to 163d, 191
a, 191b, 192a, 192b, 193a, 193
b, 194a to 194d ... inductor, 134a to 13
4c, 154a to 154c: input / output terminals of T-type circuit, 1
70a, 170b ... variable reactance elements, 171a,
171b, 181a, 181b, 182a, 182b,
183a to 183d, 184a to 184d: variable capacitors, 172a, 172b: resonance circuits, 270a, 27
0b: variable resistance element, 271a, 271b: variable resistance,
370a, 370b: Non-linear element, 371a, 371
b, 372a, 372b ... diode, 373a, 37
3b: Terminating resistor, 374a, 374b, 375a, 37
5b: DC blocking capacitor, 377: High frequency blocking inductor, 378: Bias terminal.

──────────────────────────────────────────────────の Continued on the front page (51) Int.Cl. ⁷ identification code FI H03H 7/25 H03H 7/25 (58) Investigated field (Int.Cl. ⁷ , DB name) H01P 1/18 H01P 1/22 H03F 1/32 H03H 7/18 H03H 7/20 H03H 7/25

Claims (34)

(57) [Claims] 1. A a first high-frequency impedance element having an impedance which is constituted from a connected and substantially reactance between the input terminal and the output terminal, one end connected to said input terminal and at the frequency f ₀ A first high-frequency phase shift element having a phase change of 90 degrees and having an impedance conversion function, and a phase at a frequency f ₀ connected between the output terminal and the other end of the first high-frequency phase shift element. A second high-frequency phase shift element having a change amount of 90 degrees and having an impedance conversion function, one end of which is connected to a common connection point of the first high-frequency phase shift element and the second high-frequency phase shift element; A second high-frequency impedance element having the other end grounded and having an impedance substantially constituted by reactance; and an impedance of the first high-frequency impedance element. Dance with the impedance of the second high-frequency impedance element, a phase shifter, characterized in that input and output reflection coefficient at the frequency f ₀ is set to approximately zero. 2. The phase shifter according to claim 1, wherein reactances of the first and second high-frequency impedance elements are variable. 3. The phase shifter according to claim 1, wherein each of the first and second high-frequency phase shift elements is a high-frequency transmission line having an electrical length of 90 degrees at a frequency f _0. Characterized phase shifter. 4. The phase shifter according to claim 1, wherein each of said first and second high-frequency phase shift elements has a high-frequency transmission line having an electrical length at frequency f ₀ of less than 90 degrees, and A phase shifter comprising a π-type circuit including two capacitors each having one end connected to both ends of a high-frequency transmission line and the other end grounded. 5. The phase shifter according to claim 1, wherein each of the first and second high-frequency phase shift elements is a lumped constant circuit including an inductor and a capacitor. Phase shifter. 6. The phase shifter according to claim 3, wherein the input impedance of the input terminal and the output impedance of the output terminal are respectively Z ₀ , the reactance of the first high-frequency impedance element is X ₁ , and the first high-frequency impedance element is X ₁ .
When the characteristic impedance of the high-frequency transmission line used as the second high-frequency phase shift element is Z ₂ , and the reactance of the second high-frequency impedance element is X ₃ , the reactance X ₃ becomes X ₃ = Z ₂ A phase shifter characterized by being set by a relational expression of ² X ₁ / (4Z ₀ ² ). 7. The phase shifter according to claim 4, wherein the input impedance of the input terminal and the output impedance of the output terminal are Z ₀ , the reactance of the first high-frequency impedance element is X ₁ , and the first high-frequency impedance element is X ₁ , respectively.
The electrical length and characteristic impedance of the high-frequency transmission line included in the first and second high-frequency phase shift elements are θ and Z, respectively, the capacitance of the capacitor included in the first and second high-frequency phase shift elements is C, When the reactance of the high-frequency impedance element is X ₃ , the capacitance C
And the reactance X _{3 is, C = 1 / (2πf 0} Ztanθ) X 3 = (Zsinθ) 2 X 1 / (4Z 0 2) phase shifter, characterized in that it is set by the relational expression. 8. The phase shifter according to claim 5, wherein each of the first and second high-frequency phase shift elements has a capacitor grounded at one end and one end connected to the other end of the capacitor. A T-type circuit composed of two inductors, wherein the input impedance of the input terminal and the output impedance of the output terminal are Z ₀ , the reactance of the first high-frequency impedance element is X ₁ , and the capacitance of the capacitor, respectively. Is C, the inductance of the inductor is L, and the reactance of the second high-frequency impedance element is X ₃ , the capacitance C and the reactance X ₃ are C = 1 / [(2πf ₀ ) ² L] X ₃ = (2πf ₀ L) ² X ₁ / (4Z ₀ ² ). 9. The phase shifter according to claim 5, wherein each of the first and second high-frequency phase shift elements has an inductor connected to one end of the inductor and one end connected to both ends of the inductor. Is a π-type circuit composed of two capacitors grounded, wherein the input impedance of the input terminal and the output impedance of the output terminal are Z ₀ , the reactance of the first high-frequency impedance element is X ₁ , When the capacitance of the capacitor is C, the inductance of the inductor is L, and the reactance of the second high-frequency impedance element is X ₃ , the capacitance C and the reactance X ₃ are C = 1 / [(2πf ₀ ) ² L ] A phase shifter characterized by being set by a relational expression of X ₃ = (2πf ₀ L) ² X ₁ / (4Z ₀ ² ). 10. The phase shifter according to claim 5, wherein each of the first and second high-frequency phase shift elements has an inductor grounded at one end and one end connected to the other end of the inductor. A T-type circuit comprising two capacitors, wherein the input impedance of the input terminal and the output impedance of the output terminal are Z ₀ , the reactance of the first high-frequency impedance element is X ₁ , and the capacitance of the capacitor Is C, the inductance of the inductor is L, and the reactance of the second high-frequency impedance element is X ₃ , the capacitance C and the reactance X ₃ are C = 1 / [(2πf ₀ ) ² L] X ₃ = (2πf ₀ L) ² X ₁ / (4Z ₀ ² ). 11. The phase shifter according to claim 5, wherein each of the first and second high-frequency phase shift elements has a capacitor and one end connected to both ends of the capacitor and the other end. Is a π-type circuit composed of two grounded inductors, wherein the input impedance of the input terminal and the output impedance of the output terminal are Z ₀ , the reactance of the first high-frequency impedance element is X ₁ , When the capacitance of the capacitor is C, the inductance of the inductor is L, and the reactance of the second high-frequency impedance element is X ₃ , the capacitance C and the reactance X ₃ are C = 1 / [(2πf ₀ ) ² L ] A phase shifter characterized by being set by a relational expression of X ₃ = (2πf ₀ L) ² X ₁ / (4Z ₀ ² ). 12. The phase shifter according to claim 1, wherein each of said first and second high-frequency impedance elements is a variable capacitor. 13. The phase shifter according to claim 1, wherein each of the first and second high-frequency impedance elements is a resonance circuit. 14. The phase shifter according to claim 13, wherein the resonance circuit is a series resonance circuit in which an inductor and a capacitor are connected in series. 15. The phase shifter according to claim 13, wherein the resonance circuit is a parallel resonance circuit in which an inductor and a capacitor are connected in parallel. 16. The phase shifter according to claim 13, wherein the resonance circuit is a composite resonance circuit in which a second capacitor is connected in parallel to a series resonance circuit in which an inductor and a first capacitor are connected in series. A phase shifter, characterized in that: 17. The phase shifter according to claim 13, wherein the resonance circuit is a composite resonance circuit in which two series resonance circuits in which an inductor and a capacitor are connected in series are connected in parallel. Phaser. 18. A first high-frequency impedance element connected between an input terminal and an output terminal and having an impedance substantially constituted by a resistor, one end connected to the input terminal and having a frequency f ₀ . A first high-frequency phase shift element having a phase change of 90 degrees and having an impedance conversion function, and a phase at a frequency f ₀ connected between the output terminal and the other end of the first high-frequency phase shift element. A second high-frequency phase shift element having a change amount of 90 degrees and having an impedance conversion function, one end of which is connected to a common connection point of the first high-frequency phase shift element and the second high-frequency phase shift element; A second high-frequency impedance element having the other end grounded and having an impedance substantially constituted by a resistor, wherein each of the first and second high-frequency phase shift elements has a
High-frequency transmission line whose electric length at wave number f ₀ is smaller than 90 degrees
And one end is connected to both ends of the high-frequency transmission line.
And two other ends grounded.
And a π-type circuit comprising an input terminal and an output impedance of the input terminal.
Has an output impedance of Z ₀ , and the first high frequency
The resistance of the wave impedance element is R ₁ ,
2 of the high-frequency transmission line included in the second high-frequency phase shift element.
The steepness and characteristic impedance are θ and Z, respectively.
Capacities included in the first and second high-frequency phase shift elements
A capacitor having a capacitance of C, the second high-frequency impedance element;
Is R ₃ , the capacitance C and the resistance R
_{3, C = 1 / (2πf 0} Ztanθ) R 3 = (Zsinθ) attenuator, characterized in that it is set by the relational expression ^{_{2 R 1 / (4Z 0 2}} ). 19. A terminal connected between an input terminal and an output terminal.
And has an impedance substantially composed of resistors
A first high-frequency impedance element having one end connected to the input terminal and a phase at a frequency f _0.
The amount of change is 90 degrees and has an impedance conversion function
A first high-frequency phase shift element, between the output terminal and the other end of the first high-frequency phase shift element;
And the phase change at frequency f ₀ is 90 degrees.
High frequency phase shifter having impedance conversion function
, The first high-frequency phase shift element, and the second high-frequency phase shift element
One end is connected to the common connection point with
And has an impedance substantially composed of resistors.
And a second high-frequency impedance element, wherein each of the first and second high-frequency phase shift elements
A capacitor having one end grounded and the other end of the capacitor
It consists of two inductors, one end of which is connected.
A T-type circuit is made, the input impedance and the output terminal of the input terminal
Has an output impedance of Z ₀ , and the first high frequency
The impedance of the wave impedance element is R ₁ ,
The capacitance is C, the inductance of the inductor is L,
If the resistance of the second high-frequency impedance element is a R ₃
In this case, the capacitance C and the resistance R ₃ are set by a relational expression of C = 1 / [(2πf ₀ ) ² L] R ₃ = (2πf ₀ L) ² R ₁ / (4Z ₀ ² ) An attenuator. 20. A power supply connected between an input terminal and an output terminal.
And has an impedance substantially composed of resistors
A first high-frequency impedance element having one end connected to the input terminal and a phase at a frequency f _0.
The amount of change is 90 degrees and has an impedance conversion function
A first high-frequency phase shift element, between the output terminal and the other end of the first high-frequency phase shift element;
And the phase change at frequency f ₀ is 90 degrees.
High frequency phase shifter having impedance conversion function
, The first high-frequency phase shift element, and the second high-frequency phase shift element
One end is connected to the common connection point with
And has an impedance substantially composed of resistors.
And a second high-frequency impedance element, wherein each of the first and second high-frequency phase shift elements
One end of each of the inductor and both ends of the inductor
Two keys that are connected and the other ends of which are grounded
A π-type circuit composed of a capacitor and an input impedance of the input terminal and the output terminal.
Has an output impedance of Z ₀ , and the first high frequency
The impedance of the wave impedance element is R ₁ ,
The capacitance is C, the inductance of the inductor is L,
If the resistance of the second high-frequency impedance element is a R ₃
In this case, the capacitance C and the resistance R ₃ are set by a relational expression of C = 1 / [(2πf ₀ ) ² L] R ₃ = (2πf ₀ L) ² R ₁ / (4Z ₀ ² ) An attenuator. 21. A power supply connected between an input terminal and an output terminal.
And has an impedance substantially composed of resistors
A first high-frequency impedance element having one end connected to the input terminal and a phase at a frequency f _0.
The amount of change is 90 degrees and has an impedance conversion function
A first high-frequency phase shift element, between the output terminal and the other end of the first high-frequency phase shift element;
And the phase change at frequency f ₀ is 90 degrees.
High frequency phase shifter having impedance conversion function
, The first high-frequency phase shift element, and the second high-frequency phase shift element
One end is connected to the common connection point with
And has an impedance substantially composed of resistors.
And a second high-frequency impedance element, wherein each of the first and second high-frequency phase shift elements
An inductor whose end is grounded, and the other end of the inductor
It consists of two capacitors, one end of each of which is connected.
A T-type circuit is made, the input impedance and the output terminal of the input terminal
Has an output impedance of Z ₀ , and the first high frequency
The impedance of the wave impedance element is R ₁ ,
The capacitance is C, the inductance of the inductor is L,
If the resistance of the second high-frequency impedance element is a R ₃
In this case, the capacitance C and the resistance R ₃ are set by a relational expression of C = 1 / [(2πf ₀ ) ² L] R ₃ = (2πf ₀ L) ² R ₁ / (4Z ₀ ² ) An attenuator. 22. A power supply connected between an input terminal and an output terminal.
And has an impedance substantially composed of resistors
A first high-frequency impedance element having one end connected to the input terminal and a phase at a frequency f _0.
The amount of change is 90 degrees and has an impedance conversion function
A first high-frequency phase shift element, between the output terminal and the other end of the first high-frequency phase shift element;
And the phase change at frequency f ₀ is 90 degrees.
The second high-frequency NamiUtsuri Aimoto with Ri impedance conversion function
, The first high-frequency phase shift element, and the second high-frequency phase shift element
One end is connected to the common connection point with
And has an impedance substantially composed of resistors.
And a second high-frequency impedance element, wherein each of the first and second high-frequency phase shift elements
One end of each of the capacitors and both ends of the capacitor
Connected and the other end is grounded.
A π-type circuit comprising an input terminal and an input terminal;
Has an output impedance of Z ₀ , and the first high frequency
The impedance of the wave impedance element is R ₁ ,
The capacitance is C, the inductance of the inductor is L,
If the resistance of the second high-frequency impedance element is a R ₃
In this case, the capacitance C and the resistance R ₃ are set by a relational expression of C = 1 / [(2πf ₀ ) ² L] R ₃ = (2πf ₀ L) ² R ₁ / (4Z ₀ ² ) An attenuator. 23. The attenuator according to claim 18 , wherein said first and second high-frequency impedance elements are provided.
An attenuator characterized in that each resistance is variable . 24. A power supply connected between an input terminal and an output terminal.
Having an impedance including a resistance component and being input
A first non-linear signal for generating a non-linear signal in accordance with the applied signal power
Element and one end connected to the input terminal and the phase at frequency f ₀
The amount of change is 90 degrees and has an impedance conversion function
A first high-frequency phase shift element, between the output terminal and the other end of the first high-frequency phase shift element;
And the phase change at frequency f ₀ is 90 degrees.
High frequency phase shifter having impedance conversion function
, The first high-frequency phase shift element, and the second high-frequency phase shift element
One end is connected to the common connection point with
Having an impedance including a resistance component and input
The same non-linear element as that of the first nonlinear element according to the signal power obtained.
A second nonlinear element for generating a linear signal, wherein a resistance component of an impedance of the first nonlinear element and
The resistance component of the impedance of the second nonlinear element is
Set so that the input / output reflection coefficient at wave number f ₀ is almost zero
A non-linear signal generator characterized in that: 25. The nonlinear signal generator according to claim 24,
Oite, wherein each of the first and second high-frequency phase shifting elements, the circumferential
This electrical length at the wavenumber f ₀ is 90 degrees of the high-frequency transmission line
And a non-linear signal generator. 26. The nonlinear signal generator according to claim 24, wherein
Oite, wherein each of the first and second high-frequency phase shifting elements, the circumferential
High-frequency transmission line whose electric length at wave number f ₀ is smaller than 90 degrees
And one end is connected to both ends of the high-frequency transmission line.
And two other ends grounded.
It is characterized by being a π-type circuit composed of
Non-linear signal generator. 27. The nonlinear signal generator according to claim 24,
Here, each of the first and second high-frequency phase shift elements is
Lumped constant circuit composed of inductor and capacitor
A non-linear signal generator, characterized in that: 28. The nonlinear signal generator according to claim 25,
And Input impedance of the input terminal and the output terminal
Output impedance is Z ₀ , The first non-linear
The resistance component of the element is R ₁ The first and second high frequencies
Characteristics of the high-frequency transmission line used as a phase shift element
Impedance is Z _Two , The resistance component of the second nonlinear element
Is R _Three , The resistance component R ₁ , R _Three Is R ₁  = 2Z ₀ R _Three  = Z _Two ^Two R ₁ / (4Z ₀ ^Two ) Characterized by being set by the relational expression
Signal generator. 29. The nonlinear signal generator according to claim 26,
Oite, input impedance and the output terminal of the input terminal
Has an output impedance of Z ₀ , and the first nonlinear
The resistance component of the element is R ₁ , and the first and second high-frequency components are
Electrical length of the high-frequency transmission line included in the phase shift element and
The characteristic impedances are θ and Z, respectively,
And the capacitance of the capacitor included in the second high-frequency phase shift element
But C, the resistance component of the second nonlinear element when is R ₃
In this case, the capacitance C and the resistance components R ₁ and R ₃ are set by a relational expression of C = 1 / (2πf ₀ Ztan θ) R ₁ = 2Z ₀ R ₃ = ( Z sin θ) ² R ₁ / (4Z ₀ ² ) A non-linear signal generator characterized in that: 30. A non-linear signal generator of claim 27, wherein each of the first and second high-frequency phase shifting elements, one
A capacitor having one end grounded and the other end of the capacitor
It consists of two inductors, one end of which is connected.
A T-type circuit is made, the input impedance and the output terminal of the input terminal
Has an output impedance of Z ₀ , and the first nonlinear
The resistance component of the element is R ₁ , the capacitance of the capacitor is C,
The inductance of the inductor is L, the second nonlinearity;
When the resistance component of the element is R ₃ ,
The resistance components R ₁ and R ₃ are set by a relational expression of C = 1 / [(2πf ₀ ) ² L] R ₁ = 2Z ₀ R ₃ = (2πf ₀ L) ² R ₁ / (4Z ₀ ² ) nonlinear signal generator, characterized by that. 31. A non-linear signal generator of claim 27, wherein each of the first and second high-frequency phase shifting elements, Lee
And inductor, Ri π type circuit der composed of two capacitors, each of the other end of which is grounded with each one at both ends of the inductor is connected, the input impedance and the output terminal of the input terminal
Has an output impedance of Z ₀ , and the first nonlinear
The resistance component of the element is R ₁ , the capacitance of the capacitor is C,
The inductance of the inductor is L, the second non-linear
When the resistance component of the element is R ₃ ,
The resistance components R ₁ and R ₃ are set by a relational expression of C = 1 / [(2πf ₀ ) ² L] R ₁ = 2Z ₀ R ₃ = (2πf ₀ L) ² R ₁ / (4Z ₀ ² ) nonlinear signal generator, characterized by that. 32. A non-linear signal generator of claim 27, wherein each of the first and second high-frequency phase shifting elements, one
An inductor whose end is grounded, and the other end of the inductor
It consists of two capacitors, one end of each of which is connected.
A T-type circuit is made, the input impedance and the output terminal of the input terminal
Has an output impedance of Z ₀ , and the first nonlinear
The resistance component of the element is R ₁ , the capacitance of the capacitor is C,
The inductance of the inductor is L, the second nonlinearity;
When the resistance component of the element is R ₃ ,
The resistance components R ₁ and R ₃ are set by a relational expression of C = 1 / [(2πf ₀ ) ² L] R ₁ = 2Z ₀ R ₃ = (2πf ₀ L) ² R ₁ / (4Z ₀ ² ) nonlinear signal generator, characterized by that. 33. The nonlinear signal generator according to claim 27 , wherein each of said first and second high-frequency phase shift elements comprises a key.
One end of each of the capacitors and both ends of the capacitor
Connected and the other end is grounded.
A π-type circuit composed of the inductor, the output impedance of the input impedance and the output terminal of said input terminals, respectively Z _0, the capacitance of the resistive component of the first nonlinear element is R _1, the capacitor C,
When the inductance of the inductor is L 1 and the resistance component of the second nonlinear element is R ₃ , the capacitance C and the resistance components R ₁ and R ₃ are represented by C = 1 / [(2πf ₀ ) ² L] R ₁ = 2Z ₀ R ₃ = (2πf ₀ L) ² R ₁ / (4Z ₀ ² ) A nonlinear signal generator characterized by the following equation: 34. The nonlinear signal generator according to claim 24 , wherein each of said first and second nonlinear elements has opposite polarity.
And two diodes connected in parallel to
And a resistor connected in parallel with the diode.
A non-linear signal generator characterized in that a bias current flows in each of them.