JP3298508B2 - Motor drive control device - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

The present invention relates to a drive control device for a motor having a plurality of excitation phases, such as a brushless motor and a linear motor.

[0002]

2. Description of the Related Art For example, a brushless motor used as a drive source of a power steering device of an automobile is described below.
It is a motor having three or more exciting phases, and its driving is performed by a rectangular wave exciting current.

[0003] In the case of a five-phase brushless motor, the motor drive circuit uses an electrical angle to describe the outer peripheral surface of the rotor of the motor.
Under control of a control circuit such as a microcomputer, four phases are simultaneously applied to five-phase (hereinafter, referred to as a-phase to e-phase) excitation coils a to e arranged so as to be separated from each other by 72 degrees. The coils are sequentially switched one phase at a time by a four-phase excitation method in which excitation is performed, and the coils are excited by a rectangular wave current to drive the rotor to rotate. In this four-phase excitation method, the motor current flows in four of the five phases. However, in order to allow the current to flow in each phase in a well-balanced manner, the resistance of each excitation coil is formed to be all equal. I have.

[0004] Such a motor drive circuit is composed of ten field effect transistors (FETs). These ten transistors are connected in series with two corresponding transistors to form five series transistor circuits, each of which is connected between the + and-terminals of the power supply and the two transistors of each series transistor circuit are connected. By connecting the connection portions of the transistors to the outer ends of five exciting coils a to e star-connected in a Y-shape, the connection portions are connected to the coil circuit of the motor.

The direction and length of the exciting current (rectangular wave) supplied from the motor drive circuit to each exciting coil are, for example, as shown in FIG. 16 with respect to the value of the rotation angle (electrical angle) of the rotor. . That is, the excitation coil is switched one phase at a time every 36 degrees in electrical angle, and one phase coil is switched in electrical angle in 144 degrees.
By exciting for a while, the rotor is continuously rotated. In this figure, when the electrical angle is θ, 0 ≦ θ <36, 36 ≦ θ <72, 72 ≦ θ <108, 108 ≦ θ
<144, 144 ≦ θ <180, 180 ≦ θ <216, 216 ≦ θ <2
52, 252 ≦ θ <288, 288 ≦ θ <324, 324 ≦ θ <360
Are represented by (1),..., (10), respectively.

In the case of this example, the current of the a-phase flows in the + direction in the sections (1) and (2), 0 in the section (3), and from the section (4).
At (7), it flows in the negative direction, at section (8), 0, at section (9) to (10)
Flows again in section (1) in the + direction. The b-phase current is
Flow in the + direction from section (1) to (4), 0 in section (5), section
From (6) to (9), it flows in the minus direction, at section (10) it goes to 0, and again it goes to section (1) in the plus direction. The current of the c-phase is in section (1)
Flows in the minus direction, 0 in section (2) and + in sections (3) to (6).
It flows in the direction of 0, flows in the section (7) through 0, flows through the sections (8) through (10) again in the section (1) in the negative direction. The current of the d-phase is the interval (1)
From (3) to-direction, 0 in section (4), and from section (5)
It flows in the + direction at (8), 0 in the section (9), and flows again in the-direction from the section (10). The e-phase current is 0 in section (1),
Flow in the-direction from section (2) to (5), 0 in section (6), section
It flows in the + direction from (7) to (10), and becomes 0 again in the section (1). Therefore, the boundary of each section from (1) to (10) (36 electrical degrees)
At the time of switching every degree), two of the five excitation coils are switched in opposite directions.

Such switching of the exciting current is represented in principle by the rise or fall of a rectangular wave as shown in FIG. 16, but the rise or fall waveform is actually relative to the horizontal axis. It does not change at right angles,
It takes a certain amount of time Δt (about three times the time constant of the motor circuit) until the exciting current rises in the + direction or falls in the − direction.

For example, at the boundary between the sections (8) and (9) in FIG. 16 (288 electrical degrees), the current of the a-phase rises from 0 to a constant value of +, while the current of the d-phase rises to a constant value of +. 0 from the value
The currents of the b-phase and the c-phase both have a constant value of "-", and the current of the e-phase has a constant value of "+". Become like

More specifically, the rising current of the a-phase gradually increases from 0 to a fixed value of + during the time Δt, while the falling current of the d-phase increases for a time Δt1 (motor (Smaller than the time constant of the circuit), and decreases from a constant value of + to 0. At this time, the other three phases b, c, e
Are the phases that cannot be switched, but the currents of the five phases are i _a ,
When represented by i _b , i _c , i _d , and i _e , these currents have the following relationship.

[0010] _{i a + i d + i e} = - (i b + i c) = I ... (1) Consequently, the current of a phase and d-phase changes as described above, b, c, also the current e-phase Change. That is, since the current change rates of the a phase and the d phase are different, the total value of the currents of the two phases does not become a steady value, and the currents of the b phase and the c phase fluctuate as shown in FIG. The phase current also changes during the time Δt. These current fluctuations cause excessive torque fluctuations.

The reason why the rising and falling current change rates of the two phase currents are different as described above is as follows.

First, it is assumed that the power supply voltage supplied to the motor drive circuit is Vb, and the voltage at the center connection point of the star-connected exciting coils a to e is Vn.

Next, in FIG. 17, a section of time Δt1 is defined as a section of time Δt2 (= Δt−Δt1).

In the section, d is switched from + to 0.
Phase current i _d of (OFF phase), -Vn, the change rate according to the time constant of the counter-electromotive voltage E _d and the motor circuit of the coil, half of the electric current I from the motor driving circuit to the motor (I / 2 )
To zero (0). At this time, if the voltage applied to the OFF-phase equivalent circuit is V _OFF , V _OFF = −Vn
−E _d <0, and Vn is approximately Vb / 2. On the other hand, the current i of the a-phase (ON phase) that can be switched from 0 to +
_a is the voltage Vb, -Vn, voltage change rate according to the time constant of the counter-electromotive voltage E _a and the motor circuit of the coil, rises from zero (0), applied to this case, the equivalent circuit of the ON phase the When V _ON, (duty ratio of the rectangular wave) V _ON = Vb · Duty1 a -Vn -E _a.

Describing the equation, the current _id is expressed by the following equation using an OFF-phase equivalent circuit.

[0016] _{i d (t) = (I} / 2) e -t / T + (V OFF / R) (1-e -t / T) ... (2) when ∴t = 0, i _d = I / 2 where T is the electrical time constant of the equivalent circuit, and R is the resistance of the equivalent circuit.

On the other hand, the current i _a
Is represented by the following equation.

[0018] _{i a (t) = (V} ON / R) (1-e -t / T) ... (3) when ∴t = 0, with _{i a = 0, t → ∞} , i a = V ON / R = I / 2 Therefore, OFF phase, the currents i _d in the oN phase, the rate of change of i _a, respectively as follows.

Di _d (t) / dt = − (1 / T) (I / 2) e− ^{t / T} + (1 / T) (V _OFF / R) e− ^{t / T} = − (I / 2 −V _OFF / R) (1 / T) e ^{−t / T} = − (I / 2 + Vn / R + E _d / R) (1 / T) e ^{−t / T} (4) di _a (t) / dt = (1 / T) (V _ON / R) e ^{−t / T} = (I / 2) (1 / T) e ^{−t / T} (5) In the above equations (4) and (5), (I / _{2 + Vn / R + E d} / R)>
Since it is I / 2, the current change rate of the OFF phase is larger than the current change rate of the ON phase. In particular, when the resistance R of the equivalent circuit is small, when the power supply voltage Vb (≒ 2Vn) is large, or at the time of high-speed rotation, the back electromotive force E _d (= (1/2) Km · ω, Km [volt · se
c] is the motor voltage constant, and ω is the motor rotation angular velocity). When the motor rotation angle is large, the OFF-phase current change rate is considerably larger than the ON-phase current change rate. That is, the resistor R and the power
Voltage Vb, or motor rotation angular velocity ω or motor voltage
The magnitude of several km is the current change rate of OFF phase and ON phase.
Affect the difference.

[0020] Therefore, than the time which the current i _d of OFF-phase is lowered from the I / 2 to 0 (Δt1), the longer the time the current i _a of the ON phase is increased from 0 to I / 2 (Δt). That is, the current i _a of last ON phase of Interval will not reach the I / 2, is in the middle rises.

Thereafter, in the section, the ON-phase current i
_{Although a} finally reaches the steady value (I / 2), it takes a time Δt2 (two to three times the time constant of the motor circuit) by that time. Therefore, the rising and falling currents of the two phases to be switched have different current change rates.

[0022]

As described above, in the control of the exciting current by the conventional motor drive circuit, two
Since the rates of rise and fall of the currents of the three phases (for example, the phase a and the phase d in FIG. 16) are different, the currents of the phases that cannot be switched (for example, the phases b, c, and e in FIG. 16) Fluctuate, and an excessive torque fluctuation occurs due to these current fluctuations.

In order to suppress the current fluctuation at the time of phase switching that causes such a torque fluctuation, the current of each phase may be controlled, but it is necessary to detect the current of each phase for the control. And two or more current detection circuits are required. In particular,
In the case of a five-phase brushless motor, since the four-phase excitation method is adopted, four current detection circuits and four current loops are required in the motor drive circuit, which complicates the configuration of the drive circuit and increases the cost. There was a problem of becoming.

The present invention has been made in view of the above circumstances, and provides a motor drive control device that does not use two or more current detection circuits and has a simple circuit configuration and can suppress the current fluctuation that causes the above-described torque fluctuation. It is in.

[0025]

SUMMARY OF THE INVENTION The present invention is an apparatus for controlling the driving of a motor having a plurality of excitation phases, the driving means generating an excitation current to be supplied to each excitation phase of the motor;
The drive is performed so that the excitation current is switched for each excitation phase.
Ru and control means for controlling the means.

[0026] The control means includes a drive signal for making the current change rates of the exciting phase in which the exciting current rises and the exciting phase in which the exciting current rises at the time of switching of the exciting current coincide with each other or about the same.
(G _1-10 ) is generated and supplied to the driving means .

The drive signal is obtained by synthesizing a first PWM signal for an exciting phase in which the exciting current is not switched, and a second PWM signal for an exciting phase in which the exciting current rises and / or an exciting phase in which the exciting current falls. Therefore, it is generated .

In a specific embodiment of the present invention , the driving means
Generates excitation current to be supplied to multiple excitation coils of the motor.
And a duty ratio (Duty2) of the second PWM signal, a motor current value (I) at the time of the switching, and a duty ratio (Duty2) of the first PWM signal.
1) The rotational angular velocity (ω) and voltage constant (Km) of the motor, the power supply voltage (Vb) supplied to the drive circuit, and the resistance (R) of the equivalent circuit including the excitation coil circuit and the drive circuit of the motor. Function.

[0029]

[Action and Effect] control means performs switching from the drive means of the excitation current supplied to each excitation phase of the motor. Excitation phase in which the excitation current rises when switching, and excitation in which the excitation current falls
Drive to make the current change rate of the magnetic phase equal or comparable
A signal is generated and supplied to the driving means .

The drive signal is supplied to the control means for
A first PWM signal for the excitation phase whose current is not switched
Signal and the excitation phase and / or fall
And a second PWM signal for the excitation phase
Is generated by

According to the present invention, as described above,
Switching by controlling the rate of change of the exciting current
The current change rates of the two phases must be equal or similar.
Door is Ru can. As a result, current fluctuations in the non-switching phase are suppressed, so that the aforementioned torque fluctuations are also eliminated .

The excitation current is supplied to the motor circuit as in the conventional case.
It can be generated by detecting only the flowing current.
It is necessary to detect the current of each phase to control the rate of change of the current.
It is unnecessary and the circuit configuration for control is not complicated .

In a specific embodiment of the present invention , the second P
The duty ratio (Duty2) of the WM signal includes the motor current value (I) at the time of the switching, the duty ratio (Duty1) of the first PWM signal, the rotational angular velocity (ω) of the motor, and the voltage constant (K
m), the power supply voltage (Vb) supplied to the drive circuit, and the resistance (R) of an equivalent circuit including the excitation coil circuit and the drive circuit of the motor.
A second PWM signal is automatically generated together with the above PWM signal, and a drive signal for controlling the current change rate of the phase in which the exciting current is switched can be easily obtained by a combination operation of the two signals.

If the present invention is used for drive control of a brushless motor, a high-performance motor control device that is inexpensive and has reduced current fluctuation can be obtained. For example, by incorporating the motor drive control device of the present invention into the above-described electric power steering device, it is possible to realize a power steering having no unpleasant torque fluctuation and excellent steering feeling.

[0035]

FIG. 1 is a longitudinal sectional view showing the internal structure of a five-phase brushless motor as an example of a motor driven and controlled by the apparatus of the present invention.

The five-phase brushless motor 1 has a cylindrical housing 2, a rotating shaft 4 arranged along the axis of the housing 2 and rotatably supported by bearings 3a and 3b, and a rotating shaft. 4, a motor driving permanent magnet 5 fixed to the inner peripheral surface of the housing 2 so as to surround the permanent magnet 5, and five-phase excitation coils 6a, 6
A stator (stator) 6 around which b, 6c, 6d and 6e are wound, and a rotor (rotor) 7 is constituted by the rotating shaft 4 and the permanent magnet 5.

In the vicinity of one end of the rotating shaft 4 of the rotor 7,
A ring-shaped permanent magnet 8 for phase detection is fixed, and the permanent magnets 8 are alternately magnetized to S and N poles at equal intervals in the circumferential direction. Further, the permanent magnet 5 of the rotor 7 also has the S pole and the N pole.
The poles are magnetized alternately at equal intervals in the circumferential direction.

A support substrate 10 made of a ring-shaped thin plate is provided on the end surface of the housing 2 on the side where the bearing 3b is provided, with a stay 9 interposed therebetween. It is arranged as follows. This support substrate 10
Is fixed to the surface on the side of the permanent magnet 8 so as to face the permanent magnet 8. Incidentally, in practice, five (11a to 11e) phase detection elements 11 are provided at appropriate intervals in the circumferential direction corresponding to the drive timings of the excitation coils 6a to 6e, but FIG. Only one is shown.

The phase detecting elements 11a to 11e output a sensor signal of "H" as a position detection signal when the magnetic poles of the opposing permanent magnets 8 are N poles and "L" when the magnetic poles are S poles. Are output. The rotation position of the rotor 7 can be detected by utilizing the fact that the output of each of the phase detection elements 11a to 11e changes according to the magnetic pole of the permanent magnet 8 facing each element. Depending on its rotational position,
The motor drive control device 20 (FIG. 2), which will be described later, uses the four-phase excitation method in which the four-phase excitation coils 6a to 6e are energized simultaneously and the energized excitation coils are sequentially switched one by one. Is driven to rotate.

On the other hand, the five-phase excitation coils 6a to 6e are arranged so as to surround the outer peripheral surface of the rotor 7 at an electrical angle of 72 degrees, and as shown in FIG. Thus, a coil circuit 12 of the motor is formed. In the above four-phase excitation method, the motor current flows through four phases. However, since the current is inversely proportional to the coil resistance, the coil resistance of each of the excitation coils 6a to 6e must be increased in order to flow the current in each phase in a well-balanced manner. Are formed to be all equal.

The stator 6 has, for example, 30 slots at equal intervals on the inner peripheral surface of a stator core (not shown), and has the same number of protrusions between these slots. As one set, each exciting coil 6a to 6e is wound around each set. One end of each of the excitation coils 6a to 6e is connected together and the other end is connected to the motor drive control device 20.

As shown in FIG. 2, the motor drive control device 20 includes a control circuit 21, an FET gate drive circuit 22, a motor drive circuit 23, a current detection circuit 24, and a rotor position detection circuit 25. Here, the control circuit 21 corresponds to the control means in the present invention, and the FET gate drive circuit 22 and the motor drive circuit 23 correspond to the drive means.

The control circuit 21 is constituted by a microcomputer, for example, and a constant voltage is supplied from a constant voltage source 26. The control circuit 21 receives a current command I from the external circuit 27.
_ref is input, the motor current detection value I from the current detection circuit 24, and the rotor position signal S from the rotor position detection circuit 25.
_ae (= S _a ,..., S _e ) are input. The control circuit 21 controls a drive current supplied from the motor drive circuit 23 to the coil circuit 12 of the motor based on these input signals.

Here, when the five-phase brushless motor is used as a drive source of the electric power steering apparatus, the external circuit 27 generates a pulse signal corresponding to the rotation speed of the output shaft of the transmission of the vehicle. Refer to the characteristic diagram from the vehicle speed detection value V obtained from the output of the sensor and the detection value T including the direction of the torque obtained from the output of the torque sensor for detecting the steering torque applied to the input shaft of the steering wheel. Then, a corresponding motor current value is searched, and this is output as a current command signal _Iref . This can be configured by a circuit such as a CPU that executes the above operation. Instead of this circuit, each output of the vehicle speed sensor and the torque sensor is input to the control circuit 21 and the current command _Iref is generated here. May be configured.

The motor drive circuit 23 includes five power supply circuits on the power supply side (upper side) and five motor power supply circuits on the ground side (lower side).
0 transistors (field effect transistor FET) T
a1 to Te1 and Ta2 to Te2. As for these ten transistors Ta1 to Te1, Ta2 to Te2, corresponding transistors are connected in series on the upper stage and the lower stage, and a pair of these serially connected transistors (Ta1-Ta2, Tb1-Tb2,
Tc1-Tc2, Td1-Td2, Te1-Te2), the upper terminal is the control circuit 21 and the lower terminal is the current detection circuit 24.
And the connection portion of each transistor pair is connected to the outer end of each of the exciting coils 6a to 6e (the side opposite to the center side of the star connection). Each of the gate voltage of the transistor Ta1~Te2, based on the detection signal S _ae from the rotor position detection circuit 25, the control circuit 2
1 is controlled.

The direction and magnitude of the exciting current from the motor drive circuit 23 to each of the exciting coils 6a to 6e are basically as shown in FIG. The ON / OFF timing is as shown in the gate signals (upper row) Ga1 to Ge1 and the gate signals (lower row) Ga2 to Ge2 in Table 1 below. In Table 1, gate signals Ga1 to Ge2 for turning on and off the transistors Ta1 to Te2 are represented by "1" and "0", respectively.

[0047]

[Table 1] Assuming that the rotor 7 is in the state of (1) in FIG. 16, for example , this corresponds to the section (1) in Table 1, and the upper-stage transistors Ta1 and Tb1 and the lower-stage transistors Tc2 and Td2
Are on, and the other transistors are off, so that current flows from the outer ends to the exciting coils 6a and 6b, and current flows from the connection side to the exciting coils 6c and 6d. As a result, a magnetic attraction force and a repulsive force are generated between the N pole or S pole of the rotor 7 and the N pole or S pole generated around the rotor, and the rotor 7 rotates. Then, when the rotor 7 shifts to the state of (2) in FIG.
And the upper transistors Ta1 and Tb1 and the lower transistors Td2 and Te2 are on and the other transistors are off, so that the exciting coils 6a and 6b
b, a current flows from the outer end side, and the exciting coils 6d and 6e
, A current flows from the connection side. Thereby, the rotor 7 further rotates.

When the above operation is repeated, each transistor is driven at the timings shown in Table 1, and as shown in FIG. 16, the excitation coil is switched one phase at a time every 36 degrees in electrical angle to switch one phase. Excited for 144 degrees in electrical angle. Thereby, the N pole or the S pole generated in the stator 6 moves sequentially, and the rotor 7 rotates continuously.

The current detection circuit 24 has a current detection resistor connected to the transistors Ta2 to Te2 on the lower stage of the motor drive circuit 23, amplifies the voltage generated at both ends thereof, removes noise and removes the motor current. It is output as a detection signal of value I.

The rotor position detection circuit 25 outputs the detection signals from the phase detection elements 11a to _11e as a rotor position detection signal S _ae .

As shown in Table 1 above, the control circuit 21
The storage unit stores a gate setting table indicating the correspondence between a preset combination of the detection signal S _ae from the rotor position detection circuit 25 and the upper and lower gate signals.

In this gate setting table, a combination of the detection signals S _ae corresponding to each of the sections (1) to (10) for each electrical angle of 36 degrees in FIG. 16 and the excitation coil to be set in each section are specified. Correspondence with the upper and lower gate signals Ga1 to Ge2 is set. Here, “H” of the detection signal S _ae indicates that the N pole is excited, and “L” of the detection signal S _ae indicates that the S pole is excited.

Further, in the case of the combination corresponding to each section (1) to (10), the abnormal signal is set to "0", that is, normal, and the combination is not the combination corresponding to each section (1) to (10). , The abnormal signal is set to “1”, that is, abnormal. If the combination does not correspond to each of the sections (1) to (10), all the gate signals Ga1 to Ge2 are set to "0".
Is set so that the current supply to the coil circuit 12 is not performed.

The control circuit 21 _{responds to} a combination of the detection signal S _ae sent from the rotor position detection circuit 25 with
Based on the gate setting table, corresponding gate signals Ga1 to Ge2 are sent to the FET gate drive circuit 22. Further, the control circuit 21 generates a motor drive voltage command signal by current control based on the above-described input signal, and performs pulse width modulation (Pulse Width Mo) based on the voltage command signal.
dulation) signal and gate drive signal G _1-10 (= Ga 1 to G
e2) is generated and supplied to the FET gate drive circuit 22.

The FET gate drive circuit 22 includes the control circuit 2
A predetermined voltage is supplied to the gate terminal of the specified transistor on the basis of the gate drive signal G _1-10 output from 1.

FIG. 3 is a functional block diagram of the control circuit 21. As shown in the figure, the control circuit 21 includes a current control unit 31, a PWM duty ratio calculation unit 32 for an excitation phase in which an excitation current rises (or falls) as described below, and a motor rotation speed calculation unit in terms of its function. 33, and an FET gate drive signal operation unit 34.

In the control circuit 21, the motor current command signal I _ref and the motor current value I detected by the current detection circuit 24 are input to the current control section 31.
The current control unit 31 detects the direction in which the torque is generated based on whether or not the detected torque value T from the torque sensor is higher than a predetermined neutral voltage V _C , performs a predetermined process,
Electromagnetic torque direction command signal DRC that determines motor rotation direction
Output T. Alternatively, the motor current detection value I may be provided with a sign, the electromagnetic torque direction command signal DRCT may be determined by the sign of the motor drive voltage command signal generated by the current control unit 31, and may be output. .

As described later, the current control unit 31 controls the first phase for a phase other than the phase whose current change rate is controlled at the time of phase switching.
The duty ratio Duty1 of the PWM signal is calculated and output.

On the other hand, the motor rotation speed calculation unit 33 generates a phase switching signal from the output signal S _ae from the rotor position detection circuit 25, and detects the motor rotation angular speed ω from the generation frequency of the switching signal.

The duty ratio Dut of the first PWM signal
y1, the motor current detection value I, and the motor rotation speed ω are input to a rising (or falling) phase PWM duty ratio calculation unit 32 whose current change rate is controlled at the time of phase switching. The calculation unit 32 calculates the duty ratio Duty2 of the second PWM signal for the phase whose current change rate is controlled as described later.

The above two duty ratios Duty1, Duty2,
Each signal of the electromagnetic torque direction command DRCT and the rotor position S _ae is input to the FET gate drive signal calculation unit 34.
The arithmetic unit 34 outputs a gate drive (ON / OFF) signal G _1-10 for each FET.

The FET gate drive signal calculation section 34
As shown in FIG. 4, a first FET gate drive signal logical operation unit 341 and a second FET gate drive signal logical operation unit 3
42 and an FET gate drive signal synthesis operation unit 343.

In FIG. 4, the first FET gate drive signal logic operation unit 341 determines the upper and lower stages of each phase based on the duty ratio Duty1, the position detection signal S _ae and the electromagnetic torque direction command signal DRCT of the first PWM signal. A gate drive signal G ′ _1-10 (= G ′ ₁ ,..., G ′ ₁₀ ) for the FET is generated.

Second FET gate drive signal logical operation unit 3
42 is a duty ratio Duty2 of the second PWM signal,
Rotor position detection signal S _ae and electromagnetic torque direction command DRCT
More, it generates a gate drive signal G _PC of the upper and lower FET to the exciting current rises (or falls) phase.

The FET gate drive signal synthesis operation section 343
Generates the FET gate drive signal G _1-10 from the two energized section signals G ′ _1-10 and G _PC .

FIG. 5 shows an operation block constituting the first FET gate drive signal logic operation unit 341. The arithmetic unit calculates a conventional FET energizing section signal G "from the rotor position signal S _ae and the direction command signal DRCT of the electromagnetic torque.
An arithmetic block 341a for generating _1-10 and its signal G "
From _1-10 and the duty ratio of the first PWM signal Duty1, showing the calculation block 341b to generate an FET gate drive signal G _'1-10 to the exciting phase is not controlled the rate of current change.

FIG. 6 shows an operation block constituting the second FET gate drive signal logic operation unit 342 of FIG. FIG. 6A shows an operation block 342a for generating a falling (or rising) phase FET gate drive signal G ′ _PC from the rotor position signal S _ae and the direction command signal DRCT of the electromagnetic torque, and its gate drive. Signal G ' _PC and second
From the duty ratio Duty2 of the PWM signal shows the calculation block 342b to generate a gate drive signal G _PC of FET of phase to be switched at the time of switching.

Since the end of the energizing section is when the falling of this phase starts, as shown in FIG. 6B, the operation block 342a calculates the rotor position signal S _ae and the direction command signal of the electromagnetic torque. The drive signal G ′ _PC for the phase in which the exciting current is switched may be generated using the gate drive signal G ″ _1-10 of the FET obtained from DRCT. In this case, the calculation amount is smaller than that of (A). Can be reduced.

The control circuit 21 supplies the motor current I, the duty ratio Duty1 of the first PWM signal, the motor rotation angular velocity ω and the voltage constant Km, and the motor drive circuit 23 as described above.
The supplied power supply voltage Vb and the motor coil circuit 12
The duty ratio Duty2 of the second PWM signal is calculated from the six signals of the resistance R of the equivalent circuit including the motor drive circuit 23 . The arithmetic expression is represented by the following function f.

Duty2 = f (I, Duty1, ω , Km, Vb, R ) (6) In this function f, the current change rates of two phases (for example, a phase and d phase) for switching the excitation current coincide. Or are set to be approximately the same.

Next, an example of the function f will be described.

The power supply voltage supplied to the motor drive circuit 23 in FIG. 2 is Vb, the voltage at the center connection point (the junction of each phase) of the excitation coils a to e is Vn, and Vn = (1/2) Vb. Assume. Then, the duty ratio of the PWM signal to the d phase is
Duty _2-1 , the duty ratio of the PWM signal to the a phase is D
If uty _2-2 , the voltage equation of each phase is as follows.

[0073] a _{phase: (2 · Duty 2-2 -1)} · 0.5Vb = L m (di a / dt) + i a R a + E a ... (7) b phase: (2 · Duty1-1) · 0.5 _{Vb = L m (di b /} dt) + i b R b + E b ... (8) c -phase: (2 · Duty1-1) · 0.5Vb = L m (di c / dt) + i c R c + E c ... ( 9) d _{phase: (2 · Duty 2-1 -1)} · 0.5Vb = L m (di d / dt) + i d R d + E d ... (10) e phase: (2 · Duty1-1) · 0.5Vb _{= L m (di e / dt} ) + i e R e + E e ... (11) _{However, L m = L-M (} L each phase of self-inductance,
M is the mutual inductance) between the plurality of phases.

With the rectangular wave current drive, the magnetized waveform has an electrical angle of 144
Since it is a substantially trapezoidal wave of degree, the back electromotive force is also approximately a trapezoidal wave. At the time of phase switching, the absolute value of the back electromotive voltage of each phase is substantially equal. _{That, E a = -E b = -E} c = E d = E e = E ... (12) Further, each phase of the coil resistance is the same. That is, R _a = R _b = R _c = R _d = R _e = R (13) In order to make the current change rates of the two phases to be switched (a phase and d phase in this case) the same, Are constant, and the currents of the other phases do not change. _{That, i a + i d = -i} b = -i c = i e = i ... (14) ∴ d (i a + i d) / dt = -d (i b) / dt = -d (i c) / dt = −d (i _e ) / dt = d (i) / dt = 0 (15) Equation (12) is obtained by adding the voltage equations (7) and (10) of the a-phase and d-phase.
Substituting ~ a _{(15), (2Duty 2-2 +} 2Duty 2-1 -2) · 0.5Vb = L m (d (i a + i d) / dt) + i a R a + i d R d + E _a + E _d ∴ becomes _{(Duty 2-2 + Duty 2-1 -1)} Vb = iR + 2E ... (16). Equations (12) to (1) are added to the e-phase voltage equation (11).
Substituting 5), and _{(2Duty1-1) · 0.5Vb = L m} (di e / dt) + i e R e + E e = iR + E ... (17).

From these two equations (16) and (17), the duty ratio Duty _2-1 of the OFF phase and the duty ratio Duty of the ON phase are obtained.
The relationship with _2-2 is obtained as follows.

When erasing iR Duty _2-1 + Duty _2-2 = Duty 1 + 0.5 + E / Vb = Duty 1 + 0.5 + K _m · ω / 2 Vb (18) where E = (1/2) K _m · ω , K _m [volt · sec] are voltage constants of the motor.

In the case of erasing E, Duty _2-1 + Duty _2-2 = 2 · Duty1−iR / Vb = 2 · Duty1− (2i) R / 2Vb (19) where i is a one-phase current, When detecting with one current detector, the detected current is 2i.

Using one of the above two equations (18) and (19), switching is performed as in the following Examples 1-3 (O
The duty ratios Duty _2-1 and Duty _2-2 of the second PWM signal for the (N, OFF) phase can be obtained. And
By controlling the drive current in the falling phase (or the rising phase) at the time of phase switching with the PWM signal having these duty ratios, it is possible to make the current change rates of the two phases to be switched equal or similar. In addition, the current fluctuation (FIG. 17) at the time of the conventional phase switching is suppressed. At this time, the detected current is only the motor current I, and the necessary current detection circuit is 1
It only needs one.

[Example 1] When the excitation current is switched, the current change rate of the falling phase (for example, d phase) is controlled so as to match the current change rate of the PWM signal of the rising phase (for example, a phase).

In this case, since the duty ratio Duty _2-2 = Duty1 of the PWM signal for the a-phase,
The duty ratio Duty _2-1 of the WM signal is expressed as Duty _2-1 = 0.5 + K _m · ω / 2Vb (20) from Expression (18), or Duty _2-1 = Duty1- (2i) R from Expression (19). /2Vb...(21)

FIG. 7A shows the back electromotive force E _ae (= E
_a ,..., E _e ), and FIG. 7B shows the gate drive signal G ′ _PC for controlling the current change rate in the falling phase (d phase) calculated by the functional block in FIG. C) is a gate drive signal G "of the upper and lower stage FETs of each phase calculated by the functional block of FIG.
_1-10 and FIG. _8D show the phase relationship between the waveforms of the rotor position detection signal S _ae .

In this case, the logical operation of FIG.
The gate drive signal G " _1-10 shown in FIG. 8C is generated from the rotor position detection signal S _ae and the electromagnetic torque direction command DRCT shown in FIG. 8D, and the drive signal G" _1-10 and the duty ratio Duty1 are generated.
From the PWM signal of FIG.
_1-10 is generated.

Also, the logical operation of FIG.
A gate drive signal G ′ _PC for the falling phase (d phase) of FIG. 7B is generated from the rotor position detection signal S _{ae of} FIG. 7D and the electromagnetic torque direction command DRCT.

This gate drive signal G ′ _PC is the signal shown in FIG.
Becomes "High" (1) when the gate drive signal G " _{1-10 of this} signal becomes" Low "(0), that is, the conventional drive signal G" _1-10 for this phase becomes "Low" (current when it is falling) of, in another drive signal G _'PC, controls the fall of the current. For example, Td1 in FIG.
When the gate drive signal G ” _1-10 of one of the upper FETs becomes“ Low ”at the electrical angle of 18, the gate drive signal G ′ _PC of Td1 in FIG. 7B becomes“ High ”. it then becomes "Low" at the electrical angle 54. from this gate drive signal G 'PWM signal _PC and duty ratio Duty2, the gate drive signal G _PC of a falling phase (d phase) is generated.

The actual FET gate drive signal G _1-10 is generated by combining G ′ _1-10 and G _PC .

By controlling the current change rate of the falling phase in this way, as shown in FIG. 9, the current change rate of the falling phase (d phase) and the current changing rate of the rising phase (a phase) at the time of exciting current switching are changed. Can be matched. Therefore, the current fluctuation and the torque fluctuation are significantly suppressed as compared with the torque fluctuation waveform (FIG. 17) due to the current fluctuation in the conventional FET driving method.

[Embodiment 2] When the excitation current is switched, the current change rate of the rising phase (for example, a phase) is controlled so as to be equal to the current change rate of the PWM signal of the falling phase (for example, d phase).

In this case, since the duty ratio Duty _2-1 = 0 of the PWM signal for the d-phase, the PWM signal for the a-phase
From the equation (18), the duty ratio Duty _{2-2 of the} signal is expressed as Duty _2-2 = Duty1 + 0.5 + K _m · ω / 2Vb (22) or from the equation (19), Duty _2-2 = 2 · Duty1− (2i ) R / 2Vb (23) However, if Duty _2-2 > 1,
Duty _2-2 = 1.

FIG. 10A shows the back electromotive voltage E _{ae of} each phase, and FIG. 10B shows the gate signal G ′ for controlling the current change rate of the rising phase (a phase) calculated by the functional block of FIG. _PC , Figure 1
1 (C) shows the gate drive signal G " _1-10 of the upper and lower stage FETs of each phase calculated by the functional block of FIG. 5, and FIG. 11 (D) shows the phase relationship of each waveform of the rotor position detection signal S _ae. Show.

In this case, the logical operation of FIG.
And a rotor position detection signal S _ae and the electromagnetic torque direction command DRCT of (D), the gate drive signal G _"1-10 in FIG. 11 (C)
Is generated, and the driving signal G " _1-10 and the duty ratio Duty
1 from the PWM signal of FIG.
_1-10 is generated.

Also, the logical operation of FIG.
And a 1 (D) of the rotor position detection signal S _ae and the electromagnetic torque direction command DRCT, rising phase of FIG. 10 (B) (a phase) gate drive signal G _'PC is generated.

This gate drive signal G ′ _PC is
(C) The gate drive signal G " _1-10 is" High "(1)
Becomes "High" (1). That is, when the gate drive signal G " _1-10 for this phase becomes" High "(rise of current), another drive signal G ' _PC
Control the rise of the current. For example, FIG.
When the gate drive signal G " _1-10 of Ta1 (one of the upper FETs) of FIG. 10C becomes" High "at the electrical angle of 18, the gate drive signal G ' _PC of Ta1 of FIG. Is "H
high "and immediately" Low "in order to make the rise of the gate drive signal G" _1-10 faster. This gate drive signal G ' _PC and the duty ratio Duty
From the two PWM signals, a gate drive signal G _PC of a rising phase (a phase) is generated.

The actual FET gate drive signal G _1-10 is generated by combining G ′ _1-10 and G _PC .

By controlling the current change rate of the rising phase in this way, as shown in FIG. 12, the current change rate of the rising phase (a phase) at the time of the switching of the excitation current is reduced in the aforementioned section, as shown in FIG. (D phase) current change rate. Therefore, the fluctuation is greatly suppressed as compared with the torque fluctuation waveform (FIG. 17) due to the current fluctuation in the conventional FET driving method.

Further, in the case of the second embodiment, it is difficult to completely match the current change rates of the rising phase and the falling phase. However, when compared with FIG. (Ie, the time from the start of switching until the current is stabilized) can be advantageously reduced. As a result, when the motor rotates at a high speed, the current stabilization time between the two current switching points becomes longer, which contributes to a reduction in current fluctuation and torque fluctuation.

Third Embodiment This is a combination of the first and second embodiments. That is, when the excitation current is switched, both the current change rate of the PWM signal with respect to the falling phase (for example, d phase) and the current change rate of the rising phase (for example, a phase) are controlled so that both become the same.

In this case, the duty ratio Duty _2-1 of the PWM signal with respect to the d-phase is selected in the range of 0 <Duty _2-1 <0.5 + K _m · ω / 2Vb (24) from the equation (20). Further, the duty ratio Duty _2-2 of the PWM signal with respect to the a-phase is obtained from Expression (18) as Duty _2-2 = Duty 1-Duty _2-1 + 0.5 + K _m · ω / 2 Vb (25). However, if Duty _2-2 > 1,
Duty _2-2 = 1.

Alternatively, the duty ratio Duty _2-1 of the PWM signal with respect to the d phase is selected in the range of 0 <Duty _2-1 <Duty 1− (2i) R / 2Vb (26) from the equation (21). Further, the duty ratio Duty _2-2 of the PWM signal for a phase of the formula (19) from _{Duty 2-2 = 2 · Duty1- Duty 2-1} - is obtained as (2i) R / 2Vb ... ( 27). However, if Duty _2-2 > 1,
Duty _2-2 = 1.

FIG. 13A shows the back electromotive voltage E _{ae of} each phase, and FIG. 13B shows the current of the falling phase (d phase) and the rising phase (a phase) calculated by the functional block of FIG. change rate controlling gate drive signal G _'PC, FIG 14 (C) is computed in function block in FIG. 5, the gate drive signal G _"1-10 of each phase on the lower FET, FIG. 14 (D) the rotor The phase relation of each waveform of the position detection signal S _ae is shown.

In this case, the logical operation of FIG.
From the rotor position detection signal S _ae (D) and the electromagnetic torque direction command DRCT, the gate drive signal G ″ in FIG.
_1-10 is generated, and the gate drive signal G ′ _1-10 is obtained from the drive signal G ″ _1-10 and the PWM signal having the duty ratio Duty1.
Is generated.

Also, the logical operation of FIG.
The gate drive signal G ′ _PC for the falling phase (d phase) and the rising phase (a phase) in FIG. 13B is generated from the rotor position detection signal S _ae and the electromagnetic torque direction command DRCT of FIG. You.

This gate drive signal G ′ _PC is
(C) The gate drive signal G " _1-10 is" High "(1)
Or, when it becomes "Low" (0), it becomes "Low" (0) or "High" (1). That is, the drive signal G "
_1-10 is “High” (rising current) or “Lo”
w "when it is (the fall of the current), in another drive signal G _'PC, to control the rise or fall of the current. For example, FIG. 14 (C) of Td1, Ta1 gate drive signal G" _1-10 are “Lo” at the electrical angle of 18
w ”and“ High ”, Td1 and Td1 in FIG.
Each gate drive signal G ′ _PC of Ta1 is “High”.
After that, the gate drive signal of Ta1 immediately becomes “Lo”.
w ”, and the gate drive signal of Td1 becomes“ Low ”at the electrical angle 54. These gate drive signals G ′ _PC
From a PWM signal having a duty ratio Duty2, falling phase (d phase) and a gate driving signal G _PC for rising phase (a phase) is generated.

The actual FET gate drive signal G _1-10 is generated by combining G ′ _1-10 and G _PC .

By controlling the current change rates in both the falling phase and the rising phase in this way, as shown in FIG. The current change rates of the falling phase (d phase) and the rising phase (a phase) at the time of switching the excitation current can be matched. Therefore, the fluctuation is greatly suppressed as compared with the torque fluctuation waveform (FIG. 17) due to the current fluctuation in the conventional FET driving method.

Further, also in the case of the third embodiment, it is difficult to completely match the current change rates of the rising phase and the falling phase. However, similarly to the second embodiment, the transient time of the current switching can be shortened. There is an advantage. Thus, when the motor rotates at high speed, the current stabilization time between two current switching points becomes longer, which contributes to reduction of current fluctuation and torque fluctuation.

In the above embodiment, the duty ratio Duty1 of the first PWM signal, the rotor position Sa-e and the electromagnetic torque Tm
From the direction command DRCT, the gate drive signal G ′ of each FET
Decide _1-10 . On the other hand, to control the rate of change of the falling and / or rising current of the phase to be switched, the duty ratio Duty of the second PWM signal with respect to the phase to be switched is
Calculate 2. This operation, the motor current value I as described above, the duty ratio of the first PWM signal Duty1, the motor rotation angular speed ω and the voltage constant Km, supplied to the motor drive circuit 23
Power supply voltage Vb and the motor coil circuit 12
This is performed using the function f of the resistance R of the equivalent circuit including the data drive circuit 23 . Therefore, the current detection circuit 24 detects the motor current value I, obtains the duty ratio Duty1 of the first PWM signal from the output of the current feedback control, and generates a switching signal from the rotor position detection signal Sa-e.
The motor rotational angular speed ω is detected. Then, by a logical operation of the rising and / or falling phase FET drive start signal G ′ _PC and the duty ratio Duty2, the FET drive control signal G _PC of the phase to be actually switched is determined, and these signals G ′ _1-10 , controls the driving of the motor by G _PC.

In the above description, the embodiment in which the present invention is applied to the control of a five-phase brushless motor has been described. However, the present invention is not limited to the five-phase brushless motor. The rate of change may be controlled in the same manner as in the above embodiment.

The present invention can also be used to control a ball screw type hollow shaft brushless motor for electric power steering. Further, the present invention is not limited to a brushless motor, and can be applied to control of a motor (for example, a linear motor) driven and controlled by a rectangular wave signal.

[Brief description of the drawings]

FIG. 1 is a sectional view of a five-phase brushless motor.

FIG. 2 is a circuit diagram of the electric power steering apparatus according to the embodiment.

FIG. 3 is a functional block diagram of a control circuit used in the device of FIG.

FIG. 4 is a functional block diagram of a logical operation unit that generates a gate drive signal in FIG.

FIG. 5 is a diagram showing an operation block constituting a first FET gate drive signal logic operation unit in FIG. 4;

FIG. 6 is a block diagram showing a configuration example of a second FET gate drive signal logical operation unit in FIG. 4;

FIG. 7 is a waveform diagram showing a back electromotive voltage E _ae of each phase and a gate signal G ′ _PC for controlling a current change rate of a falling phase when the excitation current is switched in the five-phase brushless motor according to the first embodiment of the present invention.

FIG. 8 shows each phase driving signal G " _1-10 generated in the first embodiment.
And a waveform diagram of the rotor position detection signal S _ae .

FIG. 9 is a diagram showing a current change and an electromagnetic torque change of each phase when the excitation current is switched according to the first embodiment.

FIG. 10 is a waveform diagram showing a back electromotive voltage E _ae of each phase of the five-phase brushless motor and a gate signal G ′ _PC for controlling a current change rate of a rising phase when the excitation current is switched according to the second embodiment of the present invention.

FIG. 11 shows each phase drive signal G ″ generated in the second embodiment.
FIG. 10 is a waveform diagram of _1-10 and a rotor position detection signal S _ae .

FIG. 12 is a diagram showing a current change and an electromagnetic torque change of each phase when the excitation current is switched according to the second embodiment.

FIG. 13 shows a back electromotive voltage E _ae of each phase of the five-phase brushless motor and a gate signal G ′ _PC for controlling a current change rate of a falling phase and a rising phase when the excitation current is switched according to the third embodiment of the present invention. Waveform diagram.

FIG. 14 is a phase drive signal G ″ generated in the third embodiment.
FIG. 10 is a waveform diagram of _1-10 and a rotor position detection signal S _ae .

FIG. 15 is a diagram showing a current change and an electromagnetic torque change of each phase when the excitation current is switched according to the third embodiment.

FIG. 16 is a waveform diagram of an exciting current of each phase of the five-phase brushless motor.

FIG. 17 is a diagram showing a current change and an electromagnetic torque change of each phase when a conventional excitation current is switched.

[Explanation of symbols]

1: 5-phase brushless motor, 2: cylindrical housing, 3
a, 3b bearing, 4 rotating shaft, 5 permanent magnet, 6 stator, 6a to 6e exciting coil, 7 rotor, 8 permanent magnet, 9 stay, 10 support substrate, 11 phase detecting element , 12 ... Coil circuit, 20 ... Motor drive control device, 2
DESCRIPTION OF SYMBOLS 1 ... Control circuit, 22 ... FET gate drive circuit, 23 ... Motor drive circuit, 24 ... Current detection circuit, 25 ... Rotor position detection circuit , 26 ... Power supply, 27 ... External circuit , 31 ... Current control part, 32 ... PW M duty calculator, 33: motor rotation speed calculator, 34: FET gate drive signal calculator.

──────────────────────────────────────────────────続 き Continued on the front page (58) Field surveyed (Int.Cl. ⁷ , DB name) H02P 6/10 H02P 6/08 H02P 6/14

Claims (2)

(57) [Claims] 1. An apparatus for controlling driving of a motor having a plurality of excitation phases, comprising: a driving unit for generating an excitation current to be supplied to each excitation phase of the motor; and a switching of the excitation current for each excitation phase. Do as above
Control means for controlling the driving means, the control means comprising an excitation phase in which the excitation current is not switched.
And the first PWM signal with respect to
The second P for the exciting phase and / or the falling exciting phase
By performing a composite operation with the WM signal, the excitation current
At the time of switching, the excitation current rises and the excitation phase falls.
Drive to make the current change rate of the exciting phase equal or comparable
A motor drive control device for generating a signal (G _1-10 ) and supplying the signal to the drive unit. 2. The motor drive control device according to claim 1, wherein said drive means includes a drive circuit for generating an excitation current to be supplied to a plurality of excitation coils of said motor, and wherein a duty ratio of said second PWM signal is provided. (Duty2)
Motor current value (I) at the time of switching, the first PWM signal
The duty ratio of (Duty1), the rotational angular velocity of the motor
(Ω), the voltage constant (Km), and the power supplied to the drive circuit.
Source voltage (Vb), and the excitation coil circuit of the motor and the
A motor drive control device, which is a function of a resistance (R) of an equivalent circuit including the drive circuit .