JP3250555B2 - Inverter device for switch reluctance motor and control method thereof - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

The present invention relates to an inverter device for driving a switched reluctance motor and a control method thereof.

[0002]

2. Description of the Related Art Compared to an induction motor (hereinafter referred to as an AC motor), a switched reluctance motor which can be downsized in principle because of its low current and large torque characteristics, and has a simple motor structure that can greatly reduce the cost. (Less than,
An SR motor is abbreviated.

Generally, the generated torque τ of an SR motor can be expressed by the following equation (1).

[0004]

(Equation 1) Where iu, iv, iw are the winding currents of each phase, Lu, L
v and Lw indicate the self-inductance of each phase winding, and θ indicates the position angle of the rotor.

FIG. 24 shows the relationship between the state of change in the self-inductance of each phase winding of the SR motor and the conduction waveform.

As shown in FIG. 24E, the change in the self-inductance of each winding with respect to the rotation angle can be approximated to a triangular wave, and each has a phase difference of 30 ° in mechanical angle. Also, as can be seen from Equation 1, the positive / negative torque is determined by the rate of change of the inductance and does not depend on the current polarity. Therefore, when a positive torque is desired, the rectangular wave current is applied to the section where the rate of change of the inductance is positive. A controllable inverter circuit and such control are required.

FIG. 24 shows that the number of poles of the stator / rotor is six.
Although the case of / 4 is illustrated, similar inverter circuits and control are required for SR motors having other combinations of pole numbers.

FIG. 25 is an electric circuit diagram showing a conventional inverter circuit for driving an SR motor. (A) in FIG.
Shows an inverter main circuit, and FIG. 25B shows a waveform control circuit.

In this inverter main circuit, a switching transistor for energization control corresponds to each phase winding by a D.
It is connected to the upper (+) side and the lower (-) side of the C link, respectively.
A corresponding winding serving as a load is connected between the collector terminal of the side switching transistor and a diode that operates when each switching transistor is OFF is a DC.
The anode of each diode on the positive side is connected to the collector terminal of the corresponding switching transistor on the negative side, and the cathode of each diode on the negative side is the emitter terminal of the corresponding switching transistor on the positive side. Are connected respectively.

The waveform control circuit turns on the switching transistor connected to the minus side corresponding to each phase winding based on a signal output from the commutation timing generator, and connects to the plus side. Switching transistor
Based on the output signal obtained by inputting the deviation between the detection current detected by the current detector and the command current output from the current amplitude command generation unit to the hysteresis comparator and the signal output from the commutation timing generation unit ON-OFF
Control, that is, pulse width modulation (hereinafter abbreviated as PWM).

That is, the switching transistor connected to the + side is turned on when the deviation is positive, and the DC voltage V is applied to both ends of the winding via both switching transistors.
DC is applied and the winding current increases. The switching transistor connected to the + side is turned off when the deviation is negative, and 0 voltage is applied to both ends of the winding by the switching transistor connected to the-side and the diode connected to the-side. The winding current is attenuated by the winding resistance.

Therefore, by repeatedly turning on and off the switching transistor connected to the + side,
A waveform having a ripple having a hysteresis width with respect to the current command can be obtained.

When the energized phase is switched, both switching transistors are turned off, and a reverse voltage (-VDC) can be applied by diodes connected to the + and-sides of the DC link, and the current can be rapidly turned off. .

It is to be noted that the switching transistor connected to the minus side may be subjected to PWM based on the output of the hysteresis comparator in place of the switching transistor connected to the plus side.
At the time of WM operation, both switching transistors are simultaneously
M may be used.

In addition to the hysteresis comparator system, for example, a triangular wave comparison system or the like can be adopted.

FIG. 26 is a block diagram showing a configuration of a current control circuit employing a triangular wave comparison method.

In this current control circuit, a current flowing through a winding connected to an inverter circuit for one phase is detected by a current detector 61, and the detected current and a current amplitude command generator 6
2 is supplied to the inverting input terminal and the non-inverting input terminal of the operational amplifier 63 to detect a current deviation, and the detected current deviation is calculated by a PI computing unit (a computing unit that performs a proportional operation and an integral operation) ) 64 to calculate a voltage command. The calculated voltage command is supplied to the + input terminal of the comparator 65, and the oscillation output from the triangular wave oscillator 66 is supplied to the-input terminal of the comparator 65. By comparing the magnitude of the oscillation output with the oscillation output, a PWM signal is output.

The commutation signal output from the commutation timing generator 67 is directly supplied to the-side switching transistor, and the PWM signal is supplied to the + side through the AND gate 68 controlled by the commutation signal. It supplies to the switching transistor.

The operation of this current control circuit is as follows.

By supplying the difference between the current amplitude command and the detected current to the PI calculator 64, the inverter output voltage command can be calculated. By comparing the inverter output voltage command with a triangular wave (carrier), the PW
The signal is converted into an M signal and supplied as an ON-OFF instruction signal to the + side switching transistor.

Therefore, a pair of switching transistors can be ON-OFF controlled similarly to the hysteresis comparator type current control circuit.

In the current control circuit shown in FIG. 26, the responsiveness of the current is related to the constant of PI calculation (proportional (p) gain / integral (I) gain). On the other hand, it is necessary to set the frequency of the triangular wave somewhat higher. Usually, for example, when it is desired to obtain a current response of 2 kHz, the frequency of the triangular wave is set to 10 times or more (20 kHz or more).

The current control circuit shown in FIG. 26 generally has lower current response than the hysteresis comparator, but can reduce current ripple and reduce current harmonics caused by PWM. Especially suitable.

It is to be noted that a sawtooth wave other than a triangular wave can be used as a carrier for converting the inverter output voltage command into a PWM signal.

[0025]

In the inverter main circuit shown in FIG. 25A, the switching transistor and the diode of each phase for controlling the winding current are arranged independently for each phase. This is an ideal circuit configuration because energization control is possible only in a section where the rate of change of the self-inductance is positive (negative section depending on the command torque polarity).
Note that the waveform to be energized is not limited to the above-described rectangular wave, and a waveform other than the rectangular wave may be adopted according to the electromagnetic characteristics of the SR motor.

However, there are the following problems as compared with the widely used AC motor inverter circuit (see FIG. 27).

(1) Since the number of connection lines to the SR motor is larger than that of the AC motor, the connection process to the SR motor becomes complicated. While the number of drawers is five, the number of power line leads is eight in the inverter circuit for the SR motor, and accordingly, the size of the module of the inverter for the SR motor is larger than the size of the module of the inverter for the AC motor. As a result, the cost is increased.

(2) In order to construct an inverter circuit for an AC motor, various modules as shown in FIGS. 28A, 28B and 28C are supplied at a low cost. However, inverter circuits for SR motors are not widely used and are expensive. In this regard, it is conceivable to divert the AC motor inverter circuit. In this case, the winding of the SR motor is connected in Y or Δ connection to the AC motor inverter circuit. become. If such a connection is made, a current flows constantly in at least two phases, and a reverse torque is inevitably generated due to the relationship that the sum of the currents of the respective phases becomes 0, so that the efficiency is significantly reduced. Conversely, if all of the inverter circuits for the SR motor are composed of discrete products, the required number of elements is one.
It becomes 2. As a result, if the capacity is several kW or more, the circuit size becomes large, the assembly becomes complicated, and the cost is increased.

The cost reduction effect of the inverter circuit impairs the cost reduction effect of the SR motor.

[0030]

SUMMARY OF THE INVENTION The present invention has been made in view of the above-mentioned problems, and has an inverter device for an SR motor capable of reducing the number of power lines drawn out without lowering the efficiency and consequently achieving cost reduction. It is intended to provide a control method thereof.

[0031]

According to a first aspect of the present invention, there is provided an inverter device for an SR motor for selectively supplying an operating voltage to each phase winding of the SR motor. A pair of series circuits consisting of a switching element and a diode for supplying an operating voltage to the line,
It includes an inverter circuit in which the midpoint of one series circuit corresponding to each phase and the midpoint of the other series circuit corresponding to the other phase are connected by an equalizing line.

The inverter device for an SR motor according to a second aspect of the present invention further includes an inverter control means for controlling each switching element so that the voltages at both middle points connected by the equalizing line become equal to each other during energization and commutation. Including.

According to a third aspect of the present invention, there is provided an inverter device for an SR motor which employs a switched reluctance motor having windings for three phases, and outputs only the inverter output current supplied to the windings for two phases from the inverter circuit. It further includes current detecting means for detecting.

According to a fourth aspect of the present invention, there is provided an inverter apparatus for an SR motor which keeps the open winding in a non-energized state irrespective of a spike voltage generated in the open phase winding during commutation. The circuit is connected to each phase winding of a switched reluctance motor.

[0035]

[0036]

In the inverter device for an SR motor according to the first aspect, when the operating voltage is selectively supplied to each phase winding of the SR motor, the operating voltage is supplied to each phase winding of the SR motor. And a series circuit composed of a switching element and a diode, and equalizes the midpoint of one series circuit corresponding to each phase and the midpoint of the other series circuit corresponding to the other phase. Since an inverter circuit connected by a line is employed, the number of power line leads can be reduced, and the cost can be reduced. In addition, the number of necessary elements can be reduced by configuring the inverter circuit using a general-purpose module product,
From this aspect, cost reduction can be achieved.

In the inverter device for an SR motor according to the second aspect, the inverter control means controls each switching element so that the voltages at both middle points connected by the equalizing line become equal to each other during energization and commutation. Can be controlled.

Therefore, the same operation as the first aspect can be achieved without lowering the efficiency of the SR motor.

In the inverter device for an SR motor according to the third aspect, when the SR motor has windings for three phases, only the inverter output currents for the two phases are detected to detect all phases. The inverter output current can be detected and controlled, the required number of current detection means can be reduced, and the same operation as that of claim 1 or claim 2 can be achieved.

According to the inverter device for an SR motor of the present invention, the open winding is kept in the non-energized state irrespective of the spike voltage generated in the open winding during commutation. Since the inverter circuit and the phase windings of the switched reluctance motor are connected to each other, it is possible to reduce the efficiency and reduce the operating range due to the reverse torque in addition to the effects of any one of claims 1 to 3. It can be prevented before it happens.

[0041]

[0042]

BRIEF DESCRIPTION OF THE DRAWINGS FIG. 1 is a block diagram of an inverter for an SR motor according to an embodiment of the present invention;

FIG. 10 shows the operation waveforms of the respective parts when the commutation from the u phase to the v phase is performed by using the conventional SR motor inverter device shown in FIG.
The operation waveforms of the respective parts when the commutation from the phase to the u-phase is performed are as shown in FIG. Note that iu and iv indicate currents flowing through the u-phase and v-phase windings, vDu- indicates a voltage across the diode Du-, vTv- indicates a voltage across the switching transistor Tv-, and Tu +, Tu
−, Tv +, Tv− are switching transistors Tu
+, Tu-, Tv +, Tv- ON (1) -OFF
(0) state is shown.

Here, regarding the commutation from the u phase to the v phase,
Focusing on vDu− and vTv− from immediately after commutation until the u-phase current is cut off (becomes 0 [A]), the diode Du− is ON during this period in which the currents overlap.
Then, the voltage vDu− between both ends becomes 0 [V]. On the other hand, since the ON signal (1 level signal) is input to the switching transistor Tu−, the voltage vTv− between both ends thereof.
Is also 0 [V]. Note that once the u-phase current is cut off, the next switching transistor Tu
Until an ON signal is input to both + and Tu-, no current flows because no voltage is applied to the u-phase winding.

Next, regarding the commutation from the v phase to the u phase, focusing on vDu− and vTv− from immediately after commutation until the v phase current is cut off (becomes 0 [A]), the u phase Unlike the case of commutation from the current to the v-phase, as the switching transistor for inputting the PWM signal, a switching transistor Tu- for the u-phase and a switching transistor Tv- for the v-phase are selected. Therefore, during this period in which the currents overlap, the diode Dv
+ Is turned on, and the voltage vDv + between both ends becomes VDC [V]. On the other hand, since an ON signal (1 level signal) is input to the switching transistor Tv +, the voltage vTv− between both ends thereof also becomes VDC [V]. You can see that it is.

From the above, by devising the way of giving the PWM signal of the conventional inverter device for SR motor shown in FIG. 25, as shown in FIGS. 10 to 11, the voltage between the adjacent elements can be made the same. As a result, as shown in FIG. 1, it can be seen that adjacent elements can be connected by a pressure equalizing line. As a result, the number of connection lines to the SR motor can be reduced to two lines as compared with the conventional SR inverter to four lines.

In the circuit configuration shown in FIG. 1, a winding in which no current flows in advance is used until an ON signal is input to both switching transistors connected to both ends of the winding.
Since no voltage is applied and no current flows, when the current of the phase before switching stops flowing when the energized phase is switched,
A control method for energizing another phase is also conceivable.

Referring to FIG. 2, the method of controlling the inverter for an SR motor according to the present invention will be described in further detail.

This inverter device includes a + side DC link 2 connected to each of a + terminal and a − terminal of a DC power supply 1.
a,-side DC link 2b. Then, switching transistors Tu +, Tv +, Tw + for energization control corresponding to each phase winding are connected to the + side DC link 2a,
The switching transistors Tu-, Tv-, Tw-
Side DC link 2b. Corresponding phase windings 3u, 3v, 3w are connected between these switching transistors Tu +, Tv +, Tw + and switching transistors Tu-, Tv-, Tw-, respectively. The diodes Du +, Dv +, and Dw + that operate when the switching transistor is OFF are connected to the + side DC.
Diodes Du-, Dv-, and Dw- that operate when the switching transistor is turned off are connected to the link 2a, respectively, to the negative DC link 2b. The anodes of the diodes Du +, Dv +, Dw + are connected to the collector terminals of the switching transistors Tu−, Tv−, Tw−,
Respectively connected, and diodes Du−, Dv−, Dw−
Are switching transistors Tu +, Tv
+ And Tw + emitter terminals. Further, a connection point between the switching transistor Tu + and the diode Du− and a connection point between the diode Dv + and the switching transistor Tv− are connected to the first equalizing line 4.
u, and a connection point between the switching transistor Tv + and the diode Dv− and a connection point between the diode Dw + and the switching transistor Tw− are connected to the second equalizing line 4.
Connected by v. Furthermore, the switching transistors Tu +, Tv +, Tw +, Tu−, Tv−, Tw−
Is provided with an inverter control unit 5 for controlling the operation.

FIG. 3 is a block diagram showing an example of the configuration of the inverter control unit 5.

The inverter control circuit 5 detects the current of each phase and outputs a current command value. The u-phase current detection / current command value output unit 5u1 and the v-phase current detection / current command value output unit 5v1, w Phase current detection / current command value output unit 5w
P which generates a PWM signal by using an output signal from
WM signal generators 5u2, 5v2, 5w2, and u-phase commutation signal generators 5u3, 5u3, which generate commutation signals of respective phases according to the position angles and supply the commutation signals to one switching transistor as a switching command. Flow signal generator 5v3, w
A phase commutation signal generator 5w3, a PWM signal generator 5u2,
The PWM signal from each of 5v2 and 5w2 and commutation signals from each of u-phase commutation signal generation unit 5u3, v-phase commutation signal generation unit 5v3 and w-phase commutation signal generation unit 5w3 are input and the other switching is performed. AND gates 5u4, 5v4, 5 that output switching commands to transistors
w4, a forward / reverse signal generator 5v5 for outputting forward / reverse signals for instructing forward / reverse rotation, and a forward / reverse signal output from the forward / reverse signal generator 5v5. And a multiplexer 5v6 for exchanging a switching command for the other switching transistor.

The PWM signal generator may be provided with, for example, a hysteresis comparator system shown in FIG.
6 can be used.

The operation of the inverter device for an SR motor having the above configuration is as follows.

First, the operation in the case of commutation from the w-phase to the u-phase as shown by the dashed arrow and the solid arrow in FIG. 4 will be described.

In order to maintain the continuity of the w-phase current by the commutation signal (due to the effect of inductance), the diode D
w + turns ON, and as a result, one end of the v-phase winding 3v is connected to the + side of the DC power supply 1. Here, when the ON / OFF control of the switching transistor Tu + is performed by the u-phase PWM signal, when the switching transistor Tu + is turned off, the diode Du− is connected to maintain the continuity of the current of the u-phase winding 3u. ON, the other end of the v-phase winding 3v is connected to the negative side of the DC power supply 1. As a result, a disadvantage occurs in that a DC voltage is applied to both ends of the v-phase winding 3v and a current flows (see FIG. 5).

However, the u-phase PWM signal is supplied to the switching transistor Tu−, and the commutation signal (always ON signal during the energization period) is supplied to the switching transistor Tu +.
, The other end of the v-phase winding 3v is always connected to the + side of the DC power supply 1 during the u-phase conduction period.
It is possible to prevent the inconvenience that a current flows through the v-phase winding 3v (see FIG. 6).

Next, the operation in the case of commutation from the u-phase to the w-phase as shown by the dashed arrow and the solid arrow in FIG. 7 will be described.

In order to maintain the continuity of the u-phase current by the commutation signal (due to the effect of inductance), the diode D
u- is turned on, and as a result, one end of the v-phase winding 3v is connected to the negative side of the DC power supply 1. Here, when the ON / OFF control of the switching transistor Tw- is performed by the w-phase PWM signal, when the switching transistor Tw- is turned off, the diode Dw + is used to maintain the continuity of the current of the w-phase winding 3w. Is turned on, and the other end of the v-phase winding 3v is connected to the + side of the DC power supply 1. As a result, a disadvantage occurs in that a DC voltage is applied to both ends of the v-phase winding 3v and a current flows (see FIG. 8).

However, the w-phase PWM signal is supplied to the switching transistor Tw +, and the commutation signal (always ON signal during the energization period) is supplied to the switching transistor Tw-.
, The other end of the v-phase winding 3v is always connected to the negative side of the DC power supply 1 during the w-phase energizing period.
It is possible to prevent the inconvenience that a current flows through the v-phase winding 3v (see FIG. 9).

As is clear from the above description, no matter which phase is commutated, v is connected in the middle of the inverter circuit.
A current can be prevented from flowing through the phase winding 3v.

From the description based on FIGS. 4 to 9, it can be understood that an ideal rectangular wave current can be supplied to each winding of the SR motor by controlling the ON / OFF of each switching transistor as shown in Table 1. In Table 1,
0 is OFF, 1 is ON, * is PWM operation (ON / O
FF operation).

[0062]

[Table 1] The control circuit of FIG. 3 shows an example of a device configuration for performing the operation of Table 1. Regardless of the forward or reverse phase rotation, a commutation signal is input to the u-phase switching transistor Tu +, and a u-phase PWM signal is input to the gate to the u-phase switching transistor Tu−, while the w-phase switching transistor Tw + is input to the gate. The w-phase PWM signal and the commutation signal are input to the gates of the w-phase switching transistor Tw-, respectively.

As for the v-phase, the v-phase PWM signal and the commutation signal are input to the gates of the v-phase switching transistors Tv + and Tv- via the multiplexer 5v6 and output from the forward / reverse signal generator 5v5. It is switched by forward / reverse signals.

The PWM signal of each phase is ANDed with the commutation signal, so that it is inputted to each gate only during the energization period.

For applications in which loads having the same rotational direction are driven, such as compressors and pumps, the forward / reverse signal generator 5v
5 and the multiplexer 5v6 can be omitted,
The configuration can be simplified.

If the inverter main circuit of FIG. 2 is driven by using the control circuit of FIG. 3, regardless of whether the rotation is normal or reverse,
Waveforms as shown in FIGS. 12 and 13 were obtained in which current flows through each winding only during the energizing period and does not flow during unnecessary periods.

In the inverter device for an SR motor shown in FIGS. 2 and 3, a winding current detector is provided to detect the currents of the windings of all the phases. The number of detectors can be reduced, and the cost can be reduced.

FIG. 14 is an electric circuit diagram showing an inverter main circuit of the inverter device for an SR motor according to the present invention.
One end of the phase winding 3u is connected to one end of the v-phase winding 3v,
The other end of the v-phase winding 3v and one end of the w-phase winding 3w are connected, and the other end of the u-phase winding 3u, the u-phase winding 3u and the v-phase winding 3v
, A connection point between the v-phase winding 3v and the w-phase winding 3w,
The other end of the w-phase winding 3w is connected to the inverter main circuit.

Here, u-phase winding 3u, v-phase winding 3v, w
The currents flowing through the phase winding 3w are iu, iv, and iw, respectively, and the current flowing at the connection point between the u-phase winding 3u and the v-phase winding 3v is defined as ia, v-phase winding 3v, and the w-phase winding 3w. Assuming that the current flowing through the connection point is ib, Equation 2 is obtained.

[0070]

(Equation 2) Then, at the time of forward rotation of the SR motor (u phase → v phase → w phase → u phase
..) Are shown in FIG. 15 as a result of obtaining the current waveform of each part by the relationship of Equation 2. Focusing on the energization control (performing current control) period of each phase, it can be seen from FIG.

[0071]

[Equation 3] On the other hand, when the SR motor rotates in reverse (u-phase → w-phase → v-phase → u-phase
FIG. 16 shows the result of obtaining the current waveform of each part of ()) according to the relationship of Expression 2. Focusing on the energization control (performing the current control) period of each phase, it can be seen from FIG. 16 that the relationship of Expression 4 holds.

[0072]

(Equation 4) As can be seen from the above, the three-phase currents iu, iv, and iw can be calculated by detecting the two-phase currents ia and ib and performing simple four arithmetic operations suitable for normal rotation and reverse rotation. Therefore, the control by the control circuit of FIG. 3 may be performed based on the three-phase currents calculated as described above.

FIGS. 17 to 19 show three-phase currents iu, iv, iw with the detected two-phase currents ia, ib as inputs.
FIG. 2 is an electric circuit diagram showing a configuration of a device for outputting a signal. Note that FIG. 17 corresponds to the case where detection of only forward rotation is sufficient, FIG. 18 corresponds to the case where detection of only reverse rotation is sufficient, and FIG. 19 illustrates detection of normal rotation and detection of reverse rotation. This corresponds to the case of performing.

The device shown in FIG. 17 supplies the detected two-phase currents ia and ib to the inverting input terminal of the operational amplifier 51 via the resistors having the same resistance value R. Then, the non-inverting input terminal of the operational amplifier 51 is connected to 1 / of the resistance.
2 is connected to the ground terminal via a resistor having a resistance value of R / 2, a resistor having a resistance value of R is connected between the non-inverting input terminal and the output terminal, and (-ia-ib) Output a signal. Further, the detected two-phase currents ia, ib,
Signal (-ia-ib) output from the operational amplifier 51
Are supplied to a multiplexer 52 that uses a u-phase commutation signal, a v-phase commutation signal, and a w-phase commutation signal as control signals, respectively, and an output signal from the multiplexer 52 is supplied to a − input terminal of a hysteresis comparator 53, The current amplitude command (the current amplitude command output from the current amplitude command generation unit 54) supplied to the input terminal is compared with the PW
An M signal is output.

The device shown in FIG. 18 is an example of a detection device configuration corresponding to the detection of reverse rotation. This device differs from the device of FIG. 17 in that an operational amplifier 55 for inverting the sign is connected between the multiplexer 52 and the hysteresis comparator 53, and the detected two-phase current ia,
ib, a signal output from the operational amplifier 51 (−ia−i
b) is supplied to the multiplexer 52 that uses the v-phase commutation signal, the w-phase commutation signal, and the u-phase commutation signal as control signals.

The device shown in FIG. 19 is an example of a detection device configuration corresponding to detection of normal rotation and detection of reverse rotation. The detected two-phase currents ia and ib have the same resistance value R, respectively.
And supplied to the inverting input terminal of the operational amplifier 51 through the resistor. Then, the non-inverting input terminal of the operational amplifier 51 is connected to the ground terminal via a resistor having a resistance R / 2, which is a half of the above-mentioned resistance, and a resistance R between the non-inverting input terminal and the output terminal.
, And from this output terminal, (-ia-ib)
The signal of is output. Further, the detected two-phase currents ia,
ib, a signal output from the operational amplifier 51 (−ia−i
b) is supplied to the multiplexer 52 that uses the v-phase commutation signal, the w-phase commutation signal, and the u-phase commutation signal as control signals. Further, the detected two-phase currents ia and ib are supplied to the inverting input terminal of the operational amplifier 51 ′ via the resistors having the same resistance value R. Then, the non-inverting input terminal of the operational amplifier 51 ′ is connected to the ground terminal via a resistor having a resistance value R / 2, which is 上 記 of the above resistance, and a resistor having a resistance value R is connected between the non-inverting input terminal and the output terminal. And outputs a signal (−ia−ib) from this output terminal. The detected two-phase currents ia and ib and the signal (−ia-ib) output from the operational amplifier 51 ′ are used to control the v-phase commutation signal, the w-phase commutation signal, and the u-phase commutation signal, respectively. And an output signal from the multiplexer 52 ′ for inverting the sign.
5 '. Further, the output signal from the multiplexer 52 and the output signal from the operational amplifier 55 'are output to the multiplexer 56 using the normal signal and the reverse signal as control signals.
And the output signal from the multiplexer 56 is supplied to the negative input terminal of the hysteresis comparator 53,
The PWM signal is output from the output terminal in comparison with the current amplitude command supplied to the input terminal (current amplitude command output from the current amplitude command generation unit 54).

FIGS. 20 to 23 show examples of the configuration of the inverter main circuit of the present invention. In FIGS. 20 to 23, the parts surrounded by broken lines indicate module parts {see FIGS. 28A, 28B and 28C}.

When the conventional inverter main circuit is constructed without using any module parts, the number of elements becomes twelve. On the other hand, in the case of FIG. 20, the number of module components is 4, the number of discrete components is 4, and the total number of elements is 8. In the case of FIG. 21, the number of module components is 2, the number of discrete components is 4, and the total number of elements is 6. Further, in the case of FIG. 22, the number of module components is 1, the number of discrete components is 2, and the total number of elements is 3. Furthermore, in the case of FIG. 23, the number of module components is 1, the number of discrete components is 2, and the total number of elements is 3.

That is, as described above, as the inverter main circuit for the SR motor, the one having the equalizing lines 4u and 4v is employed, and therefore, FIG.
(C) Any of the module components can be adopted, and the number of elements can be significantly reduced by employing the module components.

When the inverter main circuit for the SR motor is modularized, only one power line needs to be added, so that the cost increase from the AC motor inverter module (three-phase bridge configuration) is minimized. can do.

In the above embodiment, a spike voltage due to mutual induction is generated in the winding of the non-energized phase at the time of commutation, and this spike voltage turns on an inappropriate diode via the equalizing line, and Unnecessary current that generates a reverse torque may flow through the windings, and if the reverse torque is generated, the efficiency may decrease and the operating range may be reduced.

Further description will be given.

Each phase winding current iu, iv, iw is energized in the direction shown by the arrow in FIG.
Consider a case where commutation control is performed in the forward direction.

FIG. 29 is a diagram showing a simplified circuit state when commutating from the v-phase to the w-phase.

When the switching transistors Tv + and Tv- connected to both ends of the v-phase winding are turned off, the effect of the inductance of the winding (the effect of keeping the current flowing).
As a result, the diodes Dv + and Dv- are turned ON, and a current flows through a path indicated by an arrow in FIG. 29 (hereinafter, this state is referred to as a reflux state).

Further, a spike voltage is generated in the u-phase winding as shown in FIG. 30 (A) or (B) by the v-phase winding voltage at the time of commutation.

Here, when a spike voltage having the polarity shown in FIG. 30B is generated, the diode Du + is turned ON, and a current starts flowing through a path shown by an arrow in FIG. Then, once the diode Du + is turned on, the v-phase current is drawn into the path indicated by the arrow due to the effect of the inductance of the u-phase winding, and a motor reverse torque is generated in the u-phase winding until the v-phase current becomes zero. Current flows.

Conversely, when a spike voltage having the polarity shown in FIG. 30A is generated, the diode Du + enters a reverse bias state, so that a current flows through a path shown by an arrow in FIG. Will not occur.

As can be seen from the above description, by connecting each phase winding to the inverter circuit so that the polarity of the spike voltage becomes the polarity shown in FIG.
Current flows through the path indicated by the arrow in FIG. 9 and no reverse torque is generated.

This will be described in more detail.

FIG. 31 schematically shows the flow of magnetic flux in the iron core portion of the SR motor during commutation.

FIG. 31 shows the flow direction of the winding current, the v-phase main magnetic flux φv generated by the v-phase winding current (see the arrow in FIG. 31), and the u-phase winding current by the v-phase winding current. And a mutual induction magnetic flux φuv (see an arrow in FIG. 31) interlinking the line.

When the v-phase winding current decreases due to commutation, v
Both the phase main magnetic flux φv and the mutual induction magnetic flux φuv start to decrease.
Since the winding voltage is proportional to the time derivative of the magnetic flux linked to the winding, the time change of the steep mutual induction magnetic flux φuv due to the commutation appears as a spike voltage at both ends of the u-phase winding.

Since the spike voltage is a voltage associated with mutual induction between the windings, for example, the connection between the u-phase winding and the inverter circuit is changed to the connection terminal u1 of the u-phase winding as shown in FIGS. , U2 and the connection terminals u +, u− of the inverter circuit, the polarity of the spike voltage can be inverted. Here, a current is supplied to the u-phase winding from the connection terminal u1 to the connection terminal u2, and in addition, FIG.
Assuming that a winding is applied to the pole of the stator so as to generate a main magnetic flux in the direction indicated by the bold arrow in FIG. 4, by connecting as shown in FIG.
As shown in FIG. 7A, the diode Du + does not malfunction during commutation.

FIG. 35 is a simplified diagram showing a circuit state when commutating from the u phase to the v phase.

In this drawing, a spike voltage is generated in the w-phase as shown in FIG. 36 (A) or (B) depending on the connection between the winding and the inverter circuit. But,
Since the u-phase winding in the reflux state is not connected to the w-phase winding, the spike voltage temporarily exceeds VDC and the diode D
Even if w + and Dw- are turned on, it does not lead to drawing the u-phase current. That is, the polarity of the spike voltage can be freely selected for the w-phase (a phase that is not connected to the winding that is in a reflux state during commutation).

FIG. 37 is a diagram showing an example of connection between each phase winding set to prevent generation of reverse torque due to a spike voltage during commutation and an inverter device. Note that S
Winding terminals u1, u2, v1, v2, w of each phase of the R motor
38. As shown in FIG. 38, when the current is applied from the winding terminal u1 to the winding terminal u2 for the u phase,
A main magnetic flux φu is generated in the direction indicated by the thick arrow in FIG. 38A, and the v-phase and the w-phase are similarly similarly transferred from the winding terminal v1 to the winding terminal v2 or from the winding terminal w1 to the winding terminal w2. 38, the main magnetic flux φv or φw in the direction indicated by the thick arrow in FIG. 38 (B) or (C).
Each phase winding is provided so as to generate the following.

FIG. 39 shows the case where the connection is made so as to be affected by the spike voltage {see FIG. 39 (A)} and FIG.
FIG. 40 is a diagram showing current waveforms of respective phases when the connection is made as shown in FIG. 39 (see FIG. 39B).

In the current waveform shown in FIG. 39 (A), an unnecessary current which causes a reverse torque in the u-phase winding and the v-phase winding {see the portion indicated by the arrow in FIG. 39 (A)} is shown. Although it flows, in the current waveform shown in FIG. 39 (B), there is no unnecessary current that causes a reverse torque.

Therefore, by optimizing the connection between each phase winding of the SR motor and the inverter circuit, unnecessary current which causes a reverse torque can be eliminated.
It can be seen that a reduction in the efficiency of the SR motor and a reduction in the operating range can be prevented.

The same can be considered when the commutation phase is set in the order of u-phase → w-phase → v-phase (hereinafter referred to as reverse rotation). Also, if the SR motor and the inverter circuit are connected so that no reverse torque is generated due to the spike voltage in the case of forward rotation, no reverse torque is generated even in the case of reverse rotation.

In the above embodiment, the number of salient poles of the stator is set to 6, and the number of salient poles of the rotor is set to 4.
The present invention is also applicable to an SR motor having a different number of salient poles (for example, an SR motor having a stator having 12 salient poles and a rotor having 8 salient poles).

[0103]

According to the first aspect of the present invention, the number of power lines to be drawn can be reduced, and the cost can be reduced. Further, it is necessary to configure the inverter circuit using a general-purpose module product. It is possible to reduce the number of elements and to achieve a unique effect that cost reduction can be achieved from this aspect as well.

The second aspect of the invention has a unique effect that the same operation as the first aspect can be achieved without lowering the efficiency of the SR motor.

According to the third aspect of the present invention, the required number of current detecting means can be reduced, and the same effects as those of the first or second aspect can be obtained.

The invention according to claim 4 is the invention according to claims 1 to 3.
In addition to the effects of any one of the above, there is an effect that a reduction in efficiency and a reduction in the operating range due to reverse torque can be prevented beforehand.

[0107]

[Brief description of the drawings]

FIG. 1 is an electric circuit diagram showing an embodiment of an inverter main circuit for an SR motor according to the present invention.

FIG. 2 is an electric circuit diagram showing one embodiment of an inverter device for an SR motor according to the present invention.

FIG. 3 is a block diagram illustrating an example of a configuration of an inverter control unit.

FIG. 4 is a diagram illustrating an operation when commutating from a w-phase to a u-phase.

FIG. 5 is a diagram showing waveforms of respective units when u-phase current control is performed by PWM control of a switching transistor Tu-.

FIG. 6 is a diagram showing waveforms of respective parts when u-phase current control is performed by PWM control of a switching transistor Tu +.

FIG. 7 is a diagram illustrating an operation when commutating from a u phase to a w phase.

FIG. 8 is a diagram showing waveforms of respective units when w-phase current control is performed by PWM control of a switching transistor Tw-.

FIG. 9 is a diagram showing waveforms of respective units when performing w-phase current control by PWM control of a switching transistor Tw +.

FIG. 10 is a diagram showing waveforms of respective parts when commutation from the u phase to the v phase is performed.

FIG. 11 is a diagram showing waveforms of respective units when commutation from the v phase to the u phase is performed.

FIG. 12 is a diagram showing waveforms at various portions during normal rotation.

FIG. 13 is a diagram showing waveforms of respective units at the time of reverse rotation.

FIG. 14 is a diagram illustrating the connection between the inverter main circuit for the SR motor and each winding of the SR motor.

FIG. 15 is a diagram showing the sum of a winding current, a current detection signal, and a current detection signal during normal rotation.

FIG. 16 is a diagram showing a sum of a winding current, a current detection signal, and a current detection signal at the time of reverse rotation.

FIG. 17 is an electric circuit diagram showing an example of the configuration of a device for outputting a three-phase current with a two-phase current as an input.

FIG. 18 is an electric circuit diagram showing another example of the configuration of a device for outputting a three-phase current with a two-phase current as an input.

FIG. 19 is an electric circuit diagram showing another example of the configuration of the device for outputting a three-phase current by inputting a two-phase current.

FIG. 20 is a diagram illustrating an example of a configuration using module components.

FIG. 21 is a diagram showing another example of a configuration using module components.

FIG. 22 is a diagram showing still another example of a configuration using module components.

FIG. 23 is a diagram showing still another example of a configuration using module components.

FIG. 24 is a schematic diagram for explaining a conventional SR motor driving method.

FIG. 25 is an electric circuit diagram showing a conventional inverter main circuit and a waveform control circuit for an SR motor.

FIG. 26 is a block diagram showing a configuration of a current control circuit employing a triangular wave comparison method.

FIG. 27 is an electric circuit diagram showing a configuration of an AC motor inverter main circuit.

FIG. 28 is an electric circuit diagram showing a configuration of a distributed module component.

FIG. 29 is a diagram schematically illustrating a circuit state when commutating from the v-phase to the w-phase.

FIG. 30 is a diagram showing a winding voltage and a phase current when commutating from the v-phase to the w-phase.

FIG. 31 is a diagram illustrating the flow of magnetic flux during commutation.

FIG. 32 is a diagram illustrating an example of connection between a u-phase winding and an inverter circuit.

FIG. 33 is a diagram showing another example of the connection between the u-phase winding and the inverter circuit.

FIG. 34 is a diagram illustrating the direction of a main magnetic flux caused by u-phase conduction.

FIG. 35 is a diagram schematically illustrating a circuit state when commutating from the u phase to the v phase.

FIG. 36 is a diagram showing a winding voltage and a phase current when commutating from a u phase to a v phase.

FIG. 37 is a diagram showing a connection example between a motor winding and an inverter circuit set so as to eliminate generation of reverse torque due to a spike voltage at the time of commutation.

FIG. 38 is a diagram illustrating the direction of a main magnetic flux due to energization of each phase winding.

FIG. 39 is a diagram showing a current waveform of each phase when the SR motor is driven.

[Explanation of symbols]

3u, 3v, 3w Winding 4u, 4v Equalizing line 5 Waveform control circuit Du +, Du-, Dv +, Dv-, Dw +, Dw- Diode Tu +, Tu-, Tv +, Tv-, Tw +, Tw- Switching transistor

────────────────────────────────────────────────── ─── Continuation of the front page (56) References JP-A-10-323080 (JP, A) JP-A-9-168298 (JP, A) JP-A-63-129850 (JP, A) (58) Field (Int.Cl. ⁷ , DB name) H02P 7/05 H02M 7/48

Claims (4)

(57) [Claims] An inverter device for selectively supplying an operating voltage to each phase winding (3u) (3v) (3w) of a switched reluctance motor, wherein each phase winding of a switched reluctance motor is provided. (3u) (3
v) Switching elements (Tu +) (Tu−) (Tv +) (Tv−) (Tv) for supplying an operating voltage to (3w)
w +) (Tw-) and diode (Du +) (Du-)
(Dv +), (Dv-), (Dw +), and (Dw-). A series circuit corresponding to each phase has a midpoint between one series circuit and the other corresponding to another phase. An inverter device for a switched reluctance motor, comprising an inverter circuit formed by connecting a middle point of a series circuit with equalizing lines (4u) and (4v). 2. A pressure equalizing line (4u) at the time of energization and commutation.
Each switching element (Tu +) (Tu −) (Tv) such that the voltages at both middle points connected by (4v) become equal to each other.
The inverter device for a switched reluctance motor according to claim 1, further comprising inverter control means (5) for controlling (+) (Tv-) (Tw +) (Tw-). 3. The switched reluctance motor has three phases of windings (3u), (3v), and (3w).
The inverter device for a switched reluctance motor according to claim 1 or 2, further comprising a current detection means for detecting only an inverter output current supplied from the inverter circuit to windings of two phases. 4. An inverter circuit and each phase of a switched reluctance motor so as to keep the open winding in a non-energized state irrespective of a spike voltage generated in the open winding during commutation. Winding (3u) (3v) (3
The inverter device for a switched reluctance motor according to any one of claims 1 to 3, further comprising: