JP3493399B2 - Current control method and apparatus for PWM inverter - Google Patents

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a current control type PWM inverter which controls a current flowing through an inductive load by changing a conduction time of a voltage applied to the load by a PWM (pulse width modulation) inverter. In particular, the present invention relates to a method and an apparatus for comparing a current command value with an actual current and controlling a voltage energization time based on the magnitude.

[0002]

2. Description of the Related Art Conventionally, a current flowing through an inductive load is P
It is known to control using a WM inverter.
An instantaneous value comparison method is known as such a control method.

Although there are various types of inductive loads, they can be miniaturized because they have low current and large torque characteristics in principle as compared with AC motors, and a drastic cost reduction can be expected because the motor structure is simple. Switched reluctance motors (hereinafter abbreviated as SR motors) are also known as inductive loads.

Generally, the torque generated by the SR motor is described as τ = (1/2) .multidot. (Iu.iu.dLu / d.theta. + Iv.iv.dLv / d.theta. + Iw.iw.dLw / d.theta.) (1). Let Where iu, iv, iw are winding currents for each phase,
Lu, Lv, and Lw represent the self-inductance of each phase winding, and θ represents the position angle of the rotor.

In FIG. 3, the stator has 6 poles and the rotor has 4 poles.
The relationship between the change of the self-inductance of each winding of the R motor and the energization waveform is shown.

As shown in FIG. 3, the change of the self-inductance of each winding with respect to the rotation angle can be approximated to a triangular wave, and each has a phase difference of 30 ° in mechanical angle.

As can be seen from the equation (1), the positive torque /
Negative is determined by the rate of change of the inductance and does not depend on the current polarity. Therefore, in order to obtain a positive torque, an inverter and control capable of supplying a rectangular wave current in the section where the rate of change of the inductance is positive are required.

Similar energization control is generally performed in SR motors having other combinations of pole numbers (for example, the stator has 12 poles and the rotor has 8 poles).

4 and 5 are electrical circuit diagrams showing an inverter main circuit for driving an SR motor and a waveform control circuit. The transistors forming the inverter main circuit of FIG. 4 are on / off controlled by the waveform control circuit of FIG.

In the inverter main circuit, transistors for controlling energization are connected to the upper (+) side and the lower (-) side of the DC link, respectively, and the emitter terminal of the + side transistor and-
A motor winding that serves as an inductive load is connected between the collector terminal of the side transistor and the collector terminal. Then, the free wheeling diode which operates when the transistor is off is connected to the + side and the − side of the DC link, respectively, and the anode of the free wheeling diode connected to the + side is connected to the collector terminal of the transistor connected to the − side and the − side. The cathodes of the free wheeling diodes connected to are respectively connected to the emitter terminals of the transistors connected to the + side.

In this waveform control circuit, the transistor T- connected to the-side is turned on, while the transistor T + connected to the + side inputs the deviation between the detected current and the command current to the hysteresis comparator. On / off control is performed based on the obtained output result.

That is, the transistor T + connected to the + side is turned on when the deviation is negative, the DC voltage VDC is applied to both ends of the winding through the transistors T- and T +, and the winding current increases. On the contrary, when the deviation is positive, it is turned off, and a zero voltage is applied to both ends of the winding by the freewheeling diodes connected to the transistors T− and −, and the winding resistance attenuates the winding current. Then, by repeating these operations, it is possible to obtain a waveform having a ripple of a hysteresis width with respect to the current command.

When switching the energized phase, both the transistors T + and T- are turned off, and the + side of the DC link,-
Reverse voltage by the freewheeling diode connected to each side
VDC can be applied and the current can be turned off rapidly.

PWM may be performed by the transistor T- instead of the transistor T +, and the transistors T + and T- may be simultaneously turned on / off during the PWM operation although the current ripple increases.

The transistors T + and T- are turned on by inputting the base signal "1" and turned off by inputting "0". (The same applies to the following explanations)

[0016]

Problems to be Solved by the Invention Transistors T +, T
The voltage equation between the 1-phase windings of the SR motor when both are turned on is VDC = R · i + d {L (θ) · i} / dt = R · i + L (θ) · di / dt + ω · K · i It can be written as (2). Here, L (θ) is the winding self-inductance for each rotor position angle θ, R is the winding resistance, i is the winding current, ω is the rotational angular velocity of the motor, and t is time. Also, K = L (θ) / dθ. The sign of K is inverted depending on the position angle of the rotor, but when operating the motor, K
The energization control is performed in a positive period (see FIG. 3).

When the DC voltage VDC of the inverter main circuit is constant, the third term of the equation (2) increases with the angular velocity of rotation, so the inductance L of the winding of the second term of the equation (2) is L.
The voltage to be shared by (θ) decreases with the rotational angular velocity, which in turn reduces the current change rate di / dt.

On the other hand, when the transistor T- is turned on and the transistor T + is turned off, the voltage equation between the one-phase windings of the SR motor is 0 = R · i + L (θ) · di / dt + ω · K · i ·・ ・ (3) The third term of equation (3) increases with the angular velocity of rotation,
The voltage to be shared by the inductance L (θ) of the winding in the second term of the equation (3) also rises, and thus the current change rate di / d.
increase t.

In the instantaneous value comparison method, since the output of the hysteresis comparator turns on / off the transistors T + and T- so that the deviation of the actual current from the current command value becomes a predetermined value, it depends on the magnitude of the current change rate. The time for the current to reach the predetermined deviation changes, and thus the on / off frequency changes. That is, in the instantaneous value comparison method, the PWM cycle (the on / off frequency of the transistors T + and T−) changes according to the rotational angular velocity.

In general, since the switching loss of a transistor increases with the switching frequency, it is necessary to determine the PWM frequency so that there is no thermal destruction of the element due to the loss. However, in the instantaneous value comparison method, the on / off frequency of the transistors T + and T-, that is, the PWM frequency changes according to the rotation speed of the motor, so that the PW is within a predetermined range.
There is a problem that the work of adjusting the hysteresis width of the hysteresis comparator to set the M frequency becomes complicated.

To solve this problem, for example, Nishimura et al., "High-performance servo system of permanent magnet synchronous motor", Institute of Electrical Engineers of Japan, Semiconductor Power Conversion Study Group, SPC 84-13, 1
In 984, a method of adjusting the hysteresis width in response to the switching frequency has been proposed, but this causes a problem that the configuration becomes complicated.

Further, since the comparator is used in the instantaneous value comparison method, the current control gain is extremely high, and malfunction due to noise contained in the detected current is likely to occur. Particularly, in an inexpensive current detector, spike-like noise is superimposed on the detected current due to a sudden change (0 → Vdc, Vdc → 0 or Vdc → −Vdc) of the inverter output voltage at the time of switching, which causes malfunction, that is, There is a problem that unnecessary switching is performed and the switching frequency is increased.

FIG. 6 shows a waveform when a current detector having no spike noise is used, and FIG. 7 shows a waveform when an inexpensive current detector is used. Comparing FIG. 6 with FIG. 7, when an inexpensive current detector is used, a malfunction occurs due to spike noise, and the transistor T + turns on.
It can be seen that the number of off times is increasing.

A low-pass filter may be provided to reduce spike noise superimposed on the detected current. However, if the filter characteristic is set to have a cutoff frequency that can sufficiently reduce spike noise, the SR motor will be The high-frequency component contained in the rectangular wave for driving cannot be detected, and it becomes difficult to control the SR motor accurately. Specifically, the control waveform when using a low-pass filter that can sufficiently attenuate spike noise superimposed on the detected current to the extent that the hysteresis comparator does not malfunction is shown in FIG. Sufficient current tracking cannot be realized.

[0025]

SUMMARY OF THE INVENTION The present invention has been made in view of the above problems, and can improve the stability of control of the current supplied to an inductive load and simplify the adjustment of circuit constants. It is an object of the present invention to provide a current control method for a PWM inverter and a device therefor which can be performed.

[0026]

According to a first aspect of the present invention, there is provided a PWM inverter current control method, wherein a pulse signal is supplied to a PWM inverter to control a current flowing through an inductive load to control a voltage applied to the inductive load. At this time, the switching element of the PWM inverter is turned on in response to a predetermined pulse signal, and the switching element of the PWM inverter is turned off in response to the current flowing through the inductive load becoming a predetermined value or more. For a predetermined time after switching on the switching element (T +),
This is a method of reducing spike noise superimposed on the detected value of the current flowing through the inductive load.

[0027]

According to a second aspect of the present invention, there is provided a PWM inverter current control device which supplies a pulse signal to the PWM inverter to control a current flowing through the inductive load to control a voltage applied to the inductive load. A frequency limiting means for turning on the switching element in response to a predetermined pulse signal; an off control means for turning off the switching element of the PWM inverter in response to a current flowing through the inductive load exceeding a predetermined value; For a predetermined time after switching on the switching element of the inverter,
Filter means for reducing spike noise superimposed on the detected value of the current flowing through the inductive load.

[0029]

According to a third aspect of the present invention, in the current control device for the PWM inverter, as the frequency control means, a PWM carrier pulse is supplied as a clock signal, and whether or not the current flowing through the inductive load becomes a predetermined value or more. A flip-flop whose comparison result indicating is supplied as a clear input signal is adopted.

According to a fourth aspect of the present invention, there is provided a current control device for a PWM inverter, wherein the filter means is a low-pass filter whose characteristics are changed in response to a switching operation of the PWM inverter.

According to a fifth aspect of the present invention, there is provided a PWM inverter current control apparatus which employs, as the filter means, a low-pass filter whose characteristics are changed by a PWM carrier pulse.

According to a sixth aspect of the present invention, there is provided a current control device for a PWM inverter which employs, as the filter means, a series connection of a capacitor and a parallel connection circuit of a resistor and a switch which is switched by a PWM carrier pulse. .

According to a seventh aspect of the present invention, there is provided a PWM inverter current control device which employs a switched bird reluctance motor as the inductive load.

According to the current control method of the PWM inverter of claim 1, in controlling the voltage applied to the inductive load by supplying the pulse signal to the PWM inverter to control the current flowing in the inductive load, the PWM inverter is controlled. The switching element of is turned on in response to a predetermined pulse signal, and the switching element of the PWM inverter is immediately turned off in response to the current flowing through the inductive load becoming a predetermined value or more, and the switching element of the PWM inverter is turned on. Since the spike noise superimposed on the detected value of the current flowing through the inductive load is reduced for a predetermined time after switching on, it is possible to improve the responsiveness and improve the stability of current control, and to switch the switch. It is possible to prevent thermal destruction of the element due to loss, and also due to spike noise It is possible to prevent malfunction.

[0036]

According to the current control device of the PWM inverter of the third aspect, the frequency limit is applied in controlling the voltage applied to the inductive load by supplying the pulse signal to the PWM inverter to control the current flowing in the inductive load. By means
The switching element of the PWM inverter is turned on in response to a predetermined pulse signal, and the off control means immediately turns off the switching element of the PWM inverter in response to the current flowing through the inductive load becoming a predetermined value or more. The filter means can reduce spike noise superimposed on the detected value of the current flowing through the inductive load for a predetermined time after switching of the switching element of the PWM inverter.

Therefore, it is possible to improve the responsiveness to improve the stability of the current control, prevent thermal destruction of the element due to switch loss, and prevent malfunction due to spike noise. You can

[0039]

According to another aspect of the current control device of the PWM inverter of the present invention, the frequency control means is supplied with a PWM carrier pulse as a clock signal, and the current flowing through the inductive load is equal to or more than a predetermined value. Since the flip-flop to which the comparison result indicating whether or not is supplied is supplied as the clear input signal, the same operation as that of claim 2 can be achieved with a simple configuration.

According to the current control device of the PWM inverter of claim 4, since the low-pass filter whose characteristic is changed in response to the switching operation of the PWM inverter is adopted as the filter means, it is possible to prevent the occurrence of spike noise. The characteristics of the low-pass filter can be changed between when the spike noise is not generated and sufficient attenuation of spike noise and sufficient response when spike noise is not generated can be achieved, and the same as in either claim 2 or claim 3. The action of can be achieved.

According to the current control device of the PWM inverter of claim 5, since the filter means is a low-pass filter whose characteristic is changed by the PWM carrier pulse, a signal for controlling the change of the characteristic is adopted. Since it is not necessary to create the structure, the structure can be simplified and the same operation as that of the fourth aspect can be achieved.

According to the current control device of the PWM inverter of claim 6, as the filter means, a parallel connection circuit of a resistor and a switch that is switched by a PWM carrier pulse is used, and a capacitor is connected in series. Therefore, the configuration of the filter means can be simplified and the same operation as that of claim 4 or 5 can be achieved.

According to the current control device of the PWM inverter of claim 7, since a switched reluctance motor is adopted as the inductive load, low current and large torque can be realized at low cost. It is possible to achieve the same effect as in any one of claims 2 to 6.

[0045]

BEST MODE FOR CARRYING OUT THE INVENTION Embodiments of a current control method for a PWM inverter and an apparatus thereof according to the present invention will be described in detail below with reference to the accompanying drawings.

FIG. 1 is an electric circuit diagram showing an embodiment of an SR motor controller to which the current control method for a PWM inverter according to the present invention is applied. It should be noted that FIG. 1 shows only one phase winding of the inverter and the SR motor.

This SR motor control device is based on the SR motor 1
And a PWM inverter 2 that controls energization to the SR motor 1 and a transistor T- of the PWM inverter 2 that is turned on.
A commutation timing generation unit 3 that outputs a timing signal for controlling the OFF, a current amplitude command generation unit 4 that generates a current amplitude command (torque command), a conduction current of the SR motor 1 and a current amplitude command (torque command). ) As input
The PWM signal generator 5 for outputting the M signal, and the on / off signal for controlling the on / off of the transistor T- of the PWM inverter 2 with the timing signal and the PWM signal from the commutation timing generator 3 as inputs AN
And a D gate 6.

The PWM inverter 2 includes a transistor T + connected to the positive (+) side of the DC line, a transistor T- connected to the negative (-) side of the DC line, and a DC
The freewheeling diode D + is connected in reverse between the + side of the line and the transistor T-, and the freewheeling diode D- is connected in reverse between the transistor T + and the-side of the DC line. Terminal and transistor T
The stator winding of the SR motor 1 is connected between the negative collector terminal and the collector terminal. Further, a current detector 7 for detecting a current flowing through the stator winding of the SR motor 1 is provided.

In order to control the SR motor, the commutation timing generator 3, which outputs the timing for switching the energized phase, and the current amplitude command generator 4, which controls the motor torque or the rotation speed to a predetermined value, are configured as follows. Since it is well known in the art, detailed description will be omitted.

The PWM signal generator 5 has a comparator 51 without hysteresis, in which a current amplitude command (torque command) is supplied to a non-inverting input terminal, and an inverting input of the current detector 7 and the comparator 51. A "1" level signal is constantly supplied to the filter circuit 52 connected to the terminals, the pulse oscillator 53, and the data input terminal D, and the pulse signal (for example, a PWM clock pulse signal) from the pulse oscillator 53 is supplied. It is supplied to the clock input terminal CK and the comparator 5
The output signal from 1 is supplied to the clear input terminal CL,
It has a D flip-flop 54 that outputs a PWM signal from the Q output terminal. Then, the Q output signal becomes "0" level in response to the input signal of the clear input terminal CL becoming "0", and the Q output signal of the data input terminal D in response to the rising of the pulse signal from the pulse oscillator 53. "1"
The level signal is set and its value is output as the Q output signal. In the filter circuit 52, the resistor R and the capacitor C are connected in series, the connection point of the resistor R and the capacitor C is connected to the inverting input terminal of the comparator 51, and
An analog switch 55 is connected in parallel with the resistor R, and the analog switch 55 is controlled by a pulse signal from a pulse oscillator 53 (for example, in response to the pulse signal being at a "0" level, the analog switch 55 is It turns on and shorts both ends of the resistor R).

Next, referring to FIG. 2, the SR having the above-mentioned configuration
The operation of the motor control device will be described. A "1" level signal is output from the commutation timing generator 3, that is,
The case where this phase is selected as the energized phase will be shown.

The D flip-flop 54 is the pulse oscillator 5
A "1" level signal is output from the Q output terminal in response to the rising edge of the pulse signal {see (E) in FIG. 2} from FIG. Then, the "1" level signal and the "1" level signal from the commutation timing generator 3 are supplied to the AND gate 6, so that the transistor T + of the PWM inverter 2 is turned on (see (F) in FIG. 2). },At this time,
Since the transistor T- of the PWM inverter 2 is turned on by the "1" level signal from the commutation timing generator 3, the voltage VDC is applied to the stator winding of the SR motor 1 via both transistors T + and T-. When applied, the winding current rises (see (A) in FIG. 2). At this time, spike noise is generated in the detected current {see (B) in FIG. 2}. However, since the analog switch 55 is off, the resistance R
By a low pass filter consisting of
The spike noise is attenuated to the extent that the switching operation is not adversely affected {see (C) in FIG. 2}. Therefore, it is possible to prevent the output of the comparator 51 from becoming "0" level and the Q output signal of the D flip-flop 54 being cleared (being "0" level) due to spike noise (in FIG. 2). See (D)}.

The filter circuit 52 has a pulse width period of the pulse signal (CK input) from the pulse oscillator 53 (a period during which spike noise is generated, and normally, a period during which spike noise is generated is Only a few μs), and the analog switch 55 during other periods.
As a result, both ends of the resistor R are short-circuited. For this reason, the high frequency components included in the stator winding current can be accurately transmitted to the comparator 51 except during the pulse width period. This allows the control waveform to follow the rectangular wave command.
In other words, it is possible to prevent the disadvantage that the control waveform does not follow the rectangular wave command.

Then, when the voltage VDC is applied to the stator winding of the SR motor 1 through both the transistors T + and T- to increase the winding current and reach the command current, the comparator 51
Output becomes "0" level, and in response thereto, the Q output signal of the D flip-flop 54 also becomes "0" level, and A
Since the output signal from the ND gate 6 also becomes "0" level, the transistor T + is turned off and the stator winding current is attenuated. Also at this time, spike noise due to switching is superimposed on the detection current, an erroneous signal is supplied to the comparator 51, and the erroneous signal shown in (E) of FIG. 2 is output. Since the Q output signal of "1" level is not output until the next rise of the pulse of the CK input, this spike noise has no adverse effect on the current control.

In the above embodiment, the cutoff frequency of the filter circuit 52 is changed by connecting a resistor having a resistance value lower than that of the resistor R (transmitting a higher frequency component) in series with the analog switch 55. You may Further, the same operation as that of the PWM signal generator 5 of the above-described embodiment may be performed by the microcomputer processing.

If the above embodiment is adopted, the on-timing of the transistor T + is determined by the output pulse train of the pulse oscillator 53, so that the on / off frequency of the transistor T + can be set to be equal to or lower than the frequency of this pulse train. Further, the noise removing filter circuit 52 can be operated reliably in synchronization with the spike noise caused by the switching of the transistor T +.
It is possible to reliably prevent malfunction due to spike noise accompanying switching. Further, since the transistor T + is turned on only in response to the pulse train of the pulse oscillator, it is possible to prevent the inconvenience of turning on the transistor T + again due to spike noise generated when the transistor T + is turned off. Therefore, the filter circuit 52 for removing noise has the transistor T
It is sufficient to operate only with the + on-timing.

Although only the case where the SR motor 1 is adopted as the load has been described above, an inductive load other than the SR motor can be adopted.

According to the first aspect of the present invention, the responsiveness can be enhanced to improve the stability of the current control, the thermal destruction of the element due to the switch loss can be prevented, and the spike noise results. There is a unique effect that a malfunction can be prevented.

[0059]

According to the second aspect of the present invention, the responsiveness can be enhanced to improve the stability of the current control, the element can be prevented from being thermally destroyed due to the switch loss, and further, the spike noise can be caused. There is a unique effect that a malfunction can be prevented.

[0061]

The invention of claim 3 has a simple structure.
Has the same effect as.

According to the invention of claim 4, the characteristics of the low-pass filter are changed depending on whether spike noise is generated or not.
Sufficient attenuation of spike noise and sufficient response when spike noise is not generated can be achieved, and the same effect as that of either claim 2 or claim 3 is achieved.

According to the invention of claim 5, there is no need to create a signal for controlling the change of the characteristic, the structure can be simplified, and the same effect as that of claim 4 can be obtained.

The invention of claim 6 can simplify the structure of the filter means, and claim 4 or 5
Has the same effect as.

According to the invention of claim 7, low current and large torque can be realized at low cost.
It has the same effect.

[Brief description of drawings]

FIG. 1 is an electric circuit diagram showing an embodiment of an SR motor control device to which a current control method for a PWM inverter according to the present invention is applied.

FIG. 2 is a diagram showing signal waveforms of respective parts of the SR motor control device of FIG.

FIG. 3 is a diagram illustrating a driving method of an SR motor.

FIG. 4 is an electric circuit diagram showing an inverter main circuit in a conventional SR motor control device.

FIG. 5 is an electric circuit diagram showing a configuration of a waveform control circuit in a conventional SR motor control device.

6 is a diagram showing a signal waveform of each part of the device of FIG. 5 when a current detector having no spike noise is used.

7 is a diagram showing a signal waveform of each part of the device of FIG. 5 when an inexpensive current detector having spike noise is used.

FIG. 8 is a diagram showing a relationship between a command current and an actual current when a low-pass filter that sufficiently attenuates spike noise is always interposed.

[Explanation of symbols]

1 SR motor 2 PWM inverter 51 comparator 52 filter circuit 53 pulse oscillator 54 D flip-flop T + transistor

   ─────────────────────────────────────────────────── ─── Continued front page    (72) Inventor Hiroyuki Yamai               2 of 1000 Otani, Okamoto-cho, Kusatsu-shi, Shiga                 Daikin Industry Co., Ltd. Shiga Works

Claims (7)

(57) [Claims] 1. A method of controlling a voltage applied to a inductive load (1) by supplying a pulse signal to the PWM inverter (2) to control a current flowing through the inductive load (1), comprising: ), The switching element (T +) is turned on in response to a predetermined pulse signal, and the switching element of the PWM inverter is turned off in response to the current flowing through the inductive load (1) becoming a predetermined value or more, A PWM inverter characterized by reducing spike noise superimposed on a detected value of a current flowing through an inductive load (1) for a predetermined time after switching on a switching element (T +) of the PWM inverter (2). Current control method. 2. A device for controlling a voltage applied to the inductive load (1) by supplying a pulse signal to the PWM inverter (2) in order to control a current flowing through the inductive load (1). Frequency limiting means (5) for turning on the switching element (T +) of (4) in response to a predetermined pulse signal.
OFF control means (51) (54) for turning off the switching element (T +) of the PWM inverter (2) in response to the current flowing through the inductive load (1) exceeding a predetermined value. )
And a filter means (52) for reducing spike noise superimposed on the detected value of the current flowing through the inductive load (1) for a predetermined time after switching on the switching element (T +) of the PWM inverter (2). A current control device for a PWM inverter, comprising: 3. The frequency limiting means (54) is supplied with a PWM carrier pulse as a clock signal, and a comparison result indicating whether or not the current flowing through the inductive load (1) has exceeded a predetermined value. Is a flip-flop (54) supplied as a clear input signal. 4. The current control device for a PWM inverter according to claim 2, wherein the filter means (52) is a low-pass filter whose characteristics are changed in response to the switching operation of the PWM inverter (2). . 5. The current control device for a PWM inverter according to claim 4, wherein the filter means (52) is a low-pass filter whose characteristics are changed by a PWM carrier pulse. 6. The filter means (52) has a capacitor (C) connected in series to a parallel connection circuit of a resistor (R) and a switch (55) which is switched by a PWM carrier pulse. The current control device of the PWM inverter according to claim 4 or 5. 7. The current control device for a PWM inverter according to claim 2, wherein the inductive load (1) is a switched reluctance motor (1).