JP3250301B2 - Converter circuit - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a converter circuit for converting a general alternating current into a high-frequency alternating current, applying the converted alternating current to a primary side of a transformer, and extracting a direct current from a rectifier circuit provided on the secondary side of the transformer.

[0002]

2. Description of the Related Art FIG. 7 is a main circuit connection diagram showing a conventional example of a converter circuit. The main circuit of FIG.
Although it is equivalent to the circuit described in No. 53353, its circuit configuration and operation will be described below. First, the first diode 2
And the second diode 3 are connected in series to form a diode series circuit. On the other hand, four sets of switching circuits are configured by connecting a field effect transistor (hereinafter abbreviated as FET) as a semiconductor switch element and a diode in anti-parallel, and the first switching circuit 4 and the second switching circuit 5 Are connected in series to form a first switching series circuit, and the third switching circuit 6 and the fourth switching circuit 7 are connected in series to form a second switching series circuit. These diode series circuit, first switching series circuit and The second switching series circuit is connected in parallel with the second switching series circuit.

A first switching circuit 4 and a second switching circuit 5 are connected to one terminal U1 of a primary winding of a transformer 8.
And the other terminal V1 of the primary winding is connected to
A connection point between the third switching circuit 6 and the fourth switching circuit 7 is connected, and an AC input terminal V is connected to the intermediate tap terminal O1 of the primary winding. Further, an AC input terminal U is connected to a connection point between the first diode 2 and the second diode 3 via the AC reactor 1. The secondary winding of the transformer 8 also has an intermediate tap like the primary winding. The output diode 9 is connected to one terminal U2 of the secondary winding, and the output diode 10 is connected to the other terminal V2. The two cathodes of these diodes are connected in common and connected to a DC output terminal P, and the DC output terminal N is connected to an intermediate tap terminal O2 of the secondary winding. Further, a smoothing capacitor 11 is connected between the DC output terminal P and the DC output terminal N. 8A and 8B are the leakage inductances of the transformer 8.

In the above configuration, for example, AC input terminals U and V
When the first switching circuit 4 and the third switching circuit 6 are turned on when the potential of the AC input terminal U is higher than the potential of the AC input terminal V, the AC input terminal U The input current I _IN flowing from the first diode 2 to the first diode 2 depends on the path of the first switching circuit 4 → the terminal U1 → the leakage inductance 8A → the intermediate tap terminal O1, and the third switching circuit 6 → the terminal V1 → the leakage inductance 8B → the intermediate tap terminal. It shunts to the path of O1 and joins at the intermediate tap terminal O1 to reach the AC input terminal V. At this time, the terminal U1 and the terminal V1 are connected to the first switching circuit 4 and the third
Since a short circuit occurs when the switching circuit 6 is turned on, no voltage is generated in the secondary winding of the transformer 8. Therefore, the voltage between both ends of the AC reactor 1 becomes equal to the AC input voltage.
Input current I _IN flowing in AC reactor 1 increases. The period in this state is referred to as a “short circuit period”.

When the third switching circuit 6 is turned off, the current flowing through the primary winding of the transformer 8 is only the current flowing through the path of the first switching circuit 4 → terminal U1 → leakage inductance 8A → intermediate tap terminal O1. So,
Transformer 8 is energized to induce a voltage in its secondary winding, and output diode 9 conducts to charge smoothing capacitor 11, but with input current I flowing through AC reactor 1 with it.
_IN decreases. The period in this state is referred to as a “voltage generation period”.

When the third switching circuit 6 is turned on in this state, a short-circuit period occurs again, so that no voltage is generated in the secondary winding of the transformer 8, and therefore, the input current I _IN flowing through the AC reactor 1 increases. I do. Here, when the first switching circuit 4 is turned off, the voltage generation period starts again. At this time, the current flowing through the primary winding of the transformer 8 is the third switching circuit 6 → the terminal V1 → the leakage inductance 8B →
Since only the current flowing through the path of the intermediate tap terminal O1 is present, the transformer 8 is excited to induce a voltage in its secondary winding, but this time the output diode 10 conducts and charges the smoothing capacitor 11. At the same time, the input current I _IN flowing through the AC reactor 1 decreases.

As described above, when the potential of the AC input terminal U is higher than the potential of the AC input terminal V, the first switching circuit 4 and the third switching circuit 6 alternately sandwich a period in which both are simultaneously turned on. By repeating ON / OFF, the above-described short-circuit period → voltage generation period → short-circuit period → voltage generation period is repeated, and a high-frequency voltage is applied to the primary winding of the transformer 8 to realize high-frequency insulation conversion. When the potential of the AC input terminal V is higher than the potential of the AC input terminal U, the second switching circuit 5 and the fourth switching circuit 7 are alternately turned on and off alternately with the simultaneous ON period interposed therebetween. Accordingly, a high-frequency voltage is applied to the primary winding of the transformer 8 in the same manner as described above, and high-frequency insulation conversion is realized.

FIG. 8 is a block diagram showing a conventional example of a control circuit for controlling the conventional converter circuit shown in FIG. In the conventional circuit shown in FIG. 8, a current regulator 31 constituted by a proportional-integral calculator inputs a deviation between an input current value to the converter circuit and a separately set current command value. A control signal for making the deviation zero is output. The comparator 33 compares the high-frequency triangular wave signal output from the triangular wave generator 32 with the output signal of the current adjuster 31, and outputs a logic H signal during a period when the output signal of the current adjuster 31 is larger than the triangular wave signal. , And conversely, the current regulator 31
During the period when the triangular wave signal is larger than the output signal, the logic L signal is output. The pulse distribution circuit 34 controls each FET of the switching circuit so that the output of the comparator 33 is in the above-described short circuit period when the output is a logical H signal, and is in the voltage generation period when the output is a logical L signal. On / off is determined, and each of these FETs is driven via the drive circuit 35.

For example, when the input current value is smaller than the current command value, the output increases since the input deviation of the current regulator 31 is positive. Therefore, the period during which the output of the current controller 31 exceeds the triangular wave signal (ie, the comparator 33
(A period during which a logic H signal is output) becomes longer, so that a short-circuit period becomes longer. As the short-circuit period becomes longer, the input current increases and approaches the current command value. Conversely, if the input current is larger than the current command value, the current controller 31
Is decreased, and the period during which the comparator 33 outputs the logic L signal becomes longer, so that the voltage generation period becomes longer, the input current decreases, and approaches the current command value.
This is called a carrier comparison pulse width modulation and is a well-known control method. By controlling the time ratio between the short-circuit period and the voltage generation period in this manner, the input current I _IN flowing through the AC reactor 1 can be in the same phase as the AC input voltage and a sine wave. Power factor of almost 100
%.

[0010]

During the short circuit period of the conventional circuit shown in FIG. 7, the first switching circuit 4 and the third switching circuit 6 share the input current I _IN by half. That is, the current I _U flowing through the first switching circuit 4 is
I _U = I _IN / 2, and the current I _V flowing through the third switching circuit 6 is also I _V = I _IN / 2. Here, when the first switching circuit 4 is turned off after the transition from the short circuit period to the voltage generation period, the current I _U flowing through the first switching circuit 4 rapidly decreases from I _IN / 2 to zero. At the same time, the current I flowing through the third switching circuit 6
_V increases rapidly from I _IN / 2 to I _IN . At this time, in the leakage inductance 8A, a jump voltage V _8A is generated in the direction shown to prevent the sudden decrease of the current, and at the same time, in the leakage inductance 8B, the jump voltage V _8B is generated to prevent the sudden increase of the current.
In the direction shown. At this time, since the diode 5D conducts and the second switching circuit 5 is in the ON state,
Since the voltage of the sum of the jump voltages V _8A and V _8B is applied to both ends of the first switching circuit 4 that is turned off, the first switching circuit 4 may be broken by an overvoltage.

In order to prevent such a switching circuit from being damaged by overvoltage, (a) the switching circuit is composed of elements having a high withstand voltage. (b) Slow the current change rate, that is, reduce the switching speed of the switching circuit. In the former case, the device becomes large and expensive, so the price of the device increases.In the latter case, the switching loss increases, the heat generation of the device increases, and the efficiency decreases. There is.

An object of the present invention is to suppress an overvoltage caused by a leakage inductance of a transformer constituting the converter circuit when the converter circuit operates.

[0013]

In order to achieve the above object, a converter circuit according to the present invention comprises: a diode series circuit in which a first diode and a second diode are connected in series;
A first switching series circuit in which a second switching circuit having the same configuration is connected in series to a first switching circuit configured by anti-parallel connection of a semiconductor switching element and a diode, and a third switching circuit and a fourth switching circuit having the same configuration The second switching series circuit connected in series is connected in parallel with each other, one end of a transformer primary winding having an intermediate tap is connected to an intermediate connection point of the first switching series circuit, and the first primary winding of the transformer is connected. The other end and an intermediate connection point of the second switching series circuit are connected, and an AC power supply is connected to an intermediate connection point of the diode series circuit and an intermediate tap of the transformer primary winding via an AC reactor, The first switching circuit and the third switching circuit alternately turn on and off alternately with a period during which both circuits are simultaneously turned on; When the switching circuit and the fourth switching circuit are alternately turned on and off alternately with a period during which both circuits are simultaneously turned on, the output signal of the current adjusting means for matching the input current from the AC power supply with the current command value is used. Correspondingly, by changing the time ratio between ON and OFF, the AC from the AC power supply is converted into a high-frequency AC and applied to the primary winding of the transformer, and the rectifier connected to the secondary winding of the transformer is converted. In a converter circuit configured to take out a direct current from a circuit, a capacitor is connected in parallel to the diode series circuit, or a snubber circuit including a diode, a resistor, and a capacitor is connected in parallel instead of the capacitor, or The first switching circuit and the second switching circuit are connected to each other when one of the switching circuits is on, with both being off at the same time. The third switching circuit and the fourth switching circuit are also turned off, and when one of the switching circuits is on with a period in which both are off at the same time, the other switching circuit is turned off, or The output side of the current adjusting means is provided with a limiter whose limit value changes according to the magnitude of the input current to the converter circuit.

[0014]

According to the present invention, a capacitor is connected in parallel to a diode series circuit, or a snubber circuit composed of a diode, a resistor and a capacitor is connected in parallel, so that a transformer is connected with the on / off operation of the switching circuit. The current flowing through the leakage inductance continues to flow through the capacitor or the snubber circuit to the leakage inductance, so that the rate of change of the current flowing through the leakage inductance is smaller than before, thereby reducing the jump voltage. And the duty of the switching circuit is eased. In addition, by limiting the output value of the current regulator by a limiter provided on the output side of the current regulator, the maximum voltage of the capacitor or the maximum voltage of the capacitor constituting the snubber circuit is obtained for all circuit conditions. The value is suppressed to further reduce the maximum voltage applied to the switching circuit.

[0015]

FIG. 1 is a main circuit connection diagram showing a first embodiment of the present invention, which corresponds to claim 1 or claim 2.
In the circuit of the first embodiment shown in FIG.
First diode 2, second diode 3, first switching circuit 4, second switching circuit 5, third switching circuit 6, fourth switching circuit 7, transformer 8 (including leakage inductances 8A and 8B), output diode 9 , Output diode 10 and smoothing capacitor 11 have the same names, uses and functions as those of the conventional circuit described above with reference to FIG.

In the circuit of the first embodiment, a snubber capacitor 21 is connected in parallel to a diode series circuit formed by connecting a first diode 2 and a second diode 3 in series. Although different, in a first switching series circuit formed by connecting a first switching circuit 4 and a second switching circuit 5 in series,
An FET 4F constituting the switching circuit 4;
When one of the FETs 5F constituting the switching circuit 5 is on, the other is always off, and the operation is performed so that both are not turned on at the same time in order to prevent a so-called arm short circuit. Third switching circuit 6
And the fourth switching circuit 7
In the same manner as described above, the FET 7F that operates as described above also performs an operation such that when one is on, the other is off while preventing the occurrence of arm short circuit.

By the above operation, during the short-circuit period in which the first switching circuit 4 and the third switching circuit 6 are on, the first diode 2 → the first switching circuit 4 →
Terminal U1 → leakage inductance 8A → middle tap terminal O
When the current flowing through the leakage inductance 8A in the path 1 shifts to a voltage generation period in which the first switching circuit 4 is turned off, the first diode 2 → the snubber capacitor 2
1 → Diode 5D → Terminal U1 → Leakage inductance 8
Since the current continues to flow in the path from A to the intermediate tap terminal O1, the rate of change of the current flowing through the leakage inductance 8A is significantly smaller than in the conventional case. As the rate of change of the current decreases, the rate of change of the current flowing through the leakage inductance 8B also decreases. At this time, the energy stored in the leakage inductances 8A and 8B is transferred to the snubber capacitor 21 and the terminal voltage thereof is increased.
Has a value equal to the terminal voltage of the snubber capacitor 21.

As described above, when the FET 4F forming the first switching circuit 4 is turned off, the FET 5F is turned on after a short short-circuit prevention time elapses, so that the second switching circuit 5 is also conductive to the forward current. I do. Therefore, the electric charge transferred to the snubber capacitor 21 is discharged through the path of the snubber capacitor 21 → the third switching circuit 6 → the terminal V1 → the primary winding → the terminal U1 → the second switching circuit 5 → the snubber capacitor 21.
Is stored as a DC output via the transformer 8.
When transitioning from the voltage generation period to the short circuit period again, first
After the ET5F is turned off, the FET 4F is turned on after the elapse of the arm short circuit prevention time.

Similarly, when the switching circuits other than those described above are turned off from the short-circuiting period to shift to the voltage generation period, the energy of the leakage inductance is similarly reduced by the snubber capacitor 21.
When the other switching circuit connected in series with the turned off switching circuit is turned on, the energy transferred to the snubber capacitor 21 is released to the DC output side via the transformer 8. The period when the other switching circuit in the series is ON is referred to as a “regeneration period”.

[0020] Figure 2 is a waveform diagram showing a change in current and voltage of the snubber capacitor 21 in the first embodiment the circuit of FIG. 1, FIG. 2 is a change in the snubber capacitor current I _S, 2 snubber capacitor each change in voltage V _S represents. Since leakage inductances 8A and 8B are present in series in the charging / discharging path of snubber capacitor 21, if the capacitance of snubber capacitor 21 is C and the series inductance of leakage inductances 8A and 8B is L, the current waveform is periodic. Is a resonance waveform with T as
This cycle T is represented by the following equation 1.

[0021]

## EQU1 ## When the voltage generation period exceeds the period T due to T = 2π · (LC) ^1/2, there is a point in time when the voltage of the snubber capacitor 21 rises again. When the voltage generation period ends at this point, the voltage of the snubber capacitor 21 is held at a high value, and if this is repeated, the voltage of the snubber capacitor 21 may exceed the allowable value. Therefore, in order to prevent such a phenomenon from occurring, it is necessary to manage the relationship between the regeneration period and the resonance frequency, such as making the regeneration period shorter than the cycle T.

FIG. 3 is a main circuit connection diagram showing a second embodiment of the present invention, which corresponds to claim 4 or claim 5. The circuit shown in FIG. Reactor 1, first diode 2, second diode 3, first switching circuit 4, second switching circuit 5, third switching circuit 6, fourth switching circuit 7, transformer 8 (including leakage inductances 8A and 8B), The names, applications, and functions of the output diode 9, the output diode 10, and the smoothing capacitor 11 are the same as those of the conventional circuit described above with reference to FIG.

In the circuit of the second embodiment shown in FIG. 3, a snubber circuit 22 is formed by connecting a snubber capacitor 25 in series to a parallel connection circuit of a snubber diode 23 and a snubber resistor 24.
This snubber circuit 22 is connected in parallel to a diode series circuit (that is, a series connection circuit of the first diode 2 and the second diode 3). With this circuit configuration, when the converter circuit shifts from the short-circuit period to the voltage generation period, the energy stored in the leakage inductances 8A and 8B is transferred to the snubber capacitor 25 via the snubber diode 23, and then this When discharging the energy of the snubber capacitor 25 through the primary winding of the transformer 8,
Since the snubber resistor 24 is inserted into the discharge path, no vibration that may occur in the circuit of the first embodiment shown in FIG. 1 occurs. Therefore, if the regeneration period is sufficiently long, the terminal voltage of the snubber capacitor 25 becomes equal to the voltage between the terminals U1 and V1 of the transformer 8, that is, the electromotive force 2E of the transformer 8. Therefore, if the regeneration period is sufficiently long, the end point of the regeneration period can be arbitrarily determined.

[0024] Figure 4 is a waveform diagram showing a change in current and voltage of the snubber capacitor 25 in the second embodiment circuit of Figure 3, Figure 4 is the change in the snubber capacitor current I _S, 4 snubber capacitor Although each variation of the voltage V _S represents no vibrations occur, as described above. FIG. 5 is a block diagram showing a third embodiment of the present invention.
It is a block diagram regarding Claim 3 or Claim 6. In the circuit of the third embodiment shown in FIG. 5, a current controller 31, a triangular wave generator 32, a comparator 33, a pulse distribution circuit 3
4, and the names, applications, and functions of the drive circuit 35 are the same as those of the circuit of the conventional example described above with reference to FIG.

If the regeneration period is shorter than the cycle T, the discharge path is cut off before the discharge of the snubber capacitor 21 is completed, and therefore, as shown in the operation waveform diagram of FIG. is of the snubber capacitor voltage V _S is higher than the initial value. If the regenerative period is shorter in this way, the snubber capacitor voltage V _S continues to increase, but on the other hand, the difference voltage between the electromotive force 2E of the transformer 8 and the voltage increases as the V _S increases. Since the discharge current increases, the snubber capacitor voltage V _S is balanced at a certain value and becomes a constant value. This balance value increases as the cutoff current increases, that is, as the input current I _IN increases, or as the regeneration period becomes shorter.

Since the converter circuit is generally controlled so that the input current has a sine waveform with a power factor of 100%,
The instantaneous value of the input current is always proportional to the instantaneous value of the input voltage. Further, at this time, the instantaneous value of the input voltage is substantially proportional to the length of the voltage generation period. Therefore, since the input current and the length of the voltage generation period, that is, the length of the regeneration period, are in a proportional relationship, the snubber capacitor voltage V _S is maintained at a substantially constant value. However, in a transient state such as the AC input voltage fluctuates rapidly since the above-mentioned proportional relationship is not established, it may snubber capacitor voltage V _S becomes excessive.

Therefore, in the circuit of the third embodiment shown in FIG.
A variable limiter 41 is provided at the next stage of the current regulator 31 to prevent the snubber capacitor voltage V _{S from} becoming excessive. The variable limiter 41 is for limiting the output value of the current regulator 31 to a certain value or less, and changes such that the limit value is increased when the input current is small, and the limit value is decreased when the input current is large. Has a function. Therefore, for example, since the upper limit of the output value of the current regulator 31 given to the subsequent stage when the input current is small is large, the short-circuit period is long,
Therefore, the regeneration period can be shortened. Conversely, when the input current is large, the limit value of the variable limiter 41 is small, and the output value of the current regulator 31 transmitted to the subsequent stage is limited to a small value, so that the short-circuit period is short, and thus the regeneration period is long. Become.

[0028] Thus, regardless of whether it is a transient state, to ensure the regeneration period corresponding to the input current value by the above-described operation can be prevented from increasing snubber capacitor voltage V _S. Therefore, it is possible to configure the switching circuit with relatively low withstand voltage elements. Note that the current controller 31 may be constituted by, for example, a proportional adjuster or a proportional-integral-differential adjuster, instead of the proportional-integral adjuster.
The setting of the limit value at 1 is performed by setting a value proportional to the input current from a constant (or a value proportional to the square or a value proportional to the square root) as a limit value, an inverse proportional value of the input current as a limit value, or the like. Various setting methods can be adopted. Since the variable limiter 41 does not operate in a normal state, there is no possibility that the control performance when controlling the input current is adversely affected.

FIG. 6 is a block diagram showing a fourth embodiment of the present invention, similar to the circuit of the third embodiment described above with reference to FIG. In the circuit of the fourth embodiment shown in FIG. 6, a current controller 31, a triangular wave generator 32, a comparator 33, a pulse distribution circuit 34,
The names, applications, and functions of the drive circuit 35 are the same as those of the conventional circuit described above with reference to FIG.

In the circuit of the fourth embodiment shown in FIG.
Instead of 1, a limiter 42 having a constant limit value and an adder 4 for subtracting an input current value from an output value of the limiter 42
3 is different from the circuit of the third embodiment described above with reference to FIG. Normally, the current controller 31 outputs a value obtained by adding the amount subtracted by the adder 43 so that the input current value matches the current command value. Therefore, in the transient state, if the short-circuit period is lengthened even if the input current value is large, that is, if the output of the current regulator 31 is to be increased, the output of the current regulator 31 is limited by the limiter 42. Is subtracted by the adder 43, so that the input value to the comparator 33 becomes smaller. Therefore, the circuit of the fourth embodiment shown in FIG.
This is equivalent to the circuit of the third embodiment described above.

[0031]

According to the present invention, since the snubber capacitor or the snubber circuit including the snubber capacitor is connected in parallel to the diode series circuit, the operation of the converter circuit shifts from the short-circuit period to the voltage generation period. At this time, the energy stored in the leakage inductance of the transformer constituting the converter circuit is transferred to the snubber capacitor, and then, when the operation of the converter circuit shifts to the regeneration period, the electric charge stored in the snubber capacitor is transferred to the transformer. And discharges the energy to the DC output side through the primary winding. Therefore, the jump voltage caused by the energy stored in the leakage inductance is suppressed from becoming excessive, so that the withstand voltage of the semiconductor switch element constituting the switching circuit can be reduced, and the device can be reduced in size and cost. In addition, since the stored energy of the leakage inductance is recovered and output to the DC side, the effect of improving the efficiency of the device and reducing the amount of generated heat can be obtained. Further, by providing a limiter for limiting the output of the current regulator, the effect of avoiding the possibility of the snubber capacitor voltage increasing can be obtained.

[Brief description of the drawings]

FIG. 1 is a main circuit connection diagram showing a first embodiment of the present invention.

FIG. 2 is an operation waveform diagram showing changes in current and voltage of a snubber capacitor 21 in the circuit of the first embodiment of FIG.

FIG. 3 is a main circuit connection diagram showing a second embodiment of the present invention.

FIG. 4 is an operation waveform diagram showing changes in current and voltage of snubber capacitor 25 in the circuit of the second embodiment in FIG. 3;

FIG. 5 is a block diagram showing a third embodiment of the present invention.

FIG. 6 is a block diagram showing a fourth embodiment of the present invention.

FIG. 7 is a main circuit connection diagram showing a conventional example of a converter circuit.

8 is a block diagram showing a conventional example of a control circuit for controlling the conventional converter circuit shown in FIG.

[Explanation of symbols]

 DESCRIPTION OF SYMBOLS 1 AC reactor 2 1st diode 3 2nd diode 4 1st switching circuit 5 2nd switching circuit 6 3rd switching circuit 7 4th switching circuit 8 Transformer 8A, 8B Leakage inductance 9,10 Output diode 11 Smoothing capacitor 21,25 Snubber capacitor 22 Snubber circuit 23 Snubber diode 24 Snubber resistor 31 Current regulator 32 Triangular wave generator 33 Comparator 34 Pulse distribution circuit 35 Drive circuit 41 Variable limiter 42 Limiter 43 Adder

Claims (6)

(57) [Claims] 1. A second switching circuit having the same configuration is serially connected to a diode series circuit having a first diode and a second diode connected in series and a first switching circuit having an anti-parallel connection of a semiconductor switch element and a diode. And a second switching circuit in which a third switching circuit and a fourth switching circuit having the same configuration are connected in series.
A switching series circuit is connected to each other in parallel, one end of a transformer primary winding having an intermediate tap is connected to an intermediate connection point of the first switching series circuit, and the other end of the transformer primary winding is connected to the other end. (2) connecting an intermediate connection point of the switching series circuit to an intermediate connection point of the diode series circuit and an intermediate tap of the transformer primary winding via an AC reactor to connect an AC power supply to the first switching circuit; And the third switching circuit alternately turns on and off with the simultaneous ON period of both circuits alternately, or the second switching circuit and the fourth switching circuit alternate with the simultaneous ON period of both circuits alternately When repeating on / off,
The AC from the AC power supply is converted to a high-frequency AC by changing the time ratio of the ON and OFF in accordance with the output signal of the current adjusting means for matching the input current from the AC power supply to the current command value. In the converter circuit, which is applied to the primary winding of the transformer and extracts DC from a rectifier circuit connected to the secondary winding of the transformer, by connecting a capacitor in parallel to the diode series circuit, A converter circuit characterized in that a jump voltage generated in a primary winding of a transformer when alternately turning on and off from a simultaneous on period is reduced. 2. The converter circuit according to claim 1, wherein said first switching circuit and said second switching circuit are turned on simultaneously when one of said switching circuits is on with a period in which both are off. Is turned off, and the third switching circuit and the fourth switching circuit are also turned off when one of the switching circuits is on with a period in which both are off at the same time. 3. The converter circuit according to claim 1, wherein an output side of said current adjusting means is provided with a limiter whose limit value changes according to the magnitude of an input current to said converter circuit. A converter circuit characterized by the above. 4. A second switching circuit having the same configuration is connected in series to a diode series circuit having a first diode and a second diode connected in series and a first switching circuit having an anti-parallel connection of a semiconductor switching element and a diode. And a second switching circuit in which a third switching circuit and a fourth switching circuit having the same configuration are connected in series.
A switching series circuit is connected to each other in parallel, one end of a transformer primary winding having an intermediate tap is connected to an intermediate connection point of the first switching series circuit, and the other end of the transformer primary winding is connected to the other end. (2) connecting an intermediate connection point of the switching series circuit to an intermediate connection point of the diode series circuit and an intermediate tap of the transformer primary winding via an AC reactor to connect an AC power supply to the first switching circuit; And the third switching circuit alternately turns on and off with the simultaneous ON period of both circuits alternately, or the second switching circuit and the fourth switching circuit alternate with the simultaneous ON period of both circuits alternately When repeating on / off,
The AC from the AC power supply is converted to a high-frequency AC by changing the time ratio of the ON and OFF in accordance with the output signal of the current adjusting means for matching the input current from the AC power supply to the current command value. In a converter circuit configured to supply the primary winding of the transformer and take out a direct current from a rectifier circuit connected to a secondary winding of the transformer, a snubber circuit is configured by connecting a capacitor in series with a parallel circuit of a diode and a resistor. By connecting this snubber circuit in parallel with the diode series circuit, the jump voltage generated in the primary winding of the transformer when the switching circuit is turned on and off alternately during the simultaneous ON period is reduced. A converter circuit characterized by the following. 5. The converter circuit according to claim 4, wherein said first switching circuit and said second switching circuit are turned on simultaneously when one of said switching circuits is on with a period in which both are off. Is turned off, and the third switching circuit and the fourth switching circuit are also turned off when one of the switching circuits is on with a period in which both are off at the same time. 6. The converter circuit according to claim 4, wherein an output side of said current adjusting means is provided with a limiter whose limit value changes according to the magnitude of an input current to said converter circuit. A converter circuit characterized by the above.