JP3146094B2 - Microwave semiconductor circuit - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a microwave semiconductor circuit such as a semiconductor phase shifter or a semiconductor modulator in which loss fluctuation when the reactance of a reflection circuit or the like is switched between high and low is reduced.

[0002]

2. Description of the Related Art FIG. 25 shows, for example, CLCHEN et.al:
“A Low-Loss Ku-Band Monolithic Analog Phase Shif
ter ”, IEEE Trans., Vol. MTT-35, No. 3, pp. 315-320 (Ma
r.1987) is an equivalent circuit diagram showing the configuration of a conventional reflection type phase shifter of a microwave semiconductor circuit. In the figure, a varactor diode 4 having one end grounded is connected to each of two distribution terminals 2 and 3 of the hybrid circuit 1. A bias is applied to the varactor diode 4 from the outside, but a bias circuit therefor is omitted here.

Next, the operation of the reflection type phase shifter shown in FIG. 25 will be described. The signal incident from the input terminal 5 of the hybrid circuit 1 is equally distributed and appears at the distribution terminals 2 and 3,
The light is reflected by the varactor diode 4 provided at each distribution terminal and appears at the output terminal 6. At this time, when the bias applied to the varactor diode 4 is changed, the impedance exhibited by the varactor diode 4 changes and functions as an impedance variable circuit. As a result, the phase of the signal reflected at the varactor diode 4 and appearing at the output terminal 6 changes, and the device operates as a phase shifter.

FIG. 26 shows the reflection coefficient exhibited by the varactor diode 4 in order to explain the above operation in more detail. The varactor diode 4 is used by applying a reverse bias. Here, for example, the applied bias voltage is -V ₁ , -V ₂ (where V ₁ , V ₂ > 0,
| −V ₁ | <| −V ₂ |). As the magnitude of the reverse bias voltage is increased, the varactor diode 4
In contrast, when the magnitude of the reverse bias voltage is reduced, the capacitance and resistance of the varactor diode 4 are increased. Therefore, when the applied bias voltage is -V _1, the varactor diode 4 is low reactance, become highly resistive, the reflection coefficient gamma ₁ exhibited by the varactor diode 4 at this time is shown in FIG. 26. here,
θ ₁ is a Γ ₁ of phase. In general, in the case of a series circuit of reactance and resistance, as the reactance value decreases, the voltage applied to the resistance relatively increases.As a result, even if the resistance value is the same, the loss increases as the reactance value decreases, and the reflection loss increases. And the absolute value of the reflection coefficient decreases. In the varactor diode 4, since the reactance value decreases and the resistance increases, the absolute value of the reflection coefficient decreases more remarkably.

On the other hand, when the applied bias voltage is −V ₂ , the varactor diode 4 has a high reactance and a low resistance, so that the reflection coefficient Γ ₂ is advanced in phase from Γ ₁ and has a large absolute value. FIG. 26 shows the reflection coefficient Γ ₂ exhibited by the varactor diode 4 at this time. Here, θ ₂ is Γ ₂ of phase. Therefore, the phase of the reflection coefficient exhibited by the varactor diode 4 is changed to θ by the change of the applied bias.
_1, theta since ₂ and can be varied, by choosing the applied bias voltage appropriately, the gamma ₁ phase theta ₁ and gamma ₂ phase theta ₂ with the required phase shift with the difference Φ phase shifter 1 And the amount.

Next, another microwave semiconductor circuit will be described. FIG. 27 shows, for example, C.ANDRICOS et.al: “C-
Band 6-Bit GaAs Monolithic Phase Shifter ”, IEEE Tr
ans., Vol. MTT-33, No. 12, pp. 1591-1596 (Dec. 1985), is an equivalent circuit diagram showing a configuration of a conventional loaded line type phase shifter of a microwave semiconductor circuit. . Branch lines 8 are connected to two points on the main line 7 at approximately 1/4 wavelength intervals, and the other end of each branch line 8 is grounded via a field effect transistor 9.

Next, the operation of the loaded line type phase shifter shown in FIG. 27 will be described. FIG. 28 is a sectional view of FIG.
FIG. 4 is an equivalent circuit diagram when a bias voltage of 0 V equal to the ground potential is applied to the gates of two field effect transistors 9. In this case, the resistance between the drain and source of the field effect transistor 9 is low and the branch line 8 has a low impedance.
It can be considered that a low-impedance resistor is loaded at the tip of the. On the other hand, FIG. 29 is an equivalent circuit diagram when the bias voltage applied to the field effect transistor 9 is switched and a negative bias voltage lower than the pinch-off voltage is applied. In this case, between the drain and the source of the field effect transistor 9, a capacitive high impedance is exhibited, and the branch line 8
It can be considered that a high-impedance capacity is loaded at the tip of the. Here, if the electrical length of the branch line 8 is selected to be slightly longer than １ ／ wavelength, the action of the impedance conversion of the branch line 8 causes the main line 7 in the case shown in FIG.
Is loaded with a capacitive impedance including a resistance, and in the case shown in FIG. 29, the main line 7 is loaded with an inductive impedance including no resistance. As described above, the loaded line type phase shifter of the conventional configuration changes the impedance presented between the drain and the source by changing the bias voltage applied to the gate of the field effect transistor 9, and changes the branch line 8 of the main line 7. By simultaneously changing the impedance of the field-effect transistor 9 from the connection point of the field-effect transistor 9 from capacitive to inductive, or vice versa, it operates as a phase shifter that changes the phase of the radio wave propagating through the main line 7.

Next, another microwave semiconductor circuit will be described. FIG. 30 shows, for example, Takayama, Higo: “3.8G
FIG. 4 is an equivalent circuit diagram showing a configuration of a conventional phase modulator shown in “Hz band 4W 4-phase phase modulator”, MW73-108 (1973), IEICE, Microwave Research Society.
A PIN diode 12 is connected to one terminal of the circulator 10 via a lossless conversion circuit 11.

Next, a case where the phase modulator of FIG. 30 operates as a two-phase phase modulator will be described as an example. 31 and 32 show an equivalent circuit when a forward current is applied to the PIN diode 12 and an equivalent circuit when a reverse voltage is applied to the PIN diode 12.
Are shown below. When a forward current is applied, the PIN diode 12 exhibits a resistive low impedance, and when a reverse voltage is applied, the PIN diode 12 exhibits a capacitive high impedance. Therefore, the phase difference of the reflected wave from the PIN diode 12 side at the connection point between the lossless conversion circuit 11 and the PIN diode 12 becomes slightly smaller than 180 degrees. Here, for example, if an inductance is used as the lossless conversion circuit 11 and its value is appropriately selected, P
As a result, the phase delay of the IN diode 12 in the low impedance state becomes larger than the phase delay of the PIN diode 12 in the high impedance state. As a result, the phase difference of the reflected wave from the lossless conversion circuit 11 side at the connection terminal 33 of the reflection circuit of the circulator 10. Can be set to 180 degrees and operates as a two-phase modulator.

[0010]

SUMMARY OF THE INVENTION Conventional semiconductor phase shifters,
Since a microwave semiconductor circuit such as a semiconductor modulator is configured as described above, there is a difference in an absolute value of a reflection coefficient when a bias applied to a semiconductor configuring a reflection circuit or a branch circuit is switched. Alternatively, there is a difference in resistance when the reactance is switched between capacitive and inductive, so that loss fluctuation occurs. As a result, for example, in a phased array antenna using a semiconductor phase shifter, the side lobe level becomes high, causing deterioration of antenna performance,
Further, in a transmitter using a semiconductor modulator, there is a problem that distortion occurs in a modulated wave, which causes an increase in a signal error rate and an increase in unnecessary wave radiation.

SUMMARY OF THE INVENTION The present invention has been made to solve the above-mentioned problems, and a semiconductor phase shifter which reduces a loss fluctuation when a reactance of a variable impedance circuit constituting a reflection circuit or a branch circuit is switched between high and low. An object is to obtain a microwave semiconductor circuit such as a semiconductor modulator.

[0012]

To achieve the above object, a microwave semiconductor circuit according to any one of the first to sixth aspects of the present invention is provided for changing the phase amount between two predetermined distribution terminals of a hybrid circuit. Variable impedance circuit
In the connection configuration, each distribution terminal is connected
Variable impedance circuit, reactance and resistance
A plurality of series circuits are provided, and impedance
A configuration that switches the reactance and resistance of the column circuit
And multiple series circuits are switched between each other and compared
And relatively high reactance and high resistance, and
Configuration in which low reactance and low resistance are connected in series
It is obtained by the.

According to a seventh aspect of the present invention, there is provided a microwave semiconductor circuit comprising: a main line; a branch circuit connected to two points on the main line approximately １ ／ wavelength apart from each other;
An end connected to a different end from the end connected to the main line
Each of the above impedances is provided with a variable impedance circuit.
The variable circuit is a field-effect transistor or field-effect transistor.
Providing multiple series circuits of transistors and capacitors
The impedance change depends on the resistance of the field effect transistor.
The configuration is such that the resistance value is switched.
The comparison between the switching partners
Field effect transformer with a relatively high resistance due to
In this configuration, the transistor is connected to the capacitor .

The microwave semiconductor circuit of the invention according to claim 8 is provided with an impedance variable circuit using a semiconductor element as a reflection circuit connected to a predetermined terminal of the circulator and connected to the predetermined terminal. Impedance
Variable circuit consists of a field effect transistor and impedance
Provide multiple series circuits with elements and change impedance
Turns on / off the field effect transistor of the series circuit
And the plurality of series circuits are switched with each other.
Relatively high reactance and high resistance
Resistance and low reactance and low resistance in series
It is configured to be connected .

[0015]

According to the present invention, there is provided the above-mentioned structure.
In the microwave semiconductor circuit according to the invention according to the present invention, when the impedance variable circuit using the semiconductor element has a reflection circuit provided at each of two predetermined distribution terminals of the high-bridge circuit, when the impedance variable circuit exhibits low reactance, the impedance variable circuit has a low resistance component. When the impedance variable circuit has a higher reactance than the reactance, the variable impedance circuit has a higher resistance than the resistance, and the absolute value of the reflection coefficient when the reactance of the variable impedance circuit is switched. Can be made equal, loss fluctuation can be eliminated.

Further, in the microwave semiconductor circuit according to the present invention, the semiconductor element provided between the other end of the branch line connecting one end to each of two points on the main line at approximately 1/4 wavelength intervals and the ground is provided. When the used impedance variable circuit exhibits a low reactance, the impedance variable circuit has a low resistance component, and when presenting a high reactance compared to the reactance, the impedance variable circuit is higher than the resistance component. By having the resistance component and equalizing the power loss in the resistance component when the reactance of the variable impedance circuit is switched, the loss variation can be eliminated.

In the microwave semiconductor circuit according to the present invention, when the impedance variable circuit using a semiconductor element constituting a reflection circuit provided at a predetermined terminal of the circulator exhibits a low reactance, the impedance variable circuit is used. Has a low resistance component, and when presenting a high reactance compared to the reactance, the variable impedance circuit has a high resistance component compared to the above resistance component, and is used when switching the reactance of the variable impedance circuit. By making the absolute value of the reflection coefficient equal, fluctuation in loss can be eliminated.

[0018]

[Embodiment 1] FIG. 1 is an equivalent circuit diagram of a reflection type phase shifter according to a first embodiment of the present invention. In the figure, two distribution terminals 2 and 3 of the hybrid circuit 1
The variable impedance circuit which constitutes the reflection circuit provided in each case is a resonance inductor 19a between the drain and the source.
Are connected in parallel, and two circuits each having a series circuit of a resistor and a capacitor, one end of which is connected to the source of the field effect transistor and the other end is grounded, are configured in parallel. The value of the resistance of one of the circuits connected in parallel (here, the resistance 15 and the resistance Ra
) Is smaller than the resistance value of the other circuit (here, the resistance 17 and the resistance value Rb), and the value of the capacitor 16a of the same circuit (here, the capacitor 16a
And the capacitance value is Ca) larger than the capacitance value of the capacitor of the other circuit (here, the capacitance value is 18a and the capacitance value is Cb).

Next, the operation of the reflection type phase shifter shown in FIG. 1 will be described. In the following description, for simplicity, it is assumed that the two field effect transistors 13a and 14a of the variable impedance circuit forming the reflection circuit have the same characteristics. First,
A bias voltage of 0 V equal to the ground potential is applied to the gate of one field effect transistor 13a of the variable impedance circuit provided at each of the two distribution terminals of the hybrid circuit 1, and the gate of the other field effect transistor 14a has a voltage higher than the pinch-off voltage. FIG. 2 shows an equivalent circuit when a low negative bias voltage is applied. In this case, the drain-source of the field-effect transistor 13a exhibits a low resistance impedance (hereinafter, the state of the field-effect transistor is referred to as an ON state), and the field-effect transistor 14
a presents a capacitive high impedance between the drain and source (hereinafter, the state of this field effect transistor is referred to as OF
F state). The drain-source impedance of the field effect transistor 13a is sufficiently low, but always with a drawing to their close of resistor R _1. On the other hand, by setting the value of the resonance inductor 19a connected between the drain and source of the field effect transistor 14a so as to resonate in parallel with the capacitance between the drain and source,
Between the sources can be considered almost open.

FIG. 3 shows a simplified equivalent circuit when considered as described above. Each of the distribution terminals 2 and 3 has a reflection circuit including a series circuit of the resistor R ₁ (resistance when the field-effect transistor is ON), the resistor 15 and the capacitor 16a. Figure 4 illustrates a reflection coefficient gamma ₁ reflection circuit exhibits in this case. Here, Z ₀ the impedance viewed hybrid circuit side from the distribution terminal of the hybrid circuit 1, the angular frequency omega, the impedance viewed reflected circuit side from the distribution terminal of the hybrid circuit 1 as Z _1, gamma ₁ is expressed by the following equation Make it appear. Γ ₁ = (Z ₁ −Z ₀ ) / (Z ₁ + Z ₀ ) Z ₁ = (R ₁ + Ra) −j / (ωCa) In this case, the reactance of the capacitor 16a of the reflection circuit is equal to that of the other capacitor 18a. Since the reactance is lower than the reactance presented, the delay θ _{1 of the} reflection phase exhibited by this reflection circuit is larger than the delay θ _{2 of the} reflection phase exhibited by the other reflection circuit. Further, the influence of the resistance R _1, Ra, the absolute value of the gamma ₁ is has decreased somewhat in comparison with the case of the capacitor 16a alone, that reduced due to R _1, Ra is small, small.

Next, when the bias applied to the gate of one of the field effect transistors 13a of the reflection circuit and the bias applied to the gate of the field effect transistor 14a are exchanged with those described above, the distribution terminals 2, 3 Includes a resistor R ₁ (resistance when the field effect transistor 14a is in an ON state), a resistor 17 and a capacitor 18a, respectively.
And a reflection circuit composed of a series circuit of In FIG.
In this case, the reflection coefficient Γ ₂ exhibited by the reflection circuit is calculated by the aforementioned Γ ₁
Shown together with As in the case of gamma _1, the impedance viewed hybrid circuit side from the distribution terminal of the hybrid circuit 1 Z _0, the angular frequency omega, the hybrid circuit 1
The impedance looking into the reflection circuit side from the distribution terminal as Z _2, gamma ₂ is expressed by the following equation. Γ ₂ = (Z ₂ −Z ₀ ) / (Z ₂ + Z ₀ ) Z ₂ = (R ₁ + Rb) −j / (ωCb) Here, assuming that Ca> Cb and Ra <Rb, | Γ ₁ | = | Γ
_The element constants are determined so that ₂ |. This allows
In this case, the reactance exhibited by the capacitor 18a of the reflection circuit is larger than the reactance exhibited by the other capacitor 16a. Therefore, the delay θ _{2 of the} reflection phase exhibited by this reflection circuit is delayed by the delay θ _{1 of the} reflection phase exhibited by the other reflection circuit.
Less than. Further, a large resistance Rb from the resistor Ra so connected in series, can be reduced to be equal to the absolute value of the gamma ₂ to the absolute value of the gamma _1.

As a result, the bias applied to the gate of the field effect transistor 13a and the bias applied to the gate of the field effect transistor 14a constituting the reflection circuit connected to the two distribution terminals 2 and 3 of the hybrid circuit 1, respectively. Thus, a reflection type phase shifter having no loss fluctuation and switching between the reflection phases θ ₁ and θ ₂ to obtain a required phase shift amount Φ can be obtained.

Embodiment 2 FIG. FIG. 5 is an equivalent circuit diagram of a reflection type phase shifter according to a second embodiment of the present invention. The variable impedance circuit constituting the reflection circuit provided at each of the two distribution terminals 2 and 3 of the hybrid circuit 1 is composed of field effect transistors 13b and 14b having resonance inductors 19b and 19c provided between the drain and the source, and the above-mentioned sources. capacitor 16b is grounded and the other end connected to one end, constituted by two parallel-connected circuit having a 18b, the exhibits when the field effect transistor 13b of the one circuit of the parallel connection to a low impedance state resistance The value is determined to be smaller than the resistance exhibited when the field effect transistor 14b of the other circuit is set to the low impedance state, and the same value (Cc) of the capacitor 16b of the one circuit as that of the capacitor 18b of the other circuit is set. It is set to a large value (Cd).

Next, the operation of the reflection type phase shifter shown in FIG. 5 will be described. Unlike the first embodiment described above, the two field effect transistors 13a and 14a of the variable impedance circuit constituting the respective reflection circuits connected to the two distribution terminals 2 and 3 of the hybrid circuit 1 are not the same characteristics but have different characteristics. Use something. FIG. 6 is an equivalent circuit diagram for explaining the operation of the reflection type phase shifter of FIG. The field effect transistor 13b of the variable impedance circuit constituting the reflection circuit of FIG.
4B shows a case where the switch 4b is turned off. As in the first embodiment, the equivalent circuit is further simplified as shown in FIG. In the second embodiment, if the resistance value exhibited when the field effect transistor 13b is turned on is selected to be the sum of R1 and Ra in the _first embodiment, the reflection coefficient of the reflection circuit at this time is selected. Can be made the same as Γ1 in the _first embodiment.

On the other hand, the field effect transistor 1 of the reflection circuit
A bias applied to the gate of 3b, by replacing the case and bias the applied to the gate of the field effect transistor 14b, and the reflection coefficient of the reflection circuit in this case, the same as the gamma ₂ of Example 1 above can do. In this way, by using the resistance exhibited by the field effect transistor 13b and the resistance exhibited by the field effect transistor 14b as resistors for equalizing the reflection coefficient, a function equivalent to that of the phase shifter described in the first embodiment can be obtained. There is an advantage that the phase shifter can be realized with a small number of elements.

Embodiment 3 FIG. FIG. 8 is an equivalent circuit diagram of a reflection type phase shifter according to a third embodiment of the present invention. In the first and second embodiments, the case where the inductor is connected between the drain and the source of the field effect transistor so as to resonate in parallel with the capacitance between the drain and the source has been described. However, since a resonance inductor element is required and an operation band is narrowed because parallel resonance is used, as shown in FIG. 8, the configuration is such that no resonance inductor is provided, and between the drain and source of the field effect transistors 13c and 14c. By utilizing the capacitance as a part of the capacitance constituting the reflection circuit, there is an advantage that the number of elements can be reduced and the band can be widened.

Embodiment 4 FIG. FIG. 9 is an equivalent circuit diagram of a reflection type phase shifter according to a fourth embodiment of the present invention. The same reason as in Example 3 above,
That is, as shown in FIG. 9, an impedance variable circuit constituting a reflection circuit is replaced with a first field effect transistor 13 in order to avoid band narrowing by utilizing parallel resonance.
d, 14d and a capacitor 16d, 1 having one end connected to the source of the first field effect transistor and the other end grounded.
8d and two second field effect transistors 22a and 23a having a drain connected to the source of the first field effect transistor and a source grounded in parallel with the capacitor, and two circuits connected in parallel. I have. A resistance value that is exhibited when the first field-effect transistor of one circuit of the parallel connection is set to a low impedance state is smaller than a resistance value that is exhibited when the first field-effect transistor of the other circuit is set to a low impedance state. And the value of the capacitor of one circuit is set to a value larger than that of the capacitor of the other circuit. Here, the bias applied to the field effect transistors 22a and 23a loaded in parallel to the capacitors 16d and 18d, respectively, is alternately set to 0V, so that the cutoff characteristics can be favorably obtained, and there is an advantage that the band can be widened.

Embodiment 5 FIG. FIG. 10 is an equivalent circuit diagram of a reflection type phase shifter according to a fifth embodiment of the present invention. In the fourth embodiment, as shown in FIG. 10, a matching inductor 24 composed of a distributed constant line or a lumped constant element is used.
By providing a and 24b to improve the matching, there is an advantage that the band can be further widened.

Embodiment 6 FIG. FIG. 11 is an equivalent circuit diagram of a loaded line type phase shifter according to a sixth embodiment of the present invention. A main line, a branch line 8 having one end connected to two points at approximately 1/4 wavelength intervals on the main line, and an impedance variable circuit using a semiconductor element between the other end of the branch line and ground. When looking at the variable impedance circuit side from the connection point of the branch line and the variable impedance circuit, when the variable impedance circuit exhibits a low reactance, the variable impedance circuit has a low resistance component, and when the variable impedance circuit exhibits a high reactance compared to the reactance. The variable impedance circuit has a higher resistance than the above resistance.

Next, the operation of the loaded line type phase shifter of FIG. 11 will be described. FIG. 12 shows an equivalent circuit in the case where the field-effect transistor 25a shown in FIG. 11 is turned on and the field-effect transistor 26a is turned off. The drain-source impedance of the field effect transistor 25a is sufficiently low, but always with a drawing to their close of the resistor R _2. Further, the above-mentioned field effect transistor 2
Inductor 19e connected between drain and source of 6a
Is set so that it resonates in parallel with the drain-source capacitance, the drain-source can be regarded as almost open. The resistor R ₂ is resonant inductor 19d
, The effect of the resonance inductor 19d can be neglected. FIG. 13 shows an equivalent circuit when the above is considered. In the branch line 8,
So that the resistor R ₂ is connected.

On the other hand, FIG. 1 shows an equivalent circuit when the applied bias is switched to turn on the field effect transistor 26a and turn off the field effect transistor 25a.
It is shown in FIG. By setting the value of the inductor 19e connected between the drain and the source of the field effect transistor 25a so as to resonate in parallel with the capacitance between the drain and the source, the drain and the source can be regarded as substantially open. FIG. 15 is a diagram showing an equivalent circuit when the above is considered. The branch line 8 includes a field effect transistor 26
a connecting third capacitor 27a through the resistor R ₃ exhibited by. Here, the magnitudes of the resistors R ₂ and R ₃ , that is, the field effect transistor 25a and the field effect transistor 26a
Preparative By appropriately choosing, it is possible to equalize the power consumed by the resistor R ₂ and R ₃ in FIGS. 13 and 15 above. Therefore, it is possible to eliminate the loss fluctuation when the bias applied to the field effect transistor 25a and the field effect transistor 26a is switched to change the load impedance on the main line 7 between inductive and capacitive.

Embodiment 7 FIG. FIG. 16 is an equivalent circuit diagram of a loaded line type phase shifter according to a seventh embodiment of the present invention. By using the drain-source capacitance of the field-effect transistors 25b and 26b as a part of a circuit element that changes the impedance as a configuration without the resonance inductor, there is an advantage that the size and the bandwidth can be reduced.

Embodiment 8 FIG. FIG. 17 is an equivalent circuit diagram of a phase modulator according to an eighth embodiment of the present invention. FIG.
7, the variable impedance circuit forming the reflection circuit provided at the predetermined terminal 33 of the circulator 10 includes a field effect transistor 28a, a reflection inductor 30a having one end connected to the source of the field effect transistor 28a and the other end grounded. And a circuit comprising a field-effect transistor 29a and a capacitor 31a having one end connected to the source of the field-effect transistor 29a and the other end grounded.

FIG. 18 shows one field-effect transistor 2
8a is turned on, and the other field-effect transistor 29
FIG. 4 is an equivalent circuit diagram when a is turned off. The reactance exhibited by the reflected inductor 30a and a resistor R ₄ exhibited by the field effect transistor 28a on the one ON state, the field effect transistor 29 on the other in the OFF state
If the reactance a and the capacitor 31a are lower than each other, the equivalent circuit is approximately represented as shown in FIG.

FIG. 20 shows that the applied bias is switched from the state shown in FIG.
FIG. 9 is an equivalent circuit diagram in a case where the FF state is set and the field-effect transistor 29a is set to an ON state. Reactance exhibiting a resistor R ₅ and capacitor 31a exhibited by the field effect transistor 29a is found if lower than the reactance exhibiting a field effect transistor 28a of the reflection inductor 30a, the equivalent circuit is represented as approximately 21.

Next, the case where the above configuration example is operated as a two-phase phase modulator will be described in more detail.
FIG. 22 is a characteristic diagram showing reflection coefficients Γ ₃ and Γ ₄ in the above two states of the loading circuit when the field effect transistor side is viewed from the circulator of the phase modulator of FIG. In the state shown in FIG. 19, the reflection inductor 30a
位相₃ phase delay θ
₃ is greater than 180 degrees. Further, the influence of the resistance R ₄ exhibited by the field effect transistor 28a in the ON state,
The absolute value of the gamma ₃ is although a slight decrease in comparison with the case of the reflection inductors 30a alone, that reduced because R ₄ is small is small. On the other hand, in the state shown in FIG. 21, since the reactance exhibited by the capacitor 31a is large, gamma ₄ phase delay theta ₄ is little. The resistor R ₅ is the resistance R ₄ exhibited by the field effect transistor 29a which is turned ON
So that the two field effect transistors 2
8a and 29a, the capacitor 31a
Since a relatively large resistance is connected in series, the absolute value of the gamma ₄ has been reduced to the absolute value and the same degree of gamma _3. Therefore, by switching between the bias applied to the gate of the field effect transistor 28a and the bias applied to the gate of the field effect transistor 29a, the loss is kept constant, and the phase θ ₃ and the phase θ ₄ are switched to 0-π Can be realized.

Embodiment 9 FIG. FIG. 23 is an equivalent circuit diagram of a phase modulator according to a ninth embodiment of the present invention. In the eighth embodiment, the case where the field effect transistor is directly connected to the circulator 10 has been described. However, the present invention is not limited to this, and as shown in FIG.
In this case, there is an advantage that the degree of freedom of phase setting can be increased and more accurate phase modulation can be performed.

Embodiment 10 FIG. FIG. 24 is an equivalent circuit diagram of a phase modulator according to the tenth embodiment of the present invention.
In the ninth embodiment, a line is used as a circuit for converting the reflection phase. However, as shown in FIG. 24, there is an advantage that the circuit can be downsized by forming the circuit with a lumped constant element.

[0039]

As described above, according to the first to fifth aspects of the present invention, loss can be reduced by equalizing the absolute value of the reflection coefficient when the reactance of the variable impedance circuit constituting the reflection circuit is switched between high and low. A microwave semiconductor circuit such as a semiconductor phase shifter with reduced fluctuation can be obtained.

According to the seventh aspect of the present invention, the semiconductor phase shifter in which the loss fluctuation is reduced by equalizing the power loss in the resistance when the reactance of the variable impedance circuit constituting the branch circuit is switched between high and low. A microwave semiconductor circuit such as a vessel can be obtained.

According to the eighth aspect of the present invention, the semiconductor phase shifter in which the loss fluctuation is reduced by equalizing the absolute value of the reflection coefficient when the reactance of the variable impedance circuit constituting the reflection circuit is switched between high and low. Thus, a microwave semiconductor circuit such as a semiconductor modulator can be obtained.

[Brief description of the drawings]

FIG. 1 is an equivalent circuit diagram of a reflection type phase shifter according to a first embodiment of the present invention.

FIG. 2 is an equivalent circuit diagram for explaining the operation of the reflection type phase shifter of FIG.

FIG. 3 is a simplified equivalent circuit diagram for explaining the operation of the reflection type phase shifter of FIG. 1;

FIG. 4 is a characteristic diagram showing a reflection coefficient of a reflection circuit of the reflection type phase shifter of FIG. 1;

FIG. 5 is an equivalent circuit diagram of a reflection type phase shifter showing a second embodiment of the invention according to claim 3;

FIG. 6 is an equivalent circuit diagram for explaining the operation of the reflection type phase shifter of FIG.

FIG. 7 is a simplified equivalent circuit diagram for explaining the operation of the reflection type phase shifter of FIG. 5;

FIG. 8 is an equivalent circuit diagram of a reflection type phase shifter according to a third embodiment of the present invention.

FIG. 9 is an equivalent circuit diagram of a reflection type phase shifter according to a fourth embodiment of the present invention.

FIG. 10 is an equivalent circuit diagram of a reflection type phase shifter according to a fifth embodiment of the present invention.

FIG. 11 is an equivalent circuit diagram of a loaded line type phase shifter according to a sixth embodiment of the present invention.

12 is an equivalent circuit diagram illustrating an operation of the loaded line type phase shifter of FIG.

FIG. 13 is a simplified equivalent circuit diagram illustrating the operation of the loaded line phase shifter of FIG.

FIG. 14 is an equivalent circuit diagram for explaining the operation of the loaded line type phase shifter of FIG.

FIG. 15 is a simplified equivalent circuit diagram for explaining the operation of the loaded line type phase shifter of FIG. 11;

FIG. 16 is an equivalent circuit diagram of a loaded line type phase shifter according to a seventh embodiment of the present invention.

FIG. 17 is an equivalent circuit diagram of a phase modulator according to an eighth embodiment of the present invention.

18 is an equivalent circuit diagram for explaining the operation of the phase modulator shown in FIG.

FIG. 19 is a simplified equivalent circuit diagram for explaining the operation of the phase modulator of FIG.

20 is an equivalent circuit diagram illustrating an operation of the phase modulator of FIG.

FIG. 21 is a simplified equivalent circuit diagram for explaining the operation of the phase modulator of FIG.

FIG. 22 is a characteristic diagram illustrating a reflection coefficient of a loading circuit of the phase modulator of FIG. 17;

FIG. 23 is an equivalent circuit diagram of a phase modulator according to a ninth embodiment of the present invention.

FIG. 24 is an equivalent circuit diagram of a phase modulator according to a tenth embodiment of the present invention.

FIG. 25 is an equivalent circuit diagram showing a configuration of a conventional reflection type phase shifter.

26 is a characteristic diagram illustrating the operation of the reflection type phase shifter of FIG.

FIG. 27 is an equivalent circuit diagram showing a configuration of a conventional loaded line type phase shifter.

FIG. 28 is an equivalent circuit diagram for explaining the operation of the loaded line type phase shifter of FIG. 27;

FIG. 29 is an equivalent circuit diagram for explaining the operation of the loaded line phase shifter of FIG. 27;

FIG. 30 is an equivalent circuit diagram showing a configuration of a conventional phase modulator.

FIG. 31 is an equivalent circuit diagram illustrating an operation of the phase modulator of FIG. 30.

FIG. 32 is an equivalent circuit diagram illustrating the operation of the phase modulator of FIG. 30.

[Explanation of symbols]

DESCRIPTION OF SYMBOLS 1 Hybrid circuit 2 Distribution terminal 3 Distribution terminal 4 Varactor diode 5 Input terminal 6 Output terminal 7 Main line 8 Branch line 9 Field effect transistor 10 Circulator 11 Lossless conversion circuit 12 PIN diode 13a, 13b, 13c Field effect transistor 13d, 13e Electric field Effect transistor 14a, 14b, 14c, 14d, 14e Field effect transistor 15 Resistance 16a, 16b, 16c, 16d, 16e Capacitor 17 Resistance 18a, 18b, 18c, 18d, 18e Capacitor 19a, 19b, 19c, 19d, 19e Resonance inductor 22a, 22b Field effect transistor 23a, 23b Field effect transistor 24a, 24b Matching inductor 25a, 25b Field effect transistor 26a, 26b Field effect transistor Dielectric 27a, 27b Capacitor 28a, 28b, 28c Field effect transistor 29a, 29b, 29c Field effect transistor 30a, 30b, 30c Reflective inductor 31a, 31b, 31c Capacitor 32 Reflection phase conversion line 33 Reflection circuit connection terminal

────────────────────────────────────────────────── ─── Continued on the front page (51) Int.Cl. ⁷ Identification code FI H04L 27/36 H04L 27/00 F (72) Inventor Naoki Takagi 5-1-1, Ofuna, Kamakura-shi Mitsubishi Electric Corporation Electronic system In the laboratory (56) References JP-A-3-49401 (JP, A) JP-A-2-60319 (JP, U) JP-A-61-188328 (JP, U) Patent 2943480 (JP, B2) (58) Field surveyed (Int.Cl. ⁷ , DB name) H01P 1/185 H01C 3/10 H03H 7/20 H03H 11/20 H03H 11/46 H04L 27/36

Claims (8)

(57) [Claims] An impedance variable to change a phase amount to each of two predetermined distribution terminals of a hybrid circuit.
In the configuration where the circuits are connected, the impedance
The transformer circuit is composed of multiple series circuits of reactance and resistance.
And the impedance change depends on the reactance of the series circuit.
And the series circuit switches each value of the resistor and the resistor.
And relatively high reactance and high resistance, and
Configuration in which low reactance and low resistance are connected in series
Microwave semiconductor circuit, characterized in that the the. 2. A plurality of components constituting an impedance variable circuit.
Are connected between the drain and source for resonance.
A field effect transistor with an inductor, a resistor, and a key.
A series connection circuit of capacitors, and each of the above series connection circuits whose impedance is to be changed
Is compared with the other series connection circuit.
In the case of low resistance, the above capacitor is a large capacity capacitor,
2. The microwave semiconductor circuit according to claim 1, wherein the resistor is a small-capacity capacitor . 3. A plurality of variable impedance circuits constituting a variable impedance circuit.
Are connected between the drain and source for resonance.
Field effect transistor with inductor and capacitor
The series connection circuit whose impedance is to be changed
Field-effect transistor compared to one series connection circuit
If the resistance in the ON state is relatively low,
The capacitor should be a large capacity capacitor.
2. The microwave semiconductor circuit according to claim 1, wherein the microwave semiconductor circuit is a sita. 4. A plurality of variable impedance circuits constituting a variable impedance circuit.
Series circuits are each a field-effect transistor and a capacitor.
The series connection circuit of the capacitor and the series connection circuit whose impedance is to be changed
Field-effect transistor compared to the
If the resistance in the ON state is relatively low,
Use a large capacity capacitor for the capacitor and a small capacity capacitor for high resistance.
2. The microwave semiconductor circuit according to claim 1, wherein the microwave semiconductor circuit is a capacitor . 5. A plurality of variable impedance circuits constituting a variable impedance circuit.
Series circuits are first field-effect transistors, respectively.
And a second field-effect transistor and a capacitor
And the series connection circuit whose impedance is to be changed
The first field-effect transistor as compared with the other series-connected circuit.
If the resistance when the transistor is on is relatively low,
The capacitor is a large-capacity capacitor, and a high-capacity capacitor is
2. The microwave semiconductor circuit according to claim 1, wherein the microwave semiconductor circuit is a quantity capacitor . 6. A plurality of variable impedance circuits constituting a variable impedance circuit.
Series circuit constitutes part of series connection or parallel circuit
Connected a matching inductor in series with the
The microwave semiconductor circuit according to claim 5, wherein: 7. A main line, a branch circuit connected to the two points away schematic quarter-wave on the main line, the respective branch circuits
Connected to a different end than the one connected to the main line
Each of the above impedance variable circuits is provided with a field effect transistor.
Or a series connection of a field-effect transistor and a capacitor.
Circuit, and impedance change is based on the electric field
The resistance value of the effect transistor is switched, and the plurality of field effect transistors are mutually switched.
In comparison between the above, the resistance value by the above switching is relatively high
Field effect transistor with resistance connected to capacitor
Microwave semiconductor circuit, characterized in that where the structure is. 8. A variable impedance circuit connected to a predetermined terminal of a circulator and using a semiconductor element as a reflection circuit, wherein the variable impedance circuit connected to the predetermined terminal is provided between a field effect transistor and an impedance element. Provide a plurality of series circuits , and change the impedance
To switch on / off the field effect transistor on the circuit
And the series circuits are switched between each other and compared
And relatively high reactance and high resistance, and
Configuration in which low reactance and low resistance are connected in series
Microwave semiconductor circuit, characterized in that the the.