JP3129947B2 - Digital quadrature detector - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a digital quadrature detection device used in digital radio communication, and more particularly to a digital quadrature detection device for reducing the amount of operation of a digital filter constituting the device.

[0002]

2. Description of the Related Art A conventional digital quadrature detector is shown in FIG.
As shown in FIG. 2, an IF signal input terminal 1 to which a reception signal S (t) converted to an intermediate frequency (IF) is input, and a reception IF
An A / D converter 2 for sampling the signal S (t) at a sampling frequency fs and converting the signal to a digital signal;
The output sample value sequence S (nTs) of the D converter 2 (where T
s = 1 / fs) at the local frequency fs / 4 to perform quadrature detection. The multipliers 3 and 4 multiply the local signals, and the outputs Xi (nTs) and Xq (nTs) of the multipliers 3 and 4 are band-limited. Receive FIR low-pass filters 17 and 18 and receive FIR
It has an in-phase output terminal 8 and a quadrature output terminal 9 for outputting the outputs of the IR low-pass filters 17 and 18 as baseband signals.

In this digital quadrature detector, the reception F
In order to reduce the operation amount of the IR low-pass filters 17 and 18, the center frequency f _{IF of} the input IF signal S (t) and the A / D converter 2
F _IF = (4k + 1) fs / 4 (k = 0,1,2, ‥) (1) so that the sampling frequency fs of the A / D converter 2 is The local frequency of the quadrature detector is set to fs / 4.

In this case, the input IF signal S (t) (= I
_{(T) cos2πf IF t + Q} (t) sin2πf IF t)
Is converted by the A / D converter 2 into an output sample value sequence S (nTs) represented by the following equation (2).

S (nTs) = I (nTs) cos (πn / 2) + Q (nTs) sin (πn / 2) (2) Then, this signal is distributed with the frequency fc of the following equation (3) as a center frequency. .

Fc = mfs ± (fs / 4) (m = 0, ± 1, ± 2, ‥) (3) As an example, when f _IF = 5fs / 4 (k = 1 in equation (1)) FIGS. 7A and 7B show the frequency spectra of the input and output of the A / D converter 2 at the time.

On the other hand, in the quadrature detector, a multiplier 3 is used to perform quadrature detection on S (nTs) at a local frequency fs / 4.
Multiplies S (nTs) by cos (πn / 2) at
In the multiplier 4, S (nTs) is replaced by sin (πn / 2).
(N = 0, ± 1, ± 2, ‥). The local signal cos (πn / 2) of this local frequency fs / 4, s
in (πn / 2) is (1,0, -1,0), (0,1,
Considering that the sequence is a periodic sequence having one cycle of (0, -1), the outputs Xi (nTs) and Xq (nT
s) is expressed by the following equations (4) and (5), and has a form in which 0 is interpolated every other sample.

Xi (nTs) = S (nTs) cos (πn / 2) = I (nTs) (1 + (− 1) ⁿ ) / 2 (4) Xq (nTs) = S (nTs) sin (πn / 2) ) = Q (nTs) (1 − (− 1) ⁿ ) / 2 (5) As a result, the output of the quadrature detector has a sampling frequency equivalent to fs / 2. Further, the center frequency of the output of the quadrature detection unit is obtained by multiplication by the multipliers 3 and 4 so that fc ± (fs
/ 4), and mfs and mfs ± (fs / 2) (m =
0, ± 1, ± 2, ‥). FIG. 7C shows the frequency spectrum of the output of the quadrature detector.

Next, mfs ± (fs / 2) (m = 0, ± 2) corresponding to the second harmonic component included in the output of the quadrature detector.
In order to suppress the components of (1, ± 2, ‥), band limitation is performed using the reception FIR low-pass filters 17 and 18. As a result, baseband I and Q signals are obtained as shown in FIG.

At this time, the reception FIR low-pass filter 1
If the number of taps 7 and 18 is L, the number of product-sum operations per output sample is reduced to L / 2 because the sampling frequency of the input signal is fs / 2.

As described above, in the conventional digital quadrature detector, the baseband signal is detected from the input IF signal while reducing the amount of calculation in the reception FIR low-pass filter.

[0012]

However, in the conventional digital quadrature detector, the center frequency f _IF
And the sampling frequency fs of the A / D converter is required to satisfy the relationship of Expression (1). If the relationship deviates from this condition, a frequency offset occurs or a quadrature detection unit causes a frequency offset. There is a problem that the 0 interpolation effect cannot be obtained and the amount of calculation in the reception FIR low-pass filter increases.

The present invention solves such a conventional problem, and reduces the amount of calculation by a reception FIR low-pass filter with respect to an input IF signal having an arbitrary center frequency while reducing a baseband signal. It is an object of the present invention to provide a digital quadrature detector that can be obtained with high accuracy.

[0014]

Therefore, according to the present invention, an input IF signal is sampled at a sampling frequency fs, A / D converted into a sample value signal having a center frequency fc, and then subjected to quadrature detection to obtain a baseband signal. In the digital quadrature detector, the A / D converted sample value is
quadrature detection at a local frequency corresponding to the minimum value of fc |
Quadrature detection means for generating in-phase output and quadrature output
Frequency mfs + (f
s / 2) (m = 0, ± 1, ± 2, ‥)
A first and a second low-pass filter for suppressing the frequency component;
And the output sample value sequence of the second low-pass filter
Thinning out every sample, set sampling frequency to fs / 2
First and second decimation processing means for converting, and first and second decimation processing means;
2 for limiting the band of the output of the thinning processing means
And a second reception low-pass filter.

[0015]

[0016]

[0017]

[Action] center frequency when the f _IF input IF signal to the A / D conversion at a sampling frequency fs, the center frequency of the signal after A / D conversion is fc = ((mD ± N) / D) fs ( where
D and N are positive integers, and m = 0, ± 1, ± 2, ‥). The A / D-converted signal is converted to the local value of the minimum value of | fc |
Orthogonal transform by frequency and sample value sequence after orthogonal transform
Decimation processing after passing through a simple low-pass filter
To convert the sampling frequency to fs / 2,
Reduce the number of product-sum operations in filter processing performed later to half
Reduce. This simple low-pass filter uses the input IF signal
If interfering waves are mixed in the signal, the signal
Works to suppress interference wave components that may overlap
You. Therefore, even if the IF signal contains interfering waves,
Filter with a small number of calculations and high-precision baseband signals.
No. can be obtained.

[0018]

[0019]

[0020]

[0021]

【Example】

(First Embodiment) As shown in FIG. 1, a digital quadrature detection apparatus according to a first embodiment includes an IF signal input terminal 1 to which a received IF signal S (t) having a center frequency _fIF is input, and an IF signal input terminal 1. An A / D converter 2 that samples at a sampling frequency fs and converts it to a sample value signal, and performs quadrature detection on the A / D converted sample value sequence S (nTs) (Ts = 1 / fs) at a local frequency fs / 4 Cos (πn
/ 2), sin (πn / 2) (n = 0, ± 1, ± 2,
‥), receiving FIR band-pass filters 5 and 6 for band-limiting the outputs of the multipliers 3 and 4,
A phase compensator 7 for compensating for a frequency offset of the outputs of the reception FIR bandpass filters 5 and 6; a compensation function memory 19 for supplying a compensation function to the phase compensator 7;
And an in-phase output terminal 8 and a quadrature output terminal 9 for outputting the in-phase output and the quadrature output as baseband signals.

The I input to this digital quadrature detector is
The center frequency f _IF of the F signal S (t) is generally represented by the following equation.

F _IF = (N / D) fs (N, D: positive integer) (6) At this time, the output sample value sequence S (nT
s) is distributed with the frequency fc of the following equation (7) as the center frequency.

Fc = mfs ± f _IF = ((mD ± N) / D) fs (m = 0, ± 1, ± 2, ‥) (7) As an example, an operation when N = 9 and D = 8 Will be described. At this time, the equations (6) and (7) are replaced by the following equations (8) and (9).
The frequency spectra of the input and output of the A / D converter 2 are as shown in FIGS. 3 (a) and 3 (b).

F _IF = (9/8) fs (8) fc = (m ± (9/8)) fs (m = 0, ± 1, ± 2, ‥) (9) At this time, S (nTs) Is represented by the following equation (10).

S (nTs) = I (nTs) cos (2π (fs / 8) nTs) + Q (nTs) sin (2π (fs / 8) nTs) (10) In the quadrature detector, this S (nTs) is Local frequency f
To perform quadrature detection at s / 4, the multiplier 3 outputs the local signal cos (2π (fs / 4) nTs) = cos (πn /
2) is multiplied by S (nTs), and the multiplier 4 generates a local signal sin (2π (fs / 4) nTs) = sin (πn /
2) multiply S (nTs) by This cos (πn /
2), sin (πn / 2) is (1, 0, −1, 0),
Considering a periodic sequence having (0, 1, 0, -1) as one cycle, the outputs Xi (nTs), X
q (nTs) has a form in which 0 is interpolated every other sample, and is expressed by the following equation.

Xi (nTs) = S (nTs) cos (πn / 2) = I (nTs) (cos (2π (fs / 8) nTs) + cos (2π (3fs / 8) nTs)) / 2 + Q (nTs) ) (Sin (2π (3fs / 8) nTs) −sin (2π (fs / 8) nTs)) / 2 (11) Xq (nTs) = S (nTs) sin (πn / 2) = I (nTs) ( sin (2π (3fs / 8) nTs) + sin (2π (fs / 8) nTs)) / 2 + Q (nTs) (cos (2π (fs / 8) nTs) −cos (2π (3fs / 8) nTs)) / 2 (12) The center frequency of the output of the quadrature detector becomes fc ± (fs / 4) by the multiplication in the multipliers 3 and 4, and mfs + (11fs /
8) (= mfs + (3fs / 8)), mfs + (7fs
/ 8), mfs- (7fs / 8), mfs- (11fs
/ 8) (= mfs + (5fs / 8)) (m = 0, ± 1,
± 2, ‥), and the frequency spectrum of the output of the quadrature detector is as shown in FIG. Also, the sampling frequency is equivalently fs / 2. As shown in the figure, a frequency offset Δf expressed by the following equation occurs in the output of the quadrature detector.

Δf = | fc− (fs / 4) | == fs / 8 (13) Next, mfs + (3fs / 8) corresponding to the second harmonic component included in the output of the quadrature detector, mfs + (5fs / 8)
(M = 0, ± 1, ± 2, ‥)
Received FIR bandpass filters 5, 6 with center frequency Δf
To limit the bandwidth.

As a result, the frequency spectrum shown in FIG. 3D is obtained. At this time, if the number of taps of the reception FIR bandpass filters 5 and 6 is L taps, the number of product-sum operations per output sample is reduced to L / 2 since the sampling frequency of the input signal is fs / 2. Is done.

On the other hand, this filtering processing is performed by the equation (1)
This corresponds to the removal of the 3fs / 8 component in 1) and (12), and the filter outputs Yi (nTs) and Yq (nT
s) is represented by the following equation.

Yi (nTs) = R (nTs) cos (2π (fs / 8) nTs + φ (nTs)) / 2 (14) Yq (nTs) = R (nTs) sin (2π (fs / 8) nTs + φ (NTs)) / 2 (15) R (nTs) = (I (nTs) ² + Q (nTs) ² ) ^1/2 (16) φ (nTs) = tan ⁻¹ (Q (nTs) / I (nTs) (17) The phase compensator 7 performs the operation as shown in the following equations (18) and (19) to thereby obtain the reception FIR bandpass filters 5 and 6.
To compensate for the frequency offset Δf = fs / 8 occurring in the output. Note that cos (nπ / 4), s
in (nπ / 4) is supplied from the compensation function memory 19.
Therefore, the compensation function memory 19 stores a plurality of compensation function values cos (nπ / 4) and sin (nπ
/ 4) is stored.

Ie (nTs) = Yi (nTs) cos (nπ / 4) + Yq (nTs) sin (nπ / 4) (18) Qe (nTs) = − Yi (nTs) sin (nπ / 4) + Yq (nTs) ) Cos (nπ / 4) (19) As a result, a baseband signal is obtained as shown by the following equations (20) and (21).

Ie (nTs) = R (nTs) cos (φ (nTs)) / 2 = I (nTs) / 2 (20) Qe (nTs) = R (nTs) sin (φ (nTs)) / 2 = Q (nTs) / 2 (21 ) Thus, the digital quadrature detection device of the first embodiment, the input IF signal, regardless of the value of the center frequency f _IF, quadrature detection at a local frequency fs / 4 , Equivalently converting the sampling frequency to fs / 2, and
The amount of calculation of the bandpass filter is reduced. And
A high-precision baseband signal is obtained by compensating the frequency offset of the filter output by the phase compensator.

(Second Embodiment) The digital quadrature detector of the second embodiment compensates for the phase error contained in the filter output after performing baseband differential detection.

This device, as shown in FIG.
Output sample value sequence Yi of R bandpass filters 5 and 6
(NTs), Yq (nTs) (Ts = 1 / fs), a symbol discrimination point detection unit 10 for detecting symbol discrimination point values Yi (nT), Yq (nT) (T: symbol period), and a symbol discrimination check. The cosine ID of the modulation phase difference from the output of the output unit 10
(NT) and a baseband delay detector 11 for generating a sine QD (nT), and an output ID of the baseband delay detector 11
(NT), frequency offset included in QD (nT)}
A phase compensator 12 for compensating a phase error θe (= 2π △ fT) caused by f, and a compensation coefficient memory 20 for supplying a compensation coefficient to the phase compensator 12 are provided. Other configurations are first
There is no difference from the device of the embodiment.

This apparatus performs quadrature detection in the operation procedure described in the first embodiment, and filters 5 and 6 are given by the following equation (14).
Yi (nTs) and Yq (nTs) represented by (17)
Is output as a filter output.

Yi (nTs) = R (nTs) cos (2π (fs / 8) nTs + φ (nTs)) / 2 (14) Yq (nTs) = R (nTs) sin (2π (fs / 8) nTs + φ (NTs)) / 2 (15) R (nTs) = (I (nTs) ² + Q (nTs) ² ) ^1/2 (16) φ (nTs) = tan ⁻¹ (Q (nTs) / I (nTs) (17) The symbol discriminating point detecting unit 10 calculates Yi (nTs), Yq (n
Ts), the value Yi at the symbol identification point represented by the following equation
(NT) and Yq (nT) (T: symbol period).

Yi (nT) = Acos (2π (fs / 8) nT + φ (nT)) / 2 (22) Yq (nT) = Asin (2π (fs / 8) nT + φ (nT)) / 2 (23) A = R (nT) (24) φ (nT) = tan ^-1 (Q (nT) / I (nT)) (25) The baseband differential detection unit 11 calculates the following equations (26) and (27). Perform baseband differential detection.

ID (nT) = Yi (nT) Yi ((n−1) T) + Yq (nT) Yq ((n−1) T) (26) QD (nT) = Yi ((n−1) T ) Yq (nT) −Yi (nT) Yq ((n−1) T) (27) As a result, the cosine and sine of the modulation phase difference are generated as shown in the following equation.

ID (nT) = A ² cos (△ φ (nT) + θe) (28) QD (nT) = A ² sin (△ φ (nT) + θe) (29) Δφ (nT) = φ (nT ) −φ ((n−1) T) (30) θe = 2π △ fT (△ f = fs / 8) (31) The phase compensating unit 12 performs operations such as the following equations (32) and (33). By doing so, the phase error θe caused by the frequency offset Δf generated in the output of the baseband differential detection unit 11 is compensated. Note that cos θe, si
nθe is supplied from the compensation coefficient memory 20.

Ie (nT) = ID (nT) cosθe + QD (nT) sinθe (32) Qe (nT) = QD (nT) cosθe−ID (nT) sinθe (33) As a result, the baseband represented by the following equation is obtained. A signal is obtained.

Ie (nT) = A ² cos (△ φ (nT)) (34) Qe (nT) = A ² sin (△ φ (nT)) (35) Thus, digital quadrature detection of the second embodiment In the device,
Performs quadrature detection on the input IF signal at the local frequency fs / 4, regardless of the value of the center frequency f _IF ,
Equivalently, the sampling frequency is converted to fs / 2 to reduce the calculation amount of the reception FIR bandpass filter.
A high-precision baseband signal is obtained by phase-compensating the frequency offset Δf of the filter output by compensating for a constant phase error θe after baseband differential detection. In this case, the compensation coefficient memory 20 only needs to store only two data cos θe and sin θe that do not depend on the time n as the compensation function values. Therefore, a large memory capacity is not required.

(Third Embodiment) The digital quadrature detector of the third embodiment reduces the sampling frequency by thinning out the sample value sequence after performing the quadrature detection process.

This device, as shown in FIG. 4, samples an IF signal input terminal 1 to which an IF signal is input and an input IF signal at a sampling frequency fs, and obtains a center frequency fc = mfs ± f _IF (m = 0 , ± 1, ± 2, ‥) and converts the A / D-converted sample value sequence S (nTs) to a local frequency f
Cos for quadrature detection at ₀ (= minimum value of | fc |)
(2πf ₀ nTs), multipliers 3 and 4 for multiplying sin (2πf ₀ nTs), a simple FIR low-pass filter 13 for suppressing an interference wave component mixed into the received IF signal,
14, thinning-out processing units 15 and 16 for thinning out the output sample value sequences of the FIR low-pass filters 13 and 14 every other sample, reception FIR low-pass filters 17 and 18 for band-limiting the outputs of the thinning-out processing units 15 and 16, An in-phase output terminal 8 and a quadrature output terminal 9 for outputting a band signal are provided.

In this device, by providing the simple FIR low-pass filters 13 and 14, even if an interfering wave is mixed in the input IF signal, it is possible to generate a baseband signal excluding the interfering wave.

[0046] represented as the center frequency f _IF 1 of the center frequency f _IF 0 and disturbance IF signal of the desired wave IF signal input to the device generally following equation.

[0047] _{f IF 0 = (N0 / D0} ) fs (N0, D0: positive _{integer) (36) f IF 1 =} (N1 / D1) fs (N1, D1: positive integer) (37) when this, A / The desired wave component and the interference wave component in the output sample value sequence of the D converter 2 are distributed with the frequencies fc0 and fc1 of the following equations (38) and (39) as center frequencies.

[0048] _{fc0 = mfs ± f IF 0 =} ((mD0 ± N0) / D0) fs (m = 0, ± 1, ± 2, ‥) (38) fc1 = mfs ± f IF 1 = ((mD1 ± N1 ) / D1) fs (m = 0, ± 1, ± 2, ‥) (39) Hereinafter, for example, N0 = 9, D0 = 8, N1 = 11, D
The operation when 1 = 8 will be described. At this time, equation (3)
6) to (39) are represented by the following equations (40) to (42), and the frequency spectra at the input and output of the A / D converter 2 are as shown in FIGS.

[0049] _{f IF 0 = (9/8) fs} (40) f IF 1 = (11/8) fs (41) fc0 = (m ± (9/8)) fs (42) fc1 = (m ± ( 11/8)) fs (42) (m = 0, ± 1, ± 2, ‥) At this time, the output S (nTs) of the A / D converter 2 is represented by the following equation.

S (nTs) = SD (nTs) + SU (nTs) (43) SD (nTs) = I (nTs) cos (2π (fs / 8) nTs) + Q (nTs) sin (2π (fs / 8) nTs) (44) SU (nTs) = Wi (nTs) cos (2π (3fs / 8) nTs) + Wq (nTs) sin (2π (3fs / 8) nTs) (45) where SD (nTs) is a desired wave The component SU (nTs) indicates an interference wave component.

On the other hand, the quadrature detection section performs quadrature detection at a local frequency f ₀ represented by the following equation. f ₀ = | fc ₀ | min = fs / 8 (46) Therefore, the multiplier 3 performs cos on S (nTs).
(2πf ₀ nTs), and multiplier 4 calculates sin (2πf ₀ nTs).
f ₀ nTs). As a result, the outputs Xi (nTs) and Xq (nTs) of the quadrature detector are expressed as follows.

[0052] Xi (nTs) = S (nTs ) cos (2πf 0 nTs) = XDi (nTs) + XUi (nTs) (47) XDi (nTs) = SD (nTs) cos (2πf 0 nTs) = I (nTs) / 2 + (I (nTs) cos (2π (fs / 4) nTs) + Q (nTs) sin (2π (fs / 4) nTs)) / 2 (48) XUi (nTs) = SU (nTs) cos (2πf) ₀ nTs) = (Wi (nTs) cos (2π (fs / 4) nTs) + Wq (nTs) sin (2π (fs / 4) nTs)) / 2+ (Wi (nTs) cos (2π (fs / 2)) nTs) + Wq (nTs) sin (2π (fs / 2) nTs)) / 2 (49) Xq (nTs) = S (nTs) sin (2πf 0 nTs) = XDq (nTs) + XUq (nTs) (50) X q (nTs) = SD (nTs ) sin (2πf 0 nTs) = Q (nTs) / 2 + (I (nTs) sin (2π (fs / 4) nTs) -Q (nTs) cos (2π (fs / 4 ) nTs)) / 2 (51 ) XUq (nTs) = SU (nTs) sin (2πf 0 nTs) = (Wi (nTs) sin (2π (fs / 2) nTs) -Wq (nTs) cos (2π (fs / 2) nTs)) / 2-(Wi (nTs) sin (2π (fs / 4) nTs) -Wq (nTs) cos (2π (fs / 4) nTs)) / 2 (52) where XDi (nTs) ), XUi (nTs) are the desired wave component and the interference wave component in the in-phase output Xi (nTs), and XDq (nTs) and XUq (nTs) are the quadrature output X.
A desired wave component and an interference wave component at q (nTs) are shown. The frequency spectrum of the output of the quadrature detector is the multiplication of the multiplier 3, 4, fc0, fc1 and f ₀
And ± are obtained as shown in FIG. 5C.

Next, if there is a possibility that the component of the interference wave after the quadrature detection may overlap with the component of the desired wave after the thinning process, the interference wave component is suppressed by the FIR low-pass filters 13 and 14 in advance. . FIG. 5C or equation (49)
As can be seen from (52), mfs + (fs / 2) (m
= 0, ± 1, ± 2, ‥), the interference wave component is generated when a thinning process described below for converting the sampling frequency to fs / 2 is executed. It overlaps with the frequency of the desired wave component of the baseband after the thinning processing (for example, fs, 2fs, 3fs, ‥).
The desired wave component corresponding to the sampling frequency fs /
The conversion to 2 gives fs / 2, fs, 3fs / 2, ‥,
The fs / 2 and 3fs / 2 overlap with the interference wave component of mfs + (fs / 2)). After the desired wave and the interference wave are superimposed, the interference wave cannot be removed.

Therefore, this interference wave component is suppressed in advance by simple FIR low-pass filters (for example, about 7 taps) 13 and 14. Then, in the thinning processing units 15 and 16,
The output sample value sequence of the FIR low-pass filters 13 and 14 is 1
Decimate every sample, set sampling frequency to fs / 2
Convert to As a result, the frequency spectrum shown in FIG. 5D is obtained.

Next, as shown in FIG. 5D, unnecessary components other than the desired wave components distributed around, for example, the frequency mfs (m = 0, ± 1, ± 2, ‥) are received by the reception FIR low-pass filter 17. By suppressing at 18, a baseband signal can be obtained as shown in FIG. At this time, the number of taps of the reception FIR low-pass filters 17 and 18 is L
If it is a tap, since the sampling frequency of the signal input to the filters 17 and 18 has been converted to fs / 2, the number of product-sum operations per output sample is reduced to L / 2.

As described above, in the digital quadrature detection apparatus according to the third embodiment, the sampling frequency is converted to fs / 2 by performing the decimation processing after the quadrature detection processing, and the number of product-sum operations in the subsequent reception FIR low-pass filter is performed. To 1 /
Reduced to 2. Further, even when an interfering wave is mixed in the input IF signal, a part of the component that greatly affects the result is suppressed by a simple FIR low-pass filter,
Thereafter, by performing the thinning process and the process of the reception FIR low-pass filter, an excellent baseband signal can be obtained.

[0057]

As is apparent from the above description of the embodiment, the digital quadrature detector of the present invention performs the thinning process.
To change the sampling frequency to fs / 2.
Calculation of the reception FIR bandpass filter
The amount can be reduced. Also interferes with input IF signal
This component can be used in a simple FIR
High-pass filter
Command signal with a small number of operations.
You.

[0058]

[0059]

[Brief description of the drawings]

FIG. 1 is a block diagram showing a configuration of a digital quadrature detection apparatus according to a first embodiment;

FIG. 2 is a block diagram showing a configuration of a digital quadrature detection apparatus according to a second embodiment;

FIG. 3 is a frequency spectrum diagram of each part in the digital quadrature detector of the first and second embodiments;

FIG. 4 is a block diagram showing a configuration of a digital quadrature detection apparatus according to a third embodiment;

FIG. 5 is a frequency spectrum diagram of each part in a digital quadrature detection apparatus according to a third embodiment;

FIG. 6 is a block diagram showing a configuration of a conventional digital quadrature detection device;

FIG. 7 is a frequency spectrum diagram of each part in a conventional digital quadrature detection device.

[Explanation of symbols]

 Reference Signs List 1 IF signal input terminal 2 A / D converter 3, 4 Multiplier 5, 6, 17, 18 Received FIR bandpass filter 7 Phase compensator 8 In-phase output terminal 9 Quadrature output terminal 10 Symbol identification point detector 11 Baseband delay Detection unit 12 Phase compensation unit 13, 14 FIR low-pass filter 15, 16 Decimation processing unit 19 Memory for compensation function 20 Memory for compensation coefficient

──────────────────────────────────────────────────続 き Continuation of the front page (56) References JP-A-6-112984 (JP, A) JP-A-62-178046 (JP, A) JP-A-5-292133 (JP, A) JP-A-6-178133 181475 (JP, A) (58) Field surveyed (Int. Cl. ⁷ , DB name) H04L 27/00-27/38

Claims (1)

(57) [Claims] 1. An input IF signal having a sampling frequency fs
In the digital quadrature detector that samples the data by A / D conversion to a sample value signal of the center frequency fc and then performs quadrature detection to obtain a baseband signal, the A / D converted sample value is reduced to the minimum value of | fc | Equivalent
Quadrature detection at the local frequency
A quadrature detection means for generating a force, and a frequency mfs + included in the in-phase output and the quadrature output.
(Fs / 2) (m = 0, ± 1, ± 2, ‥)
First and second low-pass filters for suppressing the components of the first and second components, and output sample values of the first and second low-pass filters
The column is thinned out every other sample, and the sampling frequency is set to f
first and second thinning-out means for converting to s / 2, and band limiting the output of the first and second thinning-out means
Digital quadrature detection apparatus comprising first and second low-pass filters for receiving signals .