JP2803299B2 - Permanent magnet rotating machine - Google Patents

Description

The present invention relates to a permanent magnet motor that improves the performance of a conventional stepping motor or vernier motor and obtains a large and stable steady torque.

B. Summary of the Invention The present invention compensates for the disadvantages of stepping motors and vernier motors, in which the number of stator core slots Z ₁ , the number of rotor core slots Z ₂ ,
And the number of pole pairs P, Z ₂ = Z ₁ + P or Z ₂ = Z ₁ -P, and a three-phase winding is placed in the stator core slot. Basically, permanent magnets are arranged in at least one of all the slots and these magnets are magnetized in the same direction.If a small slot is provided in the stator teeth of the stator core, this small slot is used as a slot. Including in the number Z ₁
By detecting the rotor position and controlling the phase of the stator current based on this position, and controlling the amplitude of this stator current to generate the required torque, a high torque can be obtained even with a small current. Thus, steady torque can be obtained stably.

In addition, the rotor core is divided into two blocks along the axial direction, and these two blocks are displaced in the circumferential direction by half the rotor slot pitch. As a result, the permanent magnet causes a certain magnetic potential difference between the outer diameter of the stator core and the inner diameter of the rotor core for the permanent magnet, so that in a normal structure, the magnetic flux passes through the bearing. In order to prevent the possibility of harmful phenomena occurring in the bearing, the magnetic flux due to the above-mentioned constant magnetic potential difference is circulated between both blocks, or as another method, a non-magnetic material portion is provided on the bracket portion. The magnetic flux due to the above-mentioned constant magnetic potential difference is prevented from passing through the bearing portion.

C. Conventional technology and its problems There have been various motors, and one of them is a so-called vernier motor. The vernier motor, a stator core 1 and the rotor core 2, as shown in FIG. 9 has, first of a plurality of slots (slot number Z ₁ is the stator iron core 1 9
In the figure, 12) is provided, and the rotor core 2 is also provided with a plurality of slots (slot number Z ₂ and 10 in FIG. 9). It has a structure in which a phase winding (not shown) is provided.

Moreover, the relationship of this the number of slots the stator core 1 Z ₁ and the slot number Z ₂ of the rotor core _{2 Z 2 -Z 1 = ± P} ( where P is the number of poles of the stator three-phase winding) When selected, the rotational speed is 120 f / Z ₂ (rpm), and it has been found that ω _m = 2ω / Z _{2 when} represented by the rotational angular speed ω _m and the angular frequency ω.

Therefore, the angular frequency ω and the number of slots of the rotor core 2
Depending on the Z ₂ rotational speed is determined.

However, in this vernier motor, torque pulsation always occurs, and furthermore, the rotor winding does not depend, and the maximum torque that can be synchronized by overcoming the load inertia, that is, the pull-in torque, is small. There is a problem that the escape torque associated with the positional deviation is small.

On the other hand, there is a stepping motor as another thing. This stepping motor has a stator core 3 and a reluctance type rotor core as shown in FIG. 10, and each stator core 3 has a structure in which a stator coil 5 is wound independently. Therefore, in the example shown in FIG. 10, I, II, III
It is a structure in which phases are arranged.

By appropriately selecting each stator coil and passing a pulse signal, the magnetic field generated between the teeth between the stator core 3 and the rotor core 4 (only the I phase is shown in FIG. 10 (b)). The rotation is performed using the suction force.

However, in this stepping motor, there is a problem that torque pulsation occurs due to the structure, and furthermore, the rotational speed increases and the torque sharply decreases due to driving by the magnetic attraction force, and a stable steady torque can be obtained. Absent.

The present invention has been made in view of the above circumstances, and has as its object to provide a permanent magnet rotating machine that obtains a large and stable steady torque by suppressing torque pulsation.

The present invention to achieve the above-mentioned means purpose of solving the D. problem, Z and a rotor core having a number of slots Z ₂ in the stator core and an equal pitch having a slot number Z ₁ as the number of pole pairs of the P ₂ = Z ₁ + P or Z ₂ = Z ₁ -
P, and the slots of the stator core are 3
A phase winding is accommodated, and all slots of the stator core and all slots of the rotor core are basically provided with permanent magnets in at least one of all the slots, and are all magnetized in the same direction, and In the above-mentioned structure, a small slot is provided on the gap surface of the stator teeth of the stator core, and the small slot and the slot containing the winding are added to make the number of slots Z ₁ as a total slot. Further, the current phase is controlled as a function of the detected rotor position, and the current amplitude value is controlled so as to generate a required torque.

Furthermore, the rotor core is divided into two blocks along the axial direction, and these two blocks are displaced in the circumferential direction by half of the rotor slot pitch, and the magnet polarity is reversed in both blocks. As a result, a fixed magnetic field difference is generated between the outer periphery of the stator core and the inner periphery of the rotor core for the permanent magnet. In order to prevent the possibility of causing harmful phenomena to the bearing, the above-mentioned magnetic flux due to the constant magnetic potential difference may be circulated between the two blocks, or as another method, a non-magnetic material part may be attached to the bracket part. Is provided so that the magnetic flux due to the above-mentioned constant magnetic potential difference does not pass through the bearing portion.

E. Function By arranging permanent magnets in all slots of at least one of the rotor and the stator, the driving torque is strengthened, and a stable high torque can be obtained at a low current.

F. Embodiment An embodiment of the present invention will now be described with reference to FIG. 1 to FIG. FIG. 1 shows a rotor 10 and a stator.
11 is shown.

In FIG. 1, slots 11a of equal pitch are formed in the inner periphery of the stator 11, and its number Z ₁ (6 in FIG. 1)
It is. The slots 11a are provided with U, V, and W three-phase windings 12, so that a rotating magnetic field of 2P (P is the number of pole pairs, P = 1 in FIG. 1) is generated. Further, a permanent magnet 15 is embedded in the entrance of the slot 11a.

On the other hand, the rotor 10 has a
10c, which is the number Z ₂ (7 in FIG. 1).
The permanent magnet 16 is also provided in all the slots 10c of the rotor 10.
Is embedded.

These permanent magnets 15 and 16 are magnetized so that all magnets have the same polarity toward the center in the radial direction as shown in FIG.

And, to these slots number Z _1, Z _2, in the case of the Z ₂ = Z ₁ + P or Z ₂ = Z ₁ -P ... are formed to have a (1) the relationship (FIG. 1 Z ₂ =
Z ₁ + P).

The magnetic flux generated by each magnet forms a magnetic path closed by the magnet and the teeth through the gap.In the structure of the motor between the outer circumference of the stator core and the inner circumference of the stator core, this magnetic potential difference As a result, a magnetic flux having a magnetic path of the stator core-frame-bracket-bearing portion-shaft-fixed iron core is generated, which may cause a harmful phenomenon to the bearing. Two methods have been devised to prevent this.

The rotor 10 according to one embodiment is divided into two blocks 10a and 10b along the axial direction as shown in FIG. 2, and an AB surface which is a gap surface of the block 10a and a block 10b. The magnet planes of the permanent magnets 15 and 16 housed in the slots are opposite to each other on the AB plane and the CD plane so that the magnetic flux passes in different directions from the CD plane which is the gap plane. And For this reason, a closed magnetic path such as an arrow of block 10a-stator core 11-block 10b-shaft-block 10a is formed for the magnetic flux due to the above-mentioned constant magnetic potential difference, so that the magnetic flux passes through the bearing. Can be prevented.

Further, the block 10a and the block 10b are fixed to the shaft 14 so that the slot positions of the blocks 10a and 10b are circumferentially shifted from each other by a 1/2 slot pitch.

Now, considering the gap surface AB, when all the permanent magnets 15 of the stator slot 11a are magnetized to the S pole on the gap surface, the tooth portion between the stator slots 11a becomes the N pole, As a whole, a combination of NS poles composed of the permanent magnet 15 and adjacent teeth constitutes a total of 2Z ₁ pole magnetic poles.

On the other hand, the permanent magnet 16 in the slot 10c of the rotor 10 is
The eleven permanent magnets 15 generate a magnetic field in the same direction in the radial direction. In this case, the teeth adjacent to the slot 10c of the rotor 10 have S poles, and a total of 2Z ₂ magnetic poles are formed on the entire outer periphery of the rotor by the combination of the NS pole formed by the permanent magnet 16 and the adjacent teeth.

In such a structure, the peak values of the fundamental wave components of 2Z ₁ pole and 2Z ₂ pole of the magnetomotive force distribution generated by the permanent magnets 15 and 16 accommodated in the slots 11a and 10c are respectively _Fm1 and _Fm2.
And

Here, the center of the first phase, that is, the center of the U-phase winding group in one pole of the stator winding 12 (in the case of FIG. 1, since the number of slots per pole / phase is one, the center of the U-phase slot is ) As the origin, let the coordinates of the stator expressed by the space angle be θ ₁ and θ ₂
Are the coordinates taken on the rotor represented by spatial angles a central closest slot 10c to the origin of the stator coordinates theta ₁ as the origin at the moment of t = 0. Therefore, the relationship between θ ₁ and θ ₂ is given by the following equation (2).

θ ₂ = θ ₁ −ξΦ ₂ −ω _m t (2) where ω _m is the rotational angular velocity of the rotor 10, Φ ₂ is the spatial slot angle of the rotor, and ξΦ ₂ is the value when t = 0. θ
Since a spatial angle of _one of the origin and θ ₂ of origin, ξ is -0.
5 <ξ ≦ 0.5. That is, at t = 0, θ ₁ −θ ₂ = ξ
Φ ₂ , and when rotation starts, θ ₁ by ω _m t according to t
And the origin between θ ₂ is expanded.

Expressing the gap permeance by using the coordinates θ ₁ θ ₂ gives the following equation (3).

Here, α and γ are arbitrary integers.

The magnetomotive force of the fundamental wave component generated by the permanent magnet 15 of the stator 11 is given by the following equation (4).

F _m1 cos (Z ₁ θ ₁ ) (4) Therefore, the magnetic flux density generated by this magnetomotive force is (4)
The product of the formula and (3) child, using harmonic permeance, and substituting the relationship of formula (2) to eliminate θ ₂ , the magnetic flux density having a 2P pole as a spatial distribution is expressed by the following formula ( 5).

Similarly, the magnetomotive force generated by the permanent magnet 16 of the rotor 10 is given by the following equation (6).

F _m2 cos (Z ₂ θ ₂ ) (6) Accordingly, the magnetic flux density of the 2P pole generated by this magnetomotive force is expressed by the following equation (7), which is the same as the equation (5).

It was found that the magnetic flux densities (5) and (7) of these 2P poles had exactly the same form.
10. The magnetic flux density generated by all the permanent magnets of the stator 11 is given by the following equations (8) and (9).

On the other hand, when a three-phase alternating current flows through the stator winding 12, the fundamental wave magnetomotive force is expressed by the following equation (10).

Here, I ₁ is the effective current value, ω is the angular frequency, N ₁ is the number of one-phase series conductors, and _{kw 1} is the winding coefficient.

Equations (8) and (10) show that the number of poles in the spatial distribution is 2P
Since the poles are the same, a torque is generated between the two and Z ₂ = Z ₁ +
In the case of P, the following equation (11) is obtained.

T = KB _m I _1m sin [(Z ₂ ω _m −ω) t−ξZ ₂ Φ ₂ ] (11) where K is a constant calculated from design specifications.

From the equation (11), a steady torque is obtained by removing the pulsating torque in the state of Z ₂ ω _m −ω = 0, that is, ω _m = ω / Z ₂ (12). )

T _m = KB _m I _1m sin (ξZ ₂ φ ₂ ) (13) In the case of Z ₂ = Z ₁ -P, the rotation speed is ω _m = −ω / Z ₂ , in other words. , Ω / Z in the opposite direction to the rotating magnetic field
When rotating at an angular velocity of _2, a steady torque is obtained, and its value is the same as in equation (13).

The above explanation is an analysis on the AB surface of the rotor block.
Since the signs of F _m1 and F _m2 are reversed and the slot positions on the AB and CD surfaces are shifted by half the slot pitch,
The space angle ξφ ₂ between the origin of θ _{1 and} the origin of θ ₂ when t = 0
Is a CD surface becomes (ξ + 1/2) φ 2 _{becomes, Z 2 (ξ + 1/} 2) since it is _{Z 2 φ 2 = 2π φ 2} = Z 2 ξφ 2 + π.

Therefore, in the equation (13) of the torque, the sign of the term of the sin is reversed as in the following equation (14).

_{sinZ 2 (ξ + 1/2} ) φ 2 = -sinZ 2 ξφ 2 ... (14) whereas for the codes of all the magnetomotive force is reversed, B _m also becomes opposite sign from (10), the torque T of the CD surface Gives the torque of equation (13), which is exactly the same as the torque on the AB plane. Therefore AB
Double torque can be obtained between the surface and the CD surface. As a result, the electric motor shown in FIGS. 1 and 2 can be used as a periodic electric motor that rotates at the speed of the power supply angular frequency ω / Z ₂ .

In this case, problems such as starting, pull-in, and step-out of the synchronous motor may occur. However, as a control method without these problems, the rotor position is detected and ξZ ₂ φ ₂ = δ maintains the command value. It is conceivable to control the stator current in this way. In this case, it changes in the range of -π <δ ≦ π due to the relationship of -0.5 <ξ ≦ 0.5. Therefore, when δ is positive, δ = π in motor operation
/ 2 at maximum positive torque, when δ is negative, generator operation δ = −
The maximum negative torque is obtained at π / 2, and this negative torque can be used as a braking torque during deceleration.

described below to control when operating at the specified rotational speed omega _m while maintaining the command value of [delta] [delta] ^*.

Assuming that the coordinates of the slot center position closest to the origin of θ ₁ at the moment of t = 0 are θ ₂ , the following equation is satisfied when the above-mentioned condition of ω _m = ω / Z ₂ is satisfied and a steady torque is generated. Get.

_{θ 2 = ω m t + ξφ} 2 = ωt / Z 2 + δ / Z 2 ... (15) obtaining Therefore _{ωt = (Z 2 θ 2 -δ} ).

Therefore in order to rotate with omega _m becomes an angular velocity while maintaining a [delta] ^* detects the position theta ₂ of the center of the rotor slots calculating the following equation (16).

(Ωt) ^* = (Z ₂ θ ₂ −δ ^* ) (16) Then, the phases of the respective phase currents are controlled so as to obtain the expression (17).

On the other hand, for the current amplitude value I _1m , the command of ω _m ▲ ω ^*
_m ▼ and compared with the actual omega _m, command value I _{1 m} to the deviation to zero ▲ I ^* _1m ▼ give. In this way,
Command value ▲ i ^* _u ▼, ▲ i ^{* of} each phase current supplied to motor
_v ▼, ▲ i ^* _w ▼ it is possible to give, by controlling the current control type inverter to flow a current of the command value as can be driven electric motor always omega _m.

In order to realize the circuit in FIG. 3, a motor 21 is controlled by a current control type inverter 20, and a position detector 22 such as a rotary encoder provided in the motor 21 includes:
Signal indicating the rotor position theta ₂ is output. The rotor position theta ₂ is computed by adding the load angle command [delta] ^* by the current phase calculation circuit 23, based on the aforementioned equation (16) (.omega.t) ^* calculated, cos [of two ROM tables ( ωt) ^* ] and co
s [(ωt) ^{* −} 4 / 3π] is obtained. Note that this δ ^* is set to π / 2 during acceleration and −π / 2 during deceleration. On the other hand, the pulse signal from the position detector is converted to the rotational angular velocity omega _m in F / V converter 24. A deviation is obtained between the output ω _{m of the} F / V converter 24 and the angular velocity command ω _m ^*, and then the current command ＩI ^* _1m ▼ is determined via the PI controller 25. As a result, co
s [(.omega.t) ^*] and cos (ωt) * ^-4 / 3π] a at the multiplying type D / A converter 26 converts D / A, to ▲ I ^*
By riding _1m ▼, i _u ^* and i _w ^* are obtained. And i
by using the relationship of _{u + i v + i w =} 0, ▲ i * v ▼ arithmetic circuit 27
, The voltage source inverter 20 is controlled so as to supply a current according to the V-phase current command value to the electric motor 21.

Further, since the equation (13) of the torque T is independent of the frequency, a limiter 28 is provided on the output side of the PI controller, and a constant torque can be generated by flowing a constant current value during acceleration and deceleration.

The hysteresis comparator 29 is further in this embodiment, according to the positive or negative sign of ^{_{(▲ ω * m ▼ -ω m}} ), which is accelerated by the electric motor operating as a positive phi ^* values for positive, if the reverse negative The regenerative braking is performed by the generator operation with the value of φ ^* also being negative.

In the above description, the case where Z ₂ = Z ₁ + P has been described. In the case where Z ₂ = Z ₁ -P, the expression (16) is as follows.

− (Ωt) ^* = (Z ₂ θ _n −δ ^* ) (18) Therefore, by substituting this value into equation (17), the current command of each phase is obtained as in the case of Z ₂ = Z ₁ + P. A value can be obtained, but the phase rotation is reversed. That is, in the case of Z ₂ + P and the case of Z ₂ -P, the direction of the rotating magnetic field due to the fixed current is reversed.

The above is the first embodiment. In the second embodiment, the rotor core is not divided into two blocks as described above, and only one block is provided on the inner periphery of the bracket 18 as shown in FIG. By providing the raw portion 19, it is possible to prevent the magnetic flux from passing through the bearing portion 20. In this case, the stator core 11−
It passes through a magnetic path of bracket 18-bearing cover 21-shaft 14-rotor core 10-stator core 10-stator core 11.

The motor has been described above. However, by maintaining δ at a negative value and supplying mechanical power to the shaft from the outside, the motor can be operated as a generator.

Although the description so far has been made with a combination of a small number of slots as shown in FIG. 1, as can be seen from the equation ω _m = ω / Z _2, if the rotational angular velocity requires a low speed, the number of slots can be increased. It turns out to be good. In this case, since Z ₁ and Z ₂ have the relationship of equation (1), there is a limit to increasing Z ₂ in the structure shown in FIG. Therefore, as in the Figure 5, a small slot provided in the teeth in addition to the slot that contains the windings, the total number of slots of the sum of the two is taken as Z _1, relative to the total number of slots Z ₁ ( 1) since you can choose Z ₂ as equation holds, in such a structure can sufficiently large Z _2, can be easily obtained extremely low speed.

FIG. 5 shows a gap portion of this modified motor. A winding slot 11a into which a three-phase winding is inserted is formed on the inner peripheral side of the stator core 11, and teeth 11 between the winding slots 11a are formed.
A small slot 11c facing the rotor is also formed on the inner peripheral side of b. The entrance of the winding slot 11a and the small slot 11c have the same width. In this case, if the total number of slots and Z _1, slot pitch phi ₁ of the stator core 11 has a constant pitch of 2π / Z _1. The inlet of the winding slot 11a and the small slot 11c may have the same width in order to reduce the torque ripple during operation, but it is not particularly limited to this. A permanent magnet 15 is embedded at the entrance of the small slot and the winding slot.

On the other hand, the slot 10c of the slot number Z ₂ in the outer peripheral side of the rotor core 10 is formed, the slot pitch phi ₂ of the slot 10c has a constant pitch of 2 [pi / Z _2. Permanent magnets 16 are also embedded in the slots of these rotors.

Then, the relationship between the number of slots Z ₁ and Z ₂ is selected to be Z ₂ -Z ₁ = P or Z ₁ -Z ₂ =.

Next, the stationary torque will be described. The three-phase winding of this motor has a phase difference of 120 °, but does not change over time. DC current i _u = I _1m cosψ i _v = I _1m cos (ψ−2 / 3π) i _w = I _1m cos (ψ−4 / 3π)… (19) The torque generated when flowing is expressed by the following equation (20).

T = KI _1m B _m sin (ξZ ₂ φ ₂ -ψ) (20) where ψ is a phase angle from the peak value of the current of the first phase as shown in FIG. The value of the current flowing in each phase changes. The torque is a function of I _1m and ψ and ξ. When considering a certain constant I _1m , considering that Z ₂ φ ₂ = 2π as is apparent from the equation (20), ξ is −1. / 2
In the range from to +1/2, the waveform changes to a sine waveform as shown by the solid line in FIG. Since the torque has a peak value when ξ · 2π−ψ = π / 2, the peak value is obtained when ξ = 1/4 + ψ / 2π. Therefore, when ψ = 0, ξ = 1/4
When the value of ψ is positive, the peak value moves to the right by an amount corresponding to ψ / 2π.
When the value of is negative, it moves to the left by ψ / 2π. The range from -1/2 to +1/2 is the rotor one slot pitch, ξ = 0 corresponds to the center of the slot, and ス ロ ッ ト = 1/2 corresponds to the center of both adjacent teeth. -3π / 2 to + π
/ By changing to 2, when any position in one slot comes to the position of theta ₁ origin, it is possible to generate a positive peak torque. Since the value of the peak torque is proportional to I _{1 m,} by controlling the ψ and I _{1 m,} by generating a torque commensurate with the load torque at an arbitrary position, it is possible to rest in that position.

By adding a position control circuit having such a concept to FIG. 3, smooth and precise position control is possible.

In this structure, when a permanent magnet is embedded in the slot, in order to minimize the leakage flux of the permanent magnet passing through the teeth and increase the flux passing through the cap surface, it is necessary to use an eighth embodiment.
It is desirable to make the size of the permanent magnet 16 smaller in the depth direction D of the slot than the size W in the width direction of the slot as shown in FIG. If necessary, the leakage magnetic flux can be reduced by inserting a non-magnetic plate 17 between the permanent magnet 16 and the slot as shown in FIG. 8 (b).

In the above description, the magnetic flux passes through the bearing portion due to a certain magnetic field difference generated between the outer circumference of the stator core and the inner circumference of the rotor core, and two magnetic fluxes are provided to prevent the magnetic flux from passing through the bearing. Although the method has been described, even if a small-sized bracket with a normal structure is used, when the magnetic flux passing through the bearing portion is small and the rotation speed is low, the shaft voltage generated in the bearing portion is small and the bearing may be damaged. May not give. In such a case, it is not necessary to provide the bracket with a non-magnetic portion as shown in FIG. 4, and a bracket having a normal structure can be used.

Although the above description has been made with reference to a normal structure in which the rotor is inside the stator, an outer rotor type in which the rotor is arranged on the outer periphery of the stator may be used.

G. Effects of the Invention As described above, according to the present invention, unlike a stepping motor that obtains torque by the electromagnetic force acting between the teeth, the torque generated by the action of the armature current and the two rotating magnetic fields created by the permanent magnet is used. Since the driving system is used, stable steady torque can be obtained, and smooth speed control and precise position control can be performed. When utilizing the magnetic field generated by the slot permeance pulsation of the stator and rotor, the same torque can be generated with a smaller current compared to the reluctance type because the permanent magnet is used. It is effective in terms of conversion.

In particular, in the present invention, permanent magnets are provided in all slots of at least one of the stator core and the rotor core. However, when permanent magnets are provided in both slots, torque is increased by adding a corresponding amount. be able to.

Technological progress in permanent magnets has been remarkable, and magnets with good cost performance have been developed, so greater effects can be expected by using high-performance magnets.

[Brief description of the drawings]

1 to 6 relate to the present invention, FIG. 1 is a simplified configuration diagram of one embodiment, FIG. 2 is a sectional view, FIG. 3 is a block diagram of a control circuit, and FIG. 4 is another embodiment. 5, FIG. 5 is a block diagram of another embodiment, FIG. 6 is a waveform diagram showing a phase difference from a current peak value, FIG. 7 is a waveform diagram of a torque change with respect to 、, and FIG. FIG. 9 is a structural diagram of an example of a vernier motor, and FIGS. 10A and 10B are explanatory diagrams of a stepping motor. In the figure, 10 is a rotor, 10a and 10b are blocks, 10c, 11a and 11c are slots, 11 is a stator, 13, 15, and 16 are permanent magnets, 20 is a current-controlled inverter, 21 is a motor, and 23 is current. This is a phase calculation circuit.

Claims (4)

(57) [Claims] 1. A stator core having a number of slots Z _1, and a rotor core at equal pitch having a slot number Z _2, Z ₂ = Z ₁ as the number of pole pairs of the P + P or Z ₂ = Z _1, -P, a three-phase winding having the number of pole pairs P is accommodated in a slot of the stator core, and the rotor core is divided into two blocks along the axial direction to form a stator core. The two blocks are opposed to each other and fixed to the shaft at positions shifted from each other in the circumferential direction by 1/2 of the rotor slot pitch, and are fixed to at least one of the slots of the stator core and the rotor core. A permanent magnet rotating machine characterized by embedding permanent magnets, magnetizing all magnets in the same direction for each block, and magnetizing both blocks so that the magnets have opposite polarities. Becomes Z ₂ = Z ₁ + P or Z ₂ = Z ₁ -P pole as logarithm P and a rotor core having a wherein the number of slots Z ₂ in the stator core and an equal pitch having a slot number Z ₁ A three-phase winding having a pole pair number P is accommodated in a slot of the stator core, and a permanent magnet is embedded in at least one of all slots of the rotor core and all slots of the rotor core;
A permanent magnet rotating machine characterized in that all magnets are magnetized in the same direction and a part of magnetic flux generated by the permanent magnet does not pass through a bearing. 3. A includes a small slot gap surface of the stator teeth of the stator core, characterized in that the slot number Z ₁ as total added and the small slot and winding housed slot Slot A permanent magnet rotating machine according to claim 1 or 2. 4. The method according to claim 1, wherein the current phase is controlled as a function of the detected rotor position and the current amplitude value is controlled to generate a required torque.
The permanent magnet rotating machine according to (2) or (3).