JP2021158739A - Control device for ac rotary machine - Google Patents

Abstract

To provide a control device for an AC rotary machine capable of suppressing changes in a control response of a field current feedback control system depending on the presence or absence of magnetic saturation, even when the magnetic saturation occurs in a field winding type rotor.SOLUTION: A control device for an AC rotary machine sets proportional gain based on field current information so that the proportional gain of a proportional-integral control of a field current changes in proportion to a differential inductance that changes according to a field current by using a ratio of the change in magnetic flux of a rotor to the change in the field current at each operating point of the field current as differential inductance.SELECTED DRAWING: Figure 2

Description

The present application relates to a control device for an AC rotating machine.

Generally, as described in the equation (4-9) of Non-Patent Document 1, in the feedback control system of the armature current, the response angle frequency of the feedback control system is ωc, and the d-axis inductance is Ld. The q-axis inductance is Lq, the resistance is Ra, the proportional gain of the d-axis current proportional integration control is set to ωc × Ld, and the proportional gain of the q-axis current proportional integration control is set to ωc × Lq. The integral gain of the proportional integral control of the d-axis current and the q-axis current is set to ωc × Ra. The response angular frequency ωc is the reciprocal of the time constant τc of the feedback control system.

"Principles and Design Methods for Energy Saving Motors", by Shigeo Morimoto, Maruzen Publishing, January 28, 2018, p. 101

In the case of an AC rotating machine having a field winding on the rotor and an armature winding on the stator, magnetic saturation occurs in the rotor. In the magnetic saturation region, the increase in magnetic flux is smaller than the increase in current, but the magnetic flux of the rotor is determined by the product of the ampere turn, that is, the current flowing through the field winding and the number of turns of the field winding. If you want to achieve both small size and high output, you can obtain high output by using up to the magnetic saturation region.

In the field current control, since the magnetic saturation region is positively used to obtain even a small amount of magnetic flux, the inductance of the field winding changes to 1/10 times or less. If the proportional gain is set according to the inductance in the non-magnetic saturation region, the proportional gain is too large in the magnetic saturation region, and hunting may occur. On the contrary, if the proportional gain is set according to the inductance in the magnetic saturation region, the proportional gain is too small in the non-magnetic saturation region and the control response becomes slow.

Therefore, the present application provides a control device for an AC rotating machine capable of suppressing a change in the control response of the field current feedback control system depending on the presence or absence of magnetic saturation even when magnetic saturation occurs in the field winding type rotor. The purpose is to do.

The control device for an AC rotary machine according to the present application is a control device for an AC rotary machine that controls an AC rotary machine having a rotor provided with a field winding and a stator provided with an armature winding.
The field current command and the detected value of the field current are used so that the field current, which is the current flowing through the field winding, is detected and the detected value of the field current follows the field current command value. The field voltage command value is calculated by performing proportional integration control with respect to the deviation of, and a voltage is applied to the field winding based on the field voltage command value.
The ratio of the change in the magnetic flux of the rotor to the change in the field current at each operating point of the field current is defined as the differential inductance.
The proportional gain is set based on the information of the field current so that the proportional gain of the proportional integral control changes in proportion to the differential inductance that changes according to the field current.

According to the control device of the AC rotating machine according to the present application, even if the field current is increased up to the region where the rotor is magnetically saturated, the proportional gain changes in proportion to the differential inductance that changes according to the field current. It is possible to suppress the change in the response frequency of the field current feedback control system. Therefore, the feedback control system of the field current can be stabilized regardless of the presence or absence of magnetic saturation.

It is a schematic block diagram of the AC rotary machine and the control device of the AC rotary machine which concerns on Embodiment 1. FIG. It is a schematic block diagram of the control device which concerns on Embodiment 1. FIG. It is a hardware block diagram of the control device which concerns on Embodiment 1. FIG. It is a time chart explaining the on / off control behavior of the switching element of the inverter which concerns on Embodiment 1. FIG. It is a time chart explaining the on / off control behavior of the switching element of the inverter which concerns on Embodiment 1. FIG. The frequency of the field carrier according to the first embodiment is a diagram for explaining the relationship with the response frequency of the feedback control system of the field current. It is a block diagram of the feedback control system of the field current which concerns on Embodiment 1. FIG. It is a relationship characteristic diagram of the field current and the magnetic flux, the inductance, and the differential inductance which concerns on Embodiment 1. FIG. It is a figure explaining the table data in which the relationship between the field current and the differential inductance which concerns on Embodiment 1 is set. It is a schematic diagram of the AC rotary machine which was made into the generator motor for the vehicle which concerns on Embodiment 1. FIG. It is a schematic block diagram of the AC rotary machine and the control device of the AC rotary machine which concerns on Embodiment 2. FIG. FIG. 5 is a characteristic diagram of a relationship between a field current and a d-axis current and a differential inductance according to the second embodiment. It is a figure explaining the table data in which the relationship between the field current and the d-axis current and the differential inductance which concerns on Embodiment 2 is set. It is a figure explaining the table data in which the relationship between the conversion total current value and the differential inductance which concerns on Embodiment 2 is set.

1. 1. Embodiment 1
The control device 11 (hereinafter, simply referred to as the control device 11) of the AC rotary machine according to the first embodiment will be described with reference to the drawings. FIG. 1 is a schematic configuration diagram of an AC rotating machine 1 and a control device 11 according to the present embodiment.

1-1. AC rotating machine 1
The AC rotating machine 1 includes a stator 18 and a rotor 14 arranged inside the stator 18 in the radial direction. The AC rotating machine 1 is a field winding type synchronous rotating machine. An armature winding 12 is wound around the iron core of the stator 18. A field winding 4 is wound around the iron core of the rotor 14, and an electromagnet is provided.

In the present embodiment, the armature winding 12 is a U-phase, V-phase, and W-phase three-phase armature winding Cu, Cv, and Cw. The three-phase armature windings Cu, Cv, and Cw may be star-connected or delta-connected.

The rotor 14 is provided with a rotation sensor 15 that detects the rotation angle (rotation angle) of the rotor 14. The output signal of the rotation sensor 15 is input to the control device 11. As the rotation sensor 15, various sensors such as a Hall element, a resolver, and an encoder are used. The rotation sensor 15 may not be provided, and may be configured to estimate the rotation angle (magnetic pole position) based on the current information obtained by superimposing the harmonic component on the current command value described later (so-called). Sensorless method).

1-2. DC power supply 2
The DC power supply 2 outputs the power supply voltage Vdc to the inverter 5 and the converter 9. As the DC power supply 2, any device that outputs a DC voltage, such as a battery, a DC-DC converter, a diode rectifier, and a PWM rectifier, is used. A smoothing capacitor 3 is connected in parallel to the DC power supply 2.

1-3. Inverter 5
The inverter 5 has a plurality of switching elements and performs power conversion between the DC power supply 2 and the armature winding 12. The inverter 5 is a series circuit in which a switching element SP on the positive electrode side connected to the positive electrode side of the DC power supply 2 and a switching element SN on the negative electrode side connected to the negative electrode side of the DC power supply 2 are connected in series. Three sets are provided corresponding to the armature winding of each phase. The connection points of the two switching elements in each series circuit are connected to the armature windings of the corresponding phases.

Specifically, in the U-phase series circuit, the switching element SPu on the positive electrode side of the U phase and the switching element SNu on the negative electrode side of the U phase are connected in series, and the connection point of the two switching elements is an armature of the U phase. It is connected to the winding Cu. In the V-phase series circuit, the switching element SPv on the positive electrode side of the V phase and the switching element SNv on the negative electrode side of the V phase are connected in series, and the connection point of the two switching elements is connected to the armature winding Cv of the V phase. Has been done. In the W-phase series circuit, the switching element SPw on the positive electrode side of W and the switching element SNw on the negative electrode side of W phase are connected in series, and the connection points of the two switching elements are connected to the armature winding Cw of W phase. ing.

As the switching element of the inverter 5, an IGBT (Insulated Gate Bipolar Transistor) in which diodes are connected in antiparallel, a bipolar transistor in which diodes are connected in antiparallel, a MOSFET (Metal Oxide Semiconductor Field Effect Transistor), and the like are used. The gate terminal of each switching element is connected to the control device 11 via a gate drive circuit or the like. Therefore, each switching element is turned on or off by the switching signal output from the control device 11.

The armature current sensor 8 is a current detection circuit that detects the current flowing through the armature windings Cu, Cv, and Cw of each phase. In the present embodiment, the armature current sensor 8 is provided on the electric wire connecting the series circuit of the switching elements of each phase and the armature winding. The output signal of the armature current sensor 8 of each phase is input to the control device 11. The armature current sensor 8 is a current sensor for a Hall element, a shunt resistor, or the like. The armature current sensor 8 may be connected in series to the series circuit of the switching elements of each phase.

1-4. Converter 9
The converter 9 has a switching element and performs power conversion between the DC power supply 2 and the field winding 4. In the present embodiment, the converter 9 is a series in which the switching element SP on the positive electrode side connected to the positive electrode side of the DC power supply 2 and the switching element SN on the negative electrode side connected to the negative electrode side of the DC power supply 2 are connected in series. It is an H-bridge circuit with two sets of circuits. The connection point between the positive electrode side switching element SP1 and the negative electrode side switching element SN1 in the first set of series circuit 28 is connected to one end of the field winding 4, and the positive electrode side switching in the second set of series circuit 29 The connection point between the element SP2 and the switching element SN2 on the negative electrode side is connected to the other end of the field winding 4.

As the switching element of the converter 9, an IGBT in which diodes are connected in antiparallel, a bipolar transistor in which diodes are connected in antiparallel, a MOSFET, and the like are used. The gate terminal of each switching element is connected to the control device 11 via a gate drive circuit or the like. Therefore, each switching element is turned on or off by the switching signal output from the control device 11.

The converter 9 has other configurations such as replacing the switching element SN1 on the negative side of the series circuit 28 of the first set with a diode and replacing the switching element SN2 on the negative side of the series circuit 29 of the second set with a diode. May be.

The field current sensor 6 is a current detection circuit that detects the field current If, which is the current flowing through the field winding 4. In the present embodiment, the field current sensor 6 is provided on the electric wire on the negative electrode side of the switching element SN2 on the negative electrode side of the second set of series circuits 29. The field current sensor 6 may be provided at another location where the field current If can be detected. The output signal of the field current sensor 6 is input to the control device 11. The field current sensor 6 is a current sensor for a Hall element, a shunt resistor, or the like.

1-5. Control device 11
The control device 11 controls the AC rotating machine 1 via the inverter 5 and the converter 9. As shown in FIG. 2, the control device 11 includes functional units such as a rotation detection unit 31, an armature current detection unit 32, an armature current control unit 33, a field current detection unit 34, and a field current control unit 35. I have. Each function of the control device 11 is realized by a processing circuit provided in the control device 11. Specifically, as shown in FIG. 3, the control device 11 includes an arithmetic processing unit 90 (computer) such as a CPU (Central Processing Unit), a storage device 91 for exchanging data with the arithmetic processing unit 90, as a processing circuit. The arithmetic processing unit 90 includes an input circuit 92 for inputting an external signal, an output circuit 93 for outputting a signal from the arithmetic processing unit 90 to the outside, a communication circuit 94 for data communication with the external device, and the like.

The arithmetic processing device 90 is provided with an ASIC (Application Specific Integrated Circuit), an IC (Integrated Circuit), a DSP (Digital Signal Processor), an FPGA (Field Programmable Gate Array), various logic circuits, various signal processing circuits, and the like. You may. Further, as the arithmetic processing unit 90, a plurality of the same type or different types may be provided, and each processing may be shared and executed. As the storage device 91, a RAM (Random Access Memory) configured to be able to read and write data from the arithmetic processing unit 90, a ROM (Read Only Memory) configured to be able to read data from the arithmetic processing unit 90, and the like are used. It is equipped. The input circuit 92 is connected to various sensors and switches such as a rotation sensor 15, an armature current sensor 8, and a field current sensor 6, and A / D conversion that inputs the output signals of these sensors and switches to the arithmetic processing device 90. Equipped with vessels, etc. The output circuit 93 includes an electric load such as a gate drive circuit that drives the switching elements of the inverter 5 and the converter 9 on and off, and a drive circuit or the like that outputs a control signal from the arithmetic processing device 90 to these electric loads. The communication circuit 94 communicates with an external device.

Then, for each function of the control units 31 to 35 and the like included in the control device 11, the arithmetic processing unit 90 executes software (program) stored in the storage device 91 such as the ROM, and the storage device 91 and the input circuit 92. , And other hardware of the control device 11 such as the output circuit 93. The setting data such as table data used by each control unit 31 to 35 and the like is stored in a storage device 91 such as a ROM as a part of software (program). Hereinafter, each function of the control device 11 will be described in detail.

<Control of armature current>
The rotation detection unit 31 detects the magnetic pole position θ (rotation angle θ of the rotor) and the rotation angular velocity ω of the rotor at the electric angle. In the present embodiment, the rotation detection unit 31 detects the magnetic pole position θ (rotation angle θ) and the rotation angular velocity ω of the rotor based on the output signal of the rotation sensor 15. The magnetic pole position is set in the direction of the north pole of the electromagnet provided on the rotor. The rotation detection unit 31 is configured to estimate the rotation angle (magnetic pole position) based on the current information obtained by superimposing the harmonic component on the current command value without using the rotation sensor. It may be (so-called sensorless method).

The armature current detection unit 32 detects the armature currents Iur, Ivr, and Iwr flowing in the three-phase windings based on the output signal of the armature current sensor 8. Here, Iur is the detected value of the U-phase armature current Iu, Ivr is the detected value of the V-phase armature current Iv, and Iwr is the detected value of the W-phase armature current Iw. .. The armature current sensor 8 may be configured to detect the two-phase armature current, and the remaining one-phase armature current may be calculated based on the detected value of the two-phase armature current. For example, the armature current sensor 8 may detect the V-phase and W-phase armature currents Ivr and Iwr, and the U-phase armature current Iur may be calculated by Iur = -Ivr-Iwr.

The armature current control unit 33 calculates the armature voltage command value based on the armature current command and the armature current detection value so that the armature current detection value approaches the armature current command value, and the armature A voltage is applied to the armature winding based on the child voltage command value.

In the present embodiment, as shown in FIG. 2, the armature current control unit 33 includes a current coordinate conversion unit 331, a current command value calculation unit 332, a voltage command value calculation unit 333, a voltage coordinate conversion unit 334, and switching control. The unit 335 is provided.

The current coordinate conversion unit 331 converts the three-phase armature current detection values Iur, Ivr, and Iwr into the d-axis current detection values Idr and the q-axis current detection values Iqr on the d-axis and q-axis rotating coordinate systems. Convert. The rotation coordinate system of the d-axis and the q-axis is a two-axis rotation coordinate consisting of the d-axis defined in the direction of the detected magnetic pole position θ and the q-axis defined in the direction advanced by 90 ° in the electric angle from the d-axis. It rotates in synchronization with the rotation of the magnetic pole position θ of the rotor. Specifically, the armature current detection unit 32 performs three-phase two-phase conversion and rotational coordinate conversion on the three-phase armature current detection values Iur, Ivr, and Iwr based on the magnetic pole position θ, and performs d-axis. It is converted into the detected value Idr of the current and the detected value Iqr of the q-axis current.

The current command value calculation unit 332 calculates the armature current command value. In the present embodiment, the current command value calculation unit 332 calculates the current command value Ido on the d-axis and the current command value Iqo on the q-axis. In the present embodiment, the current command value calculation unit 332 calculates the current command values Ido and Iqo of the d-axis and the q-axis by controlling Id = 0. For example, the current command value calculation unit 332 sets the current command value Ido on the d-axis to 0, and sets the current command value Iqo on the q-axis according to the torque command value To.

The voltage command value calculation unit 333 performs proportional integration control on the deviation between the armature current command value and the armature current detection value so that the armature current detection value approaches the armature current command value. Calculate the armature voltage command value. In the present embodiment, the voltage command value calculation unit 333 sets the d-axis current command value Ido and the d-axis current detection value Idr so that the d-axis current detection value Idr approaches the d-axis current command value Ido. The voltage command value Vdo of the d-axis is calculated by performing proportional integration control with respect to the deviation ΔId of, and the current command value of the q-axis is approached so that the detected value Iqr of the q-axis current approaches the current command value Iqo of the q-axis. Proportional integration control is performed on the deviation ΔIq between Iqo and the detected value Iqr of the q-axis current to calculate the voltage command value Vqo of the q-axis. Further, a known feedforward control for non-interference between the d-axis current and the q-axis current may be performed.

Alternatively, the voltage command value calculation unit 333 changes the voltage command values of the d-axis and the q-axis by using the specifications of the AC rotating machine based on the current command values of the d-axis and the q-axis without using the current detection value. You may execute the feed forward control to make it.

The voltage coordinate conversion unit 334 performs fixed coordinate conversion and two-phase three-phase conversion on the d-axis and q-axis voltage command values Vdo and Vqo based on the magnetic pole position θ, and performs three-phase voltage command values Vuo and Vvo. , Convert to Vwo. The voltage coordinate conversion unit 334 may apply modulation such as two-phase modulation and space vector modulation so that the line voltage does not change with respect to the three-phase voltage command value.

The switching control unit 335 controls a plurality of switching elements of the inverter 5 on and off by PWM control (Pulse Width Modulation) based on the voltage command value. In the present embodiment, as shown in FIG. 4, the switching control unit 335 compares each of the three-phase voltage command values Vuo, Vvo, and Vwo with the armature carrier Cs vibrating at the period Tss of the armature carrier. This controls on / off of a plurality of switching elements. The armature carrier wave Cs is a triangular wave that oscillates with an amplitude of Vdc / 2, which is half the power supply voltage, centered on 0 in the period Tss of the armature carrier wave. The power supply voltage Vdc may be detected by the voltage sensor.

When the armature carrier Cs falls below the voltage command value for each phase, the switching control unit 335 turns on the switching signal QP of the switching element on the positive side (1 in this example) and turns on the switching element on the positive side. When the armature carrier Cs exceeds the voltage command value, the switching signal QP of the switching element on the positive side is turned off (0 in this example), and the switching element on the positive side is turned off. On the other hand, when the armature carrier Cs falls below the voltage command value for each phase, the switching control unit 335 turns off the switching signal QN of the switching element on the negative side (0 in this example) on the negative side. When the switching element is turned off, the switching element on the negative side is turned off, and the armature carrier Cs exceeds the voltage command value, the switching signal QN of the switching element on the negative side is turned on (1 in this example). , Turn on the switching element on the negative side. For each phase, between the on-period of the switching element on the positive electrode side and the on-period of the switching element on the negative electrode side, a short-circuit prevention period (dead time) in which both the switching element on the positive electrode side and the switching element on the negative electrode side are turned off. May be provided.

<Control of field current>
The field current detection unit 34 detects the field current Ifr, which is the current flowing through the field winding 4, based on the output signal of the field current sensor 6. Here, Ifr is a detected value of the field current If.

The field current control unit 35 proportionally integrates the field current command value Ifr and the field current detection value Ifr with respect to the deviation ΔIf so that the field current detection value Ifr approaches the field current command value Ifo. Control is performed to calculate the field voltage command value Vf, and a voltage is applied to the field winding 4 based on the field voltage command value Vf.

In the present embodiment, as shown in FIG. 2, the field current control unit 35 includes a current command value calculation unit 351, a voltage command value calculation unit 352, and a switching control unit 353.

The current command value calculation unit 351 sets the field current command value Ifo. For example, the current command value calculation unit 351 sets the field current command value Ifo based on the torque command value To or the like.

Then, as shown in the following equation, the voltage command value calculation unit 352 performs proportional integration control with respect to the deviation ΔIf between the field current command value Ifo and the field current detection value Ifr, and performs field voltage command value. Calculate Vf.

Here, Kpf is a proportional gain, Kif is an integral gain, s is a Laplace operator, and 1 / s represents an integral. The method of setting the proportional gain Kpf and the integrated gain Kif will be described later. Although the equation (1) is represented by a continuous system, in reality, the arithmetic processing is executed for each arithmetic cycle using the arithmetic expression in which the equation (1) is discretized by the arithmetic cycle. In the present embodiment, the calculation period is set to be the same as the period Tsf of the field carrier, which will be described later.

The switching control unit 353 controls a plurality of switching elements of the converter 9 on and off by PWM control based on the field voltage command value Vf.

For example, as shown in FIG. 5, the switching control unit 353 turns on and off a plurality of switching elements by comparing the field voltage command value Vf with the field carrier signal Cf vibrating in the period Tsf of the field carrier. Control. The field carrier signal Cf is a triangular wave that oscillates between 0 and the power supply voltage Vdc at the period Tsf of the field carrier. The power supply voltage Vdc may be detected by the voltage sensor.

When the armature carrier Cs falls below the field voltage command value Vf, the switching control unit 353 turns on the switching signal QP1 of the switching element SP1 on the positive electrode side of the first set (1 in this example) to turn on the positive electrode. The switching element SP1 on the side is turned on, and the switching signal QN1 of the switching element SN1 on the negative electrode side of the first set is turned off (0 in this example) to turn off the switching element SP1 on the negative electrode side. On the other hand, when the armature carrier Cs exceeds the field voltage command value Vf, the switching control unit 353 turns off (0) the switching signal QP1 of the switching element SP1 on the positive electrode side of the first set to turn off (0) the switching signal QP1 on the positive electrode side. The switching element SP1 is turned off, and the switching signal QN1 of the switching element SN1 on the negative electrode side of the first set is turned on (1) to turn on the switching element SN1 on the negative electrode side.

The switching control unit 353 turns off (0) the switching signal QP2 of the switching element SP2 on the positive electrode side of the second set, turns off the switching element SP2 on the positive electrode side, and turns off the switching element SP2 on the negative electrode side of the second set. The switching signal QN2 is turned on (1), and the switching element SN2 on the negative electrode side is turned on.

<Response frequency of each current control system>
Since the calorific value in the inverter 5 and the converter 9 is determined by the sum of the conduction loss and the switching loss, in order to reduce the calorific value, the conduction resistance is lowered, or the number of switchings is reduced to reduce the switching loss. The method is taken. When reducing the amount of heat generated by an inexpensive switching element, measures are taken to reduce the number of switchings, that is, to lower the frequency of the carrier wave.

The inductance of the field winding is larger than the inductance of the armature winding, and the response frequency fcf of the field current feedback control system is set lower than the response frequency fcs of the armature current feedback control system. Then, the following equation holds.

Here, the response frequency fcf of the field current feedback control system is the response frequency of the closed loop transfer function from the field current command value Ifo to the field current detection value Ifr, and the cutoff frequency of the closed loop transfer function and the response frequency of the closed loop transfer function. Corresponds to the inverse of the time constant. Further, the response frequency fcs of the feedback control system of the armature current is the response frequency of the closed loop transfer function from the current command value of the d-axis and the q-axis to the detected value of the d-axis and the q-axis current, and is the response frequency of the closed loop transfer function. Corresponds to the inverse of the cutoff frequency and time constant.

In order for the field current feedback control system to operate stably, the frequency fsf of the field carrier wave is higher than the response frequency fcf of the field current feedback control system, as shown in the following equation and the hatched region of FIG. Must also be set large. The same applies to the feedback control system for armature current. The frequency fsf of the field carrier wave is the reciprocal of the period Tsf of the field carrier wave.

In the present embodiment, in each of the field current control and the armature current control, the calculation cycle of the current feedback control is set to be the same as the cycle of the carrier wave. However, even if the calculation period is different from the period of the carrier wave, it can be considered in the same manner.

Further, in the armature current feedback control system having a relatively high response frequency, the period Tss of the armature carrier wave is shortened, so that the short-circuit prevention period (dead time) for turning off both the positive electrode side and the negative electrode side switching elements is set. It should not be long. However, in the feedback control system of the winding current having a relatively low response frequency, the period Tsf of the armature carrier wave becomes long, so that the settable range of the short circuit prevention period (dead time) becomes wide.

When an inexpensive switching element is used, it is necessary to lengthen the dead time, and the switching loss per switching becomes large. Therefore, in order to use an inexpensive switching element and reduce the cost, it is necessary to set the frequency fsf of the field carrier low.

<Feedback control system for field current in comparative example>
FIG. 7 is a block diagram of a field current feedback control system. As a comparative example, a case of designing in the same manner as the feedback control system of the armature current will be described. The field winding uses the inductance Lf of the field winding and is expressed by the voltage equation shown by the following equation. Here, Rf is the resistance of the field winding, and Vf is the applied voltage of the field winding.

The transfer function Gpf (s) of the field winding, which is obtained by Laplace transforming the equation (4), takes the applied voltage Vf of the field winding as an input, and outputs the field current If, is as follows.

Further, the transfer function Gcf (s) of the proportional integral control is as follows. Here, Kpf is a proportional gain and Kif is an integral gain.

Using the equations (5) and (6) as comparative examples, the closed-loop transfer function Gfb (s) becomes as follows, and the closed-loop transfer function Gfb (s) has a response frequency fcf (cutoff frequency, The proportional gain Kpf and the integrated gain Kif are set so as to be the reciprocal of the time constant × 2π).

In this way, when designed based on the voltage equation using the inductance Lf of the field winding, as in the armature current feedback control system, the proportional gain Kpf is proportional to the inductance Lf of the field winding. Set.

<Problem of setting proportional gain based on inductance Lf of field winding>
FIG. 8A shows the relationship characteristics between the field current If and the magnetic flux ψf of the rotor, and FIG. 8B shows the relationship characteristics between the field current If and the inductance Lf of the field winding. The inductance Lf of the field winding of FIG. 8 (b) is calculated by dividing the magnetic flux ψf of the rotor by the field current If, as shown in the following equation.

As shown in FIG. 8A, in the region where the field current If is smaller than the operating point A, the rotor is not magnetically saturated, and the magnetic flux ψf of the rotor increases in proportion to the field current If. The inductance Lf of the field winding is a constant value. On the other hand, when the field current If becomes larger than the operating point A, magnetic saturation occurs in the iron core of the rotor, and the slope of the increase in the magnetic flux ψf of the rotor with respect to the increase in the field current If decreases. Therefore, as the field current If increases, the inductance Lf of the field winding decreases.

Since the output torque of the AC rotating machine is given as shown in the following equation, the magnetic flux ψf or the q-axis current Iq of the rotor may be increased in order to increase the output torque.

Generally, in a field winding type rotor, since it is desired to increase the output torque while reducing the size of the iron core of the rotor, the field current If is increased up to the magnetic saturation region, and the magnetic flux ψf of the rotor is increased. In order to keep the response frequency fcf of the field current feedback control system constant even in the magnetic saturation region, the proportional gain Kpf from Eq. (7) can be changed according to the inductance Lf of the field winding. Conceivable. However, the voltage equation of the equation (4) is based on the premise that the inductance Lf of the field winding is time-invariant in the region where the magnetic saturation does not occur, which is insufficient.

<Control system design considering the magnetic saturation of the rotor>
Therefore, in the field winding type rotor, it is necessary to derive the equation on the premise that the field current If increases to the magnetic saturation region. Therefore, the voltage equation of the field winding is expressed by the following equation using the magnetic flux ψf of the rotor.

By transforming equation (10), the following equation is obtained. Here, the derivative of the magnetic flux ψf of the rotor with respect to the field current If is defined as the differential inductance dLf. The differential inductance dLf can also be expressed as the ratio of the change in the magnetic flux ψf of the rotor to the change in the field current If at each operating point of the field current If.

Using this equation (11), the transfer function Gpf (s) of the field winding is as follows.

Using these equations (12) and (6), the proportional gain is obtained so that the closed loop transfer function Gfb (s) has a response frequency fcf (cutoff frequency, reciprocal of time constant × 2π) as shown in the following equation. Kpf and the integrated gain Kif may be set.

Therefore, considering the magnetic saturation of the rotor, it can be seen that the proportional gain Kpf should be set to be proportional to the differential inductance dLf. As shown in FIG. 8 (c), the differential inductance dLf changes steeper than the inductance Lf in FIG. 8 (b).

For example, the cases of the operating point A, the operating point B, and the operating point C in FIG. 8 will be compared and described. At the operating point B, the magnetic flux φf is 1.9 times that of the operating point A, while the inductance Lf is 0.4 times and the differential inductance dLf is 0.1 times due to magnetic saturation. Under the condition that the d-axis current Id is 0 and the q-axis current Iq is the same, an output torque 1.9 times that of the operating point A can be obtained at the operating point B. When the proportional gain Kpf is set based on the inductance Lf of the operating point B, the value is four times the value based on the differential inductance dLf that is originally desired to be set, and the response frequency fcf is four times the target frequency. As a result, there is a possibility that the field current feedback control system becomes unstable because it extends beyond the hatched region of FIG.

Further, at the operating point C, the magnetic flux φ is doubled with respect to the operating point A, while the inductance Lf is 0.2 times and the differential inductance dLf is 0.02 times due to magnetic saturation. Under the condition that the d-axis current Id is 0 and the q-axis current Iq is the same, an output torque twice that of the operating point A can be obtained at the operating point C. When the proportional gain Kpf is set based on the inductance Lf of the operating point C, the value is 10 times the value based on the differential inductance dLf that is originally desired to be set, and the response frequency fcf is 10 times the target frequency. As a result, the field current feedback control system may become more unstable than the operating point B, protruding from the hatched region of FIG.

Further, assuming that the operating region of the field current If is up to the operating point C, if a fixed proportional gain Kpf is set based on the differential inductance dLf at the operating point A, the response frequency fcf at the operating point C is 50, which is the target frequency. Since it is doubled, the feedback control system of the field current becomes unstable. On the other hand, if a fixed proportional gain Kpf is set based on the differential inductance dLf at the operating point C, the response frequency fcf at the operating point A becomes 0.02 times the target frequency. It will be a slow response.

That is, in the field current control that positively uses up to the magnetic saturation region, in order to realize the target response frequency fcf over the entire operating region of the field current If, the proportional gain Kpf is set to the field. It is necessary to set it so that it changes in proportion to the differential inductance dLf that changes according to the current If.

<Proportional gain setting>
Based on the above derivation results, the field current control system is designed. That is, the voltage command value calculation unit 352 is proportional based on the field current information so that the proportional gain Kpf of the proportional integration control changes in proportion to the differential inductance dLf that changes according to the field current If. Set the gain Kpf. Here, the differential inductance dLf is the ratio of the change in the magnetic flux ψf of the rotor to the change in the field current If at each operating point of the field current If, as described above using the equation (11).

According to this configuration, even if the field current If is increased up to the region where the rotor is magnetically saturated, the proportional gain Kpf is set to change in proportion to the differential inductance dLf that changes according to the field current If. It is possible to suppress the change in the response frequency fcf of the field current feedback control system. Therefore, the field current feedback control system can be stabilized over the entire operating region of the field current If.

In the present embodiment, the voltage command value calculation unit 352 calculates the differential inductance dLf based on the field current information. The voltage command value calculation unit 352 uses the field current detection value Ifr as the field current information. Then, the voltage command value calculation unit 352 detects the current field current by using the setting data in which the relationship between the field current If and the differential inductance dLf is preset as shown in FIG. 8C. The differential inductance dLf corresponding to the value Ifr is calculated. As the setting data, table data as shown in FIG. 9 may be used, or a high-order approximate expression such as a polynomial may be used.

Alternatively, the voltage command value calculation unit 352 uses the closed loop transmission function Gfb (s) from the field current command value Ifo to the field current detection value Ifr with respect to the field current command value Ifo as field current information. The estimated value of the field current calculated by performing the response delay processing corresponding to the above may be used. For example, in this response delay processing, a first-order delay of the response frequency fcf as shown in the first equation of the equation (12) is used. The time constant of the first-order lag is the reciprocal of fcf × 2π. With this configuration, it is less susceptible to the field current detection error.

Then, the voltage command value calculation unit 352 calculates the proportional gain Kpf by multiplying the differential inductance dLf by the preset conversion coefficient Kc, as shown in the following equation.

The conversion coefficient Kc may be set to 2π × fcf as in the equation (13), but as described below, it is set in consideration of the influence of the waste time of the calculation cycle Tsf that occurs in the actual calculation processing. Will be done. As described above, the calculation cycle Tsf of the field current control is set as long as possible, so that the influence cannot be ignored. The transfer function Gd (s) of the dead time element of the calculation cycle Tsf is approximated by a first-order Padé approximation as shown in the following equation. Here, the dead time will be described as the calculation cycle Tsf, but it may be the time from the detection of the field current to the reflection of the field voltage command value in the output.

Using this equation (15), the closed-loop transfer function Gfb (s) becomes as follows.

In order for the response frequency (cutoff frequency) of the closed-loop transfer function Gfb (s) of the equation (16) to be the target response frequency fcf, the gain at the target response frequency fcf must satisfy the relationship of the following equation. The proportional gain Kpf and the integrated gain Kif may be set. The equation (6) is used for the transfer function Gcf (s) of the proportional integral control, and the equation (12) is used for the transfer function Gpf (s) of the field winding.

The proportional gain Kpf and the integrated gain Kif satisfying the equation (17) are as follows. Here, Kl is an adjustment gain set to a positive value of 1 or less so that the equation (17) is satisfied, and changes according to the calculation cycle Tsf.

Therefore, the voltage command value calculation unit 352 calculates the proportional gain Kpf using the first equation of the equation (18). Further, the voltage command value calculation unit 352 sets the integral gain Kif by using the second equation of the equation (18). The voltage command value calculation unit 352 may change the target response frequency fcf according to the operating state. The adjustment gain Kl may be calculated from the equations (20) and (17) with the dead time element as the equation (19).

Alternatively, the voltage command value calculation unit 352 calculates the proportional gain Kpf corresponding to the current detected value Ifr of the field current using the setting data in which the relationship between the field current If and the proportional gain Kpf is preset. You may. This setting data uses the characteristics of FIG. 8 (c) and the first equation of equation (18) so that the proportional gain Kpf changes in proportion to the differential inductance dLf that changes according to the field current If. Is preset.

<Setting of proportional gain and integrated gain>
In the above equation (14), the conversion coefficient Kc was multiplied in the calculation of the proportional gain Kpf, but as shown in the equation (20), the conversion coefficient Kc was added to the resistance Rf of the field winding as in the proportional gain Kpf. The integrated gain Kif may be calculated by multiplication.

From equations (19) and (17), the proportional gain Kpf and the integral gain Kif are as shown in equation (21).

When the dead time is TL, the adjustment gain Kl is given by the equation (22) and is determined by the dead time TL and the response frequency fcf.

That is, the adjustment gain Kl is a positive coefficient of 1 or less that reduces the response frequency that is apparently increased by the amount of the dead time TL, as in the equation (23).

When the proportional gain Kpf and the integral gain Kif are given by the equation (21), the differential inductance dLf and the resistance Rf do not participate in the adjustment gain Kl because the pole-zero cancellation is possible in the closed loop transfer function Gfb (s). That is, even when the change width of both or one of the differential inductance dLf and the resistance Rf is large, it is possible to give the adjustment gain Kl at a constant value. Further, since the adjustment gain Kl does not affect the pole-zero cancellation, overshoot can be suppressed even when the difference between the adjustment gain Kl and 1 is large.

<When used as a generator motor for vehicles>
When the control device for the AC rotary machine of the present embodiment is used for a generator motor for a vehicle, the configuration is as shown in FIG. The rotating shaft of the rotor of the AC rotating machine 1 is connected to the crankshaft of the internal combustion engine 100 via a pulley and a belt mechanism 101. The rotating shaft of the AC rotating machine 1 is connected to the wheels 103 via the internal combustion engine 100 and the transmission 102. The AC rotating machine 1 functions as an electric motor, serves as an auxiliary machine for the internal combustion engine 100, serves as a driving force source for the wheels 103, and also functions as a generator, and generates electricity by utilizing the rotation of the internal combustion engine 100. The current command value calculation unit 332 of the armature winding and the current command value calculation unit 351 of the field winding have the armature current command values Ido, Iqo and the field current based on the control command values such as the torque command value To. The command value Ifo is calculated.

In recent years, the number of idle-stop vehicles has increased, and there is a demand for shortening the rise time of the drive torque when the internal combustion engine is restarted by the drive torque of the AC rotating machine 1. Since the rising time of the field current greatly affects the rising time of the drive torque, we want to increase the response frequency of the field current control. As described above, by appropriately setting the proportional gain Kpf in consideration of the differential inductance dLf, a desired response frequency can be realized for a wide range of field currents, and the rise time of the drive torque can be shortened. ..

Further, in order to improve fuel efficiency, even when the torque of the internal combustion engine is assisted by the drive torque of the AC rotary machine 1, the rise of the drive torque during acceleration can be improved by appropriately setting the proportional gain Kpf. It is possible. Furthermore, as the frequency of use on the drive side increases, efficient power generation is required to secure the power balance, but by appropriately setting the proportional gain Kpf, it is possible to cover a wide range of field currents. Therefore, a desired response frequency can be realized, the rise time of the magnetic flux can be shortened, and the response of the amount of power generation can be improved.

2. Embodiment 2
Next, the AC rotating machine 1 and the control device 11 according to the second embodiment will be described. Description of the same components as in the first embodiment will be omitted. The basic configuration of the AC rotating machine 1 and the control device 11 according to the present embodiment is the same as that of the first embodiment, but the method of setting the proportional gain Kpf is different from that of the first embodiment. FIG. 11 shows a block diagram of the control device 11 according to the present embodiment.

In the first embodiment, Id = 0 control is performed, and the proportional gain Kpf is set based on the information of the field current. However, in the present embodiment, the d-axis current Id, which is a component of the armature current in the magnetic pole direction of the rotor, is changed from 0. Therefore, the voltage command value calculation unit 352 sets the proportional gain Kpf based on the information of the d-axis current in addition to the information of the field current.

In the present embodiment, the current command value calculation unit 332 of the armature current control unit 33 changes the current command value Ido of the d-axis from 0. The current command value calculation unit 332 reduces the current command value Ido of the d-axis from 0 by the maximum torque current control, the weakening magnetic flux control, or the like. Alternatively, the current command value calculation unit 332 may increase the current command value Ido on the d-axis from 0 in order to strengthen the magnetic flux of the rotor. As described in the first embodiment, the detected value Idr of the d-axis current is controlled so as to follow the current command value Ido of the d-axis.

When the d-axis current Id is increased from 0, the magnetic flux ψf of the rotor increases, and magnetic saturation occurs in the rotor at a field current If smaller than that in the state of Id = 0. On the other hand, when the d-axis current Id is reduced from 0, the magnetic flux ψf of the rotor is reduced, and magnetic saturation occurs in the rotor at a field current If larger than that in the state of Id = 0. Therefore, as the d-axis current Id increases or decreases from 0, the degree of magnetic saturation increases or decreases even in the same field current If, and as shown in FIG. 12, the differential inductance dLf increases or decreases.

In FIG. 12, the solid line shows the d-axis current Id of 0, the broken line shows the d-axis current Id of more than 0, and the alternate long and short dash line shows the field current If and the differential inductance dLf when the d-axis current Id is smaller than 0. Represents a relationship. In this way, when the d-axis current Id is changed from 0, the differential inductance dLf changes according to the field current If and the d-axis current Id.

Therefore, in the present embodiment, the voltage command value calculation unit 352 sets the proportional gain Kpf based on the information of the field current and the information of the d-axis current. The voltage command value calculation unit 352 determines the field current information and the d-axis current information so that the proportional gain Kpf changes in proportion to the differential inductance dLf that changes according to the field current If and the d-axis current Id. Based on, the proportional gain Kpf is set.

According to this configuration, when the d-axis current Id is changed from 0, the magnetic flux ψf of the rotor changes, the degree of magnetic saturation of the rotor changes, and the differential inductance dLf changes in the same field current If. Even in this case, the proportional gain Kpf can be set to change in proportion to the differential inductance dLf that changes according to the field current If and the d-axis current Id, and the response frequency fcf of the field current feedback control system changes. Can be suppressed. Therefore, even if the d-axis current Id is changed from 0, the field current feedback control system can be stabilized over the entire operating region of the field current If.

In the present embodiment, the voltage command value calculation unit 352 calculates the differential inductance dLf based on the field current information and the d-axis current information. Similar to the first embodiment, the voltage command value calculation unit 352 uses the detected value Ifr of the field current or the estimated value of the field current as the information of the field current.

The voltage command value calculation unit 352 uses the detected value Idr of the d-axis current as the information of the d-axis current. Alternatively, the voltage command value calculation unit 352 may use the d-axis current command value Ido as the d-axis current information. As described using the equation (2), the response frequency fcf of the feedback control system (closed loop transfer function) from the field current command value Ifo to the field current detection value Ifr is determined from the armature current command value to the armature. It is lower than the response frequency fcs of the feedback control system (closed loop transfer function) up to the detected value of the current. Therefore, the response difference between the actual d-axis current Id and the d-axis current command value Ido is smaller than the response of the field current. Therefore, the d-axis current command value Ido is used as the d-axis current information. Is also practically acceptable.

The voltage command value calculation unit 352 estimates the current field current detection value Ifr or field current using the setting data in which the relationship between the field current If and the d-axis current Id and the differential inductance dLf is preset. The value and the differential inductance dLf corresponding to the detected value Idr of the d-axis current or the current command value Ido of the d-axis are calculated. As the setting data, table data having the field current If and the d-axis current Id as the search axis as shown in FIG. 13 may be used, or a higher-order approximate expression may be used. Further, since the differential inductance dLf decreases as the d-axis current Id increases, the larger of the d-axis current detection value Idr and the d-axis current command value Ido is selected for the calculation of the differential inductance dLf. You may use it.

<Reduction of the set data amount of the differential inductance dLf>
As shown in FIG. 12, the field current If at the bending point where the differential inductance dLf changes significantly with respect to the change in the field current If changes according to the d-axis current Id. Therefore, in the table data as shown in FIG. 13, in order to improve the setting accuracy of the differential inductance dLf, it is necessary to make the step size of the search axis near the bending point finer. However, since the bending point changes according to the d-axis current Id, when trying to improve the accuracy in all the d-axis current Id, the step size of the search axis of the field current If and the search axis of the d-axis current Id It is necessary to make the step size finer, which increases the amount of data.

Therefore, a method of reducing the set data amount of the differential inductance dLf will be examined. The total magnetic flux of the rotor is given by the following equation.

When the turn ratio Kn of the armature winding to the field winding is given by the equation (25), the total magnetic flux of the rotor can be expressed by the equation (26). The turns ratio Kn is the ratio of the d-axis inductance Ld of the armature winding to the inductance Lf of the field winding when magnetic saturation does not occur.

That is, the total magnetic flux of the rotor can be expressed by a converted total current value which is a value obtained by multiplying the turns ratio Kn by the d-axis current Id and adding the field current If. Therefore, the differential inductance dLf can be expressed using the converted total current value.

Therefore, the voltage command value calculation unit 352 calculates the differential inductance dLf based on the converted total current value, which is the value obtained by multiplying the turns ratio Kn by the d-axis current Id and adding the field current If. For example, the voltage command value calculation unit 352 calculates the differential inductance dLf corresponding to the current converted total current value by referring to the setting data in which the relationship between the converted total current value and the differential inductance is set in advance. As the setting data, table data having the converted total current value as the search axis as shown in FIG. 14 can be used. That is, since the step size may be made finer at one bending point in the converted total current value, the amount of data can be reduced.

<Example of diversion>
In each of the above embodiments. The case where a set of three-phase armature windings is provided on the stator has been described as an example. However, a plurality of sets (for example, two sets) of armature windings may be provided on the stator, and armature current control units of an inverter and a control device may be provided corresponding to each set of armature windings. Also in this case, in the second embodiment, the d-axis current, which is the total current of the plurality of sets of armature windings, may be used.

Although the present application describes various exemplary embodiments and examples, the various features, embodiments, and functions described in one or more embodiments are applications of a particular embodiment. It is not limited to, but can be applied to embodiments alone or in various combinations. Therefore, innumerable variations not illustrated are envisioned within the scope of the techniques disclosed herein. For example, it is assumed that at least one component is modified, added or omitted, and further, at least one component is extracted and combined with the components of other embodiments.

4 Field winding, 11 AC rotating machine control device, 12 Armature winding, 14 Rotor, 18 stator, Id d-axis current, If field current, Ifo field current command value, Ifr field current detection value , Kn turns ratio, Kc conversion coefficient, Kpf proportional gain, Ld d-axis inductance, dLf differential inductance, fcf response frequency

Claims (17)

A control device for an AC rotor that controls an AC rotor having a rotor provided with field windings and a stator provided with armature windings.
The field current, which is the current flowing through the field winding, is detected, and the field current command value and the field current detection value are set so that the detected value of the field current approaches the field current command value. The field voltage command value is calculated by performing proportional integration control with respect to the deviation of, and a voltage is applied to the field winding based on the field voltage command value.
The ratio of the change in the magnetic flux of the rotor to the change in the field current at each operating point of the field current is defined as the differential inductance.
An AC rotating machine that sets the proportional gain based on the information of the field current so that the proportional gain of the proportional integration control changes in proportion to the differential inductance that changes according to the field current. Control device. Based on the field current information, the differential inductance is calculated.
The control device for an AC rotating machine according to claim 1, wherein the differential inductance is multiplied by a preset conversion coefficient to calculate the proportional gain. Based on the field current information, the differential inductance is calculated.
The proportional gain is Kpf, the response frequency of the feedback control system from the field current command value to the detected value of the field current is fcf, the differential inductance is dLf, and a positive value of 1 or less is preset. Set the adjustment gain to KL and set it to KL.
Kpf = KL x 2π x fcf x dLf
The control device for an AC rotary machine according to claim 1 or 2, wherein the proportional gain is calculated by using the calculation formula of. Based on the field current information, the differential inductance is calculated.
The proportional gain is Kpf, the integrated gain of the proportional integral control is Kif, the response frequency of the feedback control system from the field current command value to the detected value of the field current is fcf, and the differential inductance is dLf. The resistance of the field winding is Rf, and the adjustment gain preset to a positive value of 1 or less is KL.
Kpf = KL x 2π x fcf x dLf
Kif = KL x 2π x fcf x Rf
The control device for an AC rotating machine according to claim 1 or 2, wherein the proportional gain and the integrated gain are calculated by using the calculation formula of the above. The control device for an AC rotor according to claim 4, wherein the adjustment gain is determined based on the dead time in the field current control system and the response frequency. An AC voltage is applied to the armature winding to control the armature current, which is the current flowing through the armature winding, and the d-axis current, which is a component of the armature current in the magnetic pole direction of the rotor, is changed from 0. Change,
Based on the field current information and the d-axis current information, the proportional gain changes in proportion to the field current and the differential inductance that changes according to the d-axis current. The control device for an AC rotating machine according to any one of claims 1 to 5, wherein the gain is set. A control device for an AC rotor that controls an AC rotor having a rotor provided with field windings and a stator provided with armature windings.
The field current, which is the current flowing through the field winding, is detected, and the field current command value and the field current detection value are set so that the detected value of the field current approaches the field current command value. The field voltage command value is calculated by performing proportional integration control with respect to the deviation of, and a voltage is applied to the field winding based on the field voltage command value.
An AC voltage is applied to the armature winding to control the armature current, which is the current flowing through the armature winding, and the d-axis current, which is a component of the armature current in the magnetic pole direction of the rotor, is changed from 0. Change,
A control device for an AC rotating machine that sets a proportional gain of the proportional integral control based on the field current information and the d-axis current information. The ratio of the change in the magnetic flux of the rotor to the change in the field current at each operating point of the field current is defined as the differential inductance.
The differential inductance is calculated based on the field current information and the d-axis current information.
The control device for an AC rotating machine according to claim 6 or 7, wherein the differential inductance is multiplied by a preset conversion coefficient to calculate the proportional gain. The ratio of the change in the magnetic flux of the rotor to the change in the field current at each operating point of the field current is defined as the differential inductance.
The differentiation is based on the converted total current value, which is the value obtained by multiplying the value of the d-axis current by the turns ratio of the armature winding to the field winding and adding the value of the field current. The control device for an AC rotating machine according to claim 6 or 7, wherein the inductance is calculated. The number of turns ratio is the ratio of the components in the magnetic pole direction of the rotor to the inductance of the field winding when magnetic saturation does not occur and the inductance of the armature winding when magnetic saturation does not occur. Item 9. The control device for the AC rotating machine according to Item 9. The AC rotating machine according to claim 9 or 10, wherein the differential inductance corresponding to the current converted total current value is calculated by referring to the setting data in which the relationship between the converted total current value and the differential inductance is set in advance. Control device. The armature current command value and the armature current detection value are set so that the armature current, which is the current flowing through the armature winding, is detected and the detected value of the armature current approaches the armature current command value. The armature voltage command value is calculated based on the above, and a voltage is applied to the armature winding based on the armature voltage command value.
The control of the AC rotating machine according to any one of claims 6 to 11, wherein the armature current command value or the component in the magnetic pole direction of the rotor in the detected value of the armature current is used as the information of the d-axis current. Device. Claim 12 uses, as the d-axis current information, the larger of the component in the magnetic pole direction of the rotor in the armature current command value and the component in the magnetic pole direction of the rotor in the detected value of the armature current. The control device for the AC rotating machine described in 1. The response frequency of the feedback control system from the field current command value to the detected value of the field current is lower than the response frequency of the feedback control system from the armature current command value to the detected value of the armature current. The control device for an AC rotating machine according to claim 12 or 13, wherein the field current control cycle is longer than the armature current control cycle. The control device for an AC rotating machine according to any one of claims 1 to 14, which uses the detected value of the field current as the information of the field current. As the field current information, the field current command value is calculated by performing a response delay process corresponding to a feedback control system from the field current command value to the detected value of the field current. The control device for an AC rotating machine according to any one of claims 1 to 14, which uses an estimated value of field current. The control device for an AC rotary machine according to any one of claims 1 to 16, wherein the AC rotary machine is a generator motor for a vehicle.

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