JP2019103257A - Power conversion system and control method of the same - Google Patents

Abstract

To provide a power conversion system with high performance, capable of suppressing a switching loss, and a control method.SOLUTION: A power conversion system comprises: a first bridge circuit converting a DC voltage input into an AC voltage; a high frequency transformer transforming the AC voltage converted in the first bridge circuit; a second bridge circuit magnetically coupled with the first bridge circuit through the high frequency transformer, and converting the AC voltage transformed by the high frequency transformer into the DC voltage; and control means of controlling at least one operation of the first bridge circuit and the second bridge circuit. The high frequency transformer includes: a primary winding connected to the first bridge circuit; a secondary winding connected to the second bridge circuit; and an iron core to which the first bridge circuit and the second bridge circuit are wound. The first bridge circuit and the second bridge circuit are alternately wound to the circumference of the iron core so as to be oppositely arranged.SELECTED DRAWING: Figure 1

Description

  The present invention relates to the structure of a power conversion system and its control, and more particularly to a technology effectively applied to a large capacity power conversion system mounted on wind power generation, a railway system or the like.

  The power conversion system known in the prior art converts an input DC voltage into an AC voltage by an inverter circuit, transforms this into a DC voltage using a high frequency transformer, and converts the DC voltage by a rectifier circuit. Output system.

  Power conversion systems for such applications are also called DC-DC converters. When the power to be handled is large, a full bridge circuit is generally used as an inverter circuit. The switching elements constituting the full bridge circuit are inefficient because switching loss occurs at each switching.

  Therefore, Patent Document 1 discloses a DC-DC converter which reduces switching loss and improves efficiency. By connecting a resonance capacitor and a resonance reactor that cause voltage resonance or current resonance to the full bridge circuit, the voltage waveform across the primary winding of the high frequency transformer or all the current waveforms flowing through the primary winding Alternatively, the resonance is used to sinusoidally change a portion to realize zero voltage switching or zero current switching. A DC-DC converter to which such passive components for inducing resonance are applied is generally called a resonant DC-DC converter.

  Further, Patent Document 2 discloses a technology capable of realizing zero voltage switching at the time of turning off of a switching element without such a passive component for resonance. According to this, a snubber capacitor connected in parallel to the switching element and a means for observing the voltage of the snubber capacitor are required.

JP 2003-47245 A JP 2002-238257 A

  The rated capacity of the conventional DC-DC converter is at most about 10 kW at most, and the prior art is not always effective for a large-capacity DC-DC converter exceeding 100 kW. From the viewpoint of loss reduction, the rated capacity of the DC-DC converter and the drive frequency are generally in inverse proportion to each other. That is, in order to realize a large capacity DC-DC converter, it is necessary to lower the driving frequency. When the rated capacity is large and the driving frequency is low, the volume, weight and cost of the resonance capacitor, the resonance reactor and the snubber capacitor as disclosed in Patent Document 1 and Patent Document 2 increase. It is disadvantageous for downsizing and cost reduction of the DC-DC converter.

  Therefore, an object of the present invention is to provide a highly efficient power conversion system capable of suppressing switching loss without adding passive components such as a resonance capacitor, a resonance reactor, and a snubber capacitor, and a control method thereof. .

  In order to solve the above problems, the present invention provides a first bridge circuit that converts an input DC voltage into an AC voltage, and a high frequency transformer that transforms the AC voltage converted by the first bridge circuit. A second bridge circuit magnetically coupled to the first bridge circuit via the high frequency transformer and converting an alternating voltage transformed by the high frequency transformer into a direct current voltage; and the first bridge circuit; Control means for controlling the operation of at least one of the second bridge circuit, the high frequency transformer being a primary winding connected to the first bridge circuit, and the second bridge The secondary winding connected to the circuit, and the iron core around which the primary winding and the secondary winding are wound, the primary winding and the secondary winding face each other Around the iron core to be placed Characterized in that it is wound alternately.

  Also, the present invention is the power conversion system described above, wherein the first bridge circuit includes a first switching leg in which a first switching element and a second switching element are connected in series; Three switching elements and a fourth switching element connected in series, and a second switching leg connected in parallel with the first switching leg, and between the two ends of the first switching leg and the second switching leg Between both ends of the second switching leg is a first DC terminal, and a connection point of the first switching element and the second switching element and a connection point of the third switching element and the fourth switching element And a first smoothing capacitor connected between the first DC terminals, and a first smoothing capacitor connected between the first DC terminals, A second switching element, a third switching element, and a diode connected in anti-parallel with an output capacitance connected in parallel to each of the fourth switching element; It is characterized in that the primary winding is connected between alternating current terminals of one.

  The present invention is the control method of a power conversion system for controlling the power conversion system described above, wherein the fourth switching element is turned off after the second switching element is turned on, and the fourth switching element is turned off. After turning on the first switching element, the third switching element is turned off later.

  Further, according to the present invention, in the power conversion system control method for controlling the power conversion system described above, the second switching element is turned on, and the fourth switching element is turned off synchronously. The first switching element is turned on, and the third switching element is turned off synchronously.

  According to the present invention, it is possible to provide a highly efficient power conversion system capable of suppressing switching loss and a control method therefor without adding passive components such as a resonant capacitor, a resonant reactor, and a snubber capacitor.

  Problems, configurations, and effects other than those described above will be clarified by the description of the embodiments below.

It is a circuit block diagram of the power conversion system concerning one embodiment of the present invention. Example 1 It is a figure which shows the structure of high frequency transformer T1 which concerns on one Embodiment of this invention. Example 1 It is a horizontal sectional view which shows the X-X 'cross section in FIG. FIG. 3 is a vertical sectional view showing a Y-Y ′ cross section in FIG. 2; It is a figure which shows the comparative example of FIG. 3B. FIG. 6 is a voltage / current waveform diagram showing the circuit operation of the power conversion system according to the embodiment of the present invention. It is a circuit diagram which shows the circuit operation (mode 1) of period (1) in FIG. It is a circuit diagram showing circuit operation (mode 2-1) of the first half among circuit operations (mode 2) of period (2) in FIG. FIG. 10 is a current waveform diagram showing the relationship between the primary winding current and the S1 drain-source current in the circuit operation (mode 2). FIG. 6 is a circuit diagram showing a second half circuit operation (mode 2-2) of the circuit operation (mode 2) in a period (2) in FIG. 5; It is a circuit diagram which shows the circuit operation (mode 3) of period (3) in FIG. It is a circuit diagram which shows the circuit operation (mode 4) of period (4) in FIG. It is a circuit diagram which shows the circuit operation (mode 5) of period (5) in FIG. It is a circuit diagram showing circuit operation (mode 6-1) of the first half among circuit operations (mode 6) of period (6) in FIG. FIG. 6 is a circuit diagram showing a second half circuit operation (mode 6-2) of the circuit operation (mode 6) in a period (6) in FIG. 5; It is a circuit diagram which shows the circuit operation (mode 7) of period (7) in FIG. It is a circuit diagram which shows the circuit operation (mode 8) of period (8) in FIG. It is a vertical sectional view showing the composition of high frequency transformer T1 concerning one embodiment of the present invention. (Example 2) It is a circuit block diagram of the power conversion system concerning one embodiment of the present invention. (Example 3) It is a figure showing the composition of the power conversion system concerning one embodiment of the present invention. (Example 4) It is a figure showing the composition of the power conversion system concerning one embodiment of the present invention. (Example 4) It is a figure showing the composition of the power conversion system concerning one embodiment of the present invention. (Example 4) It is a figure showing the composition of the power conversion system concerning one embodiment of the present invention. (Example 4) It is a gate drive signal waveform diagram which shows the control method by the conventional system (phase shift system). It is a gate drive signal waveform diagram which shows the control method concerning one Embodiment of this invention. (Example 5)

  Hereinafter, embodiments of the present invention will be described with reference to the drawings. In the drawings, the same components are denoted by the same reference numerals, and detailed descriptions of overlapping components are omitted.

  The power conversion system of the first embodiment and a control method thereof will be described with reference to FIGS. 1 to 16.

  FIG. 1 is a circuit diagram of a power conversion system 1 according to a first embodiment of the present invention. As shown in FIG. 1, the power conversion system 1 of this embodiment boosts or steps down the input voltage Vin to output an output voltage Vout. The input voltage Vin may be replaced with the output of another power conversion system that outputs a DC voltage.

  In FIG. 1, a smoothing capacitor Cin is connected between the DC terminals A and A 'of the full bridge circuit 11, and an input voltage Vin is applied. The primary winding W1 and the secondary winding W2 of the high-frequency transformer T1 are connected between the AC terminals B-B 'of the full bridge circuit 11 and between the AC terminals C-C' of the rectifier circuit 21, respectively.

  The high frequency transformer T1 transforms the voltage between the AC terminals B and B 'according to the turns ratio of the primary winding W1 and the secondary winding W2 and outputs the voltage between the AC terminals C and C'. The full bridge circuit 11 includes a first switching leg 12 in which first and second switching elements S1 and S2 are connected in series, and a second switching leg 13 in which third and fourth switching elements S3 and S4 are connected in series. And consists of

  Antiparallel diodes D1 to D4 are connected to the switching elements S1 to S4, respectively. Here, in FIG. 1, the switching elements S1 to S4 are IGBTs as an example. The switching elements S1 to S4 have output capacitances C1 to C4. The output capacitances C1 to C4 are represented in a form connected in parallel with the switching elements S1 to S4, respectively. A leakage inductance L1 is present in the primary winding W1 of the high frequency transformer T1. The leakage inductance L1 is inserted between the AC terminals B and B 'of the full bridge circuit 11 and the primary winding W1 in the circuit diagram.

  The rectifier circuit 21 includes a first diode leg 22 in which diodes D5 and D6 are connected in series, and a second diode leg 23 in which the diodes D7 and D8 are connected in series and connected in parallel to the first diode leg 22; It is comprised by the smoothing reactor L2 and the smoothing capacitor Cout. One end of the smoothing reactor L 2 is connected to one end of the first diode leg 22 and one end of the second diode leg 23. The other end of the smoothing reactor L2 is connected to one end of the smoothing capacitor Cout. The other end of the first diode leg 22 and the other end of the second diode leg 23 are connected to the other end of the smoothing capacitor Cout. An AC terminal C which is a connection point of the diodes D5 and D6 is connected to one end of the secondary winding W2, and an AC terminal C 'which is a connection point of the diodes D7 and D8 is connected to the other end of the secondary winding W2.

  With such a configuration of the rectifier circuit 21, a diode with a small reverse withstand voltage can be used. Such a configuration is suitable when the output voltage is high. The smoothing reactor L2 can be omitted when the output of the power conversion system 1 is connected to a large inductance component such as a transmission line, or when the power conversion system 1 exchanges power in both directions as described later. There is a case.

  The control means 31 controls the power conversion system 1 to satisfy a desired operation. FIG. 1 shows the case of controlling the output voltage Vout as an example. The control means 31 observes the output voltage Vout by a voltage sensor not shown in FIG. 1, and sends gate drive signals to the switching elements S1 to S4 so that the output voltage Vout becomes a command value. The control means 31 is not limited to control of the output voltage Vout, and is effective for any other control, and can control the input voltage Vin, input power, output power and the like.

  FIG. 2 shows a configuration example of the high frequency transformer T1. The high frequency transformer T1 is composed of a winding body 41 mainly composed of a primary winding W1 and a secondary winding W2, and an iron core 4. The primary windings W1 and W2 are respectively wound around the legs of the iron core 4 and are magnetically coupled to each other. The configuration in which the iron core 4 is disposed inside the winding body 41 (primary winding W1 and secondary winding W2) in this way is called an inner iron core, and the weight of the iron core can be reduced, so the high frequency transformer T1 can be reduced in weight. .

  Furthermore, the winding body 41 is formed on both legs of the iron core 4, and one end of the primary winding W1 of the left leg and one end of the primary winding W1 of the right leg are electrically connected by the first wiring 45. . Further, one end of the left secondary winding W2 and one end of the right secondary winding W2 are electrically connected by the second wiring 46. The other end of the left primary winding W1 and the other end of the right primary winding W1 are respectively AC terminals B and B ', and the other end of the left secondary winding W2 and the right secondary The other ends of the winding W2 are referred to as AC terminals C and C ', respectively.

  With such a configuration, the high frequency transformer T1 has a symmetrical structure and is easy to install. In addition, since it is possible to shorten the total circumferential length of each of the primary winding W1 and the secondary winding W2 by setting the number of winding bodies 41 to two rather than one, it is possible to reduce the winding loss. Can.

  FIGS. 3A and 3B show an example of the X-X 'horizontal sectional view and the Y-Y' vertical sectional view of the winding body 41, respectively. In addition, although X-X 'and Y-Y' are shown by the winding body 41 of the right leg in FIG. 2, you may think that the winding body 41 of a left leg also has the same cross section.

  In FIGS. 3A and 3B, primary windings W1 and W2 are thin plate-like conductors. The material of these thin plate-like conductors is preferably copper or an alloy containing copper, but not limited thereto. As shown in FIG. 3A, the largest area plane of the primary winding W1 and the largest area plane of the secondary winding W2 are opposed to each other. The winding pair 42, which is a pair of the primary winding W1 and the secondary winding W2, is configured by being wound around the leg of the iron core 4 such that the vertical direction of the largest area face thereof becomes the radial direction. . The direction of the current flowing through the secondary winding W2 is opposite to the current flowing through the primary winding W1, and in FIG. 3B, the current of the primary winding W1 flows from the front side to the back side of the paper and the current of the secondary winding W2. Shows an example of flowing from the back side to the front side of the paper.

  By configuring the primary winding W1 and the secondary winding W2 with a thin plate-like conductor, the skin effect that is noticeable in the high frequency range is suppressed, and the winding resistance in the high frequency range is reduced. Line loss can be reduced. Further, by arranging the thin plate-like primary windings W1 and W2 to be opposed to each other, the proximity effect that is noticeable in a high frequency region is suppressed, and the winding resistance in a high frequency region is further reduced. Can. In addition, such an arrangement also has the effect of reducing the leakage inductance.

  The comparative example of Y-Y 'perpendicular | vertical sectional drawing (FIG. 3B) of the winding body 41 of a present Example is shown in FIG. The winding body 41 shown in FIG. 4 is configured by first winding the primary winding W1 around the legs of the iron core 4 and then winding the secondary winding W2 around the outside of the primary winding W1. . In such a configuration, the current direction of the adjacent windings is the same except between the primary winding W1 and the secondary winding W2, so that the magnetic field generated by the flow of the current is in the same direction. The leakage inductance L1 becomes large.

  On the other hand, in the winding body 41 according to the present embodiment shown in FIG. 3B, since the current directions of the adjacent windings are alternated, the magnetic fields generated by the flow of the currents are in a mutually canceling relationship. . Thereby, the leakage inductance L1 can be reduced.

  As a result of the trial evaluation by the inventors of the present application, the leakage inductance in the winding configuration shown in FIGS. 3A and 3B is about 1/15 times as large as the leakage inductance in the winding configuration shown in FIG. From this, it is clear that the difference in the winding configuration has an influence on the leakage inductance value.

  Although FIGS. 3A and 3B are described on the assumption that the primary winding W1 and the secondary winding W2 face each other over the entire range, the primary winding W1 and the secondary winding W2 are the same. It is not necessary to be opposite in all areas. The larger the facing area, the greater the effect of reducing the leakage inductance. However, by making the primary winding W1 and part of the secondary winding W2 opposite, the shape, volume and weight of the high-frequency transformer T1 are designed. You can increase your freedom.

  The influence of the leakage inductance L1 which is reduced by arranging the primary winding W1 and the secondary winding W2 opposite to each other to the loss of the power converter 1 will be described. In the following description, a voltage equivalent to or less than the voltage of the on-state switching element or the forward drop voltage of the antiparallel diode connected in parallel with the switching element is referred to as a zero voltage, Switching on and off of the switching element when the applied voltage is at zero voltage is referred to as zero voltage switching.

  In addition, even if the voltage applied to the switching element is not zero voltage, when the voltage is suppressed to a low voltage and a reduction in switching loss is expected, it is referred to as incomplete zero voltage switching. Similarly, with respect to the current, a current equal to or less than the drain-source current of the switching element in the off state or the reverse current of the antiparallel diode connected in parallel with the switching element is referred to as a zero current. The switching on and off of the switching element in a state where the current flowing through the switching element is zero current is referred to as zero current switching.

  The circuit operation (control method) of the power conversion system 1 will be described using FIGS. 5 to 16. FIG. 5 is a voltage / current waveform diagram for explaining the circuit operation of the power conversion system 1. From top to bottom in FIG. 5, gate-source voltages of S1 to S4, voltages and currents across primary windings, and drain-source voltages and currents of S1 to S4 are shown.

  The S1-S4 gate-source voltage is a voltage generated between the gate-source of the switching elements S1-S4 as the control means 31 outputs a drive signal to the switching elements S1-S4. The switching elements S1 to S4 are turned on when the drive signal waveforms output to the switching elements S1 to S4 become high, and are turned off when the driving signal waveform becomes low.

  The voltage and current across the primary winding are the voltage generated between the terminals B and B 'in FIG. 1 and the current flowing from the terminal B to the terminal B' through the leakage inductance L1 and the primary winding W1. is there. With respect to the primary winding current, a direction flowing from the terminal B as a base point to the terminal B 'through the leakage inductance L1 and the primary winding W1 is positive. The drain-source voltages and currents of S1 to S4 are drain-source voltages and currents of the switching elements S1 to S4. The drain-source current has a positive flow direction from the drain to the source. Periods (1) to (8) divided by dotted lines in FIG. 5 respectively correspond to (mode 1) to (mode 8) described below.

  6 to 16 show current paths of the power conversion system 1 corresponding to (Mode 1) to (Mode 8). 6, 7 and 9 to 16 show only the configuration necessary for the description of (mode 1) to (mode 8), and the description of the parts where the description is unnecessary is omitted. Also, (mode 5) to (mode 8) are in a dual relationship with (mode 1) to (mode 4), respectively. That is, the switching element S4 in (Mode 1) to (Mode 4) corresponds to the switching element S3, and the switching element S1 in (Mode 1) to (Mode 4) corresponds to the switching element S2.

(Mode 1)
FIG. 6 is a circuit diagram for explaining the circuit operation (mode 1) in period (1) shown in FIG. The (mode 1) is a mode following the (mode 8) described later. The switching elements S2 and S3 are off, and the switching element S1 is on. In this state, the switching element S4 is turned on. In (Mode 8), since the S4 drain-source current is zero current, turning on of the switching element S4 is zero current switching. Since the switching elements S1 and S4 are turned on by turning on the switching element S4, Vin is applied to the primary winding W1, and the current flowing through the primary winding W1 increases in the positive direction.

(Mode 2)
In (mode 2), the switching element S1 is turned off. The (mode 2) is divided into (mode 2-1) and (mode 2-2) described below.

(Mode 2-1)
FIG. 7 is a circuit diagram for explaining the first half (mode 2-1) of the circuit operation (mode 2) in the period (2) shown in FIG. When the switching element S1 is turned off, the S1 drain-source current decays rapidly. On the other hand, since the leakage inductance L1 is small due to the above-described winding configuration, the primary winding current is also rapidly attenuated when the switching element S1 is turned off.

  The relationship between the primary winding current and the S1 drain-source current at this time is shown in FIG. The primary winding current is larger than the S1 drain-source current. The difference is the discharge current from the output capacitance C2 of the switching element S2. For this reason, the S2 drain-source voltage decreases. The sum of the S1 drain-source voltage and the S2 drain-source voltage is the input voltage Vin, and the input voltage Vin can be considered constant, so the S1 drain-source voltage rises by the amount of the decrease in the S2 drain-source voltage.

  That is, at the same time as the discharge of the output capacity C2, the output capacity C1 is charged. Since the switching element S1 is an IGBT, tail current continues to flow for a while even after the turning off. On the other hand, the attenuation of the primary winding current proceeds because the leakage inductance L1 is small. Therefore, as shown in FIG. 8, the primary winding current is smaller than the S1 drain-source current. When the primary winding current becomes smaller than the S1 drain-source current, transition to (mode 2-2) is made.

(Mode 2-2)
FIG. 9 is a circuit diagram for explaining the second half (mode 2-2) of the circuit operation (mode 2) in the period (2) shown in FIG. When the primary winding current becomes smaller than the S1 drain-source current, charging of the output capacitance C1 and discharging of the output capacitance C2 are stopped, and conversely, the output capacitance C1 is discharged and the output capacitance C2 is charged. This causes the S1 drain-to-source voltage to drop to near zero and the S2 drain-to-source voltage to rise to be approximately equal to Vin. Eventually, the tail current of the switching element S1 ends and the S1 drain-source current becomes zero, and the energy stored in the leakage inductance L1 also becomes almost zero, so the primary winding current becomes almost zero.

  As described above, the output capacitor C1 is not sufficiently charged in (mode 2-1), and the switching element S1 is turned off while the output capacitor C1 is discharged in (mode 2-2). Therefore, although switching element S1 becomes imperfect zero voltage switching, the switching loss is reduced significantly compared with the time of normal switching.

(Mode 3)
FIG. 10 is a circuit diagram for explaining the circuit operation (mode 3) in period (3) shown in FIG. In (mode 3), the switching element S2 is turned on. At this time, the switching elements S2 and S4 are in the on state. Since the primary winding current is already zero in (mode 2-2), the switching element S2 can be regarded as almost zero current switching. Since the output capacitance C2 is discharged by turning on the switching element S2, the S2 drain-source voltage decreases and becomes a zero voltage. Also, since the output capacitance C1 is charged, the S1 drain-source voltage rises and becomes approximately equal to the input voltage Vin. When the discharge of the output capacitance C2 and the charging of the output capacitance C1 are completed, the S2 drain-source current and the S1 drain-source current become zero current.

(Mode 4)
FIG. 11 is a circuit diagram for explaining the circuit operation (mode 4) in period (4) shown in FIG. In (mode 4), the switching element S4 is turned off. Since S4 drain-source current can be regarded as almost zero, S4 is zero current switching. The slight current circulating in the circuit gradually charges the output capacity C4 and slightly raises the S4 drain-source voltage. At the same time, the output capacity C3 is gradually discharged, and the S3 drain-source voltage slightly decreases.

(Mode 5)
FIG. 12 is a circuit diagram for explaining the circuit operation (mode 5) in period (5) shown in FIG. The switching elements S1 and S4 are off, and the switching element S2 is on. In this state, switching element S3 is turned on. In (Mode 4), since the S3 drain-source current is zero current, turning on of the switching element S3 is zero current switching. Since the switching elements S2 and S3 are turned on by turning on the switching element S3, -Vin is applied to the primary winding W1, and the current flowing through the primary winding W1 increases in the negative direction.

(Mode 6)
In (Mode 6), the switching element S2 is turned off. Similarly to (Mode 2), (Mode 6) is divided into (Mode 6-1) and (Mode 6-2) described below.

(Mode 6-1)
FIG. 13 is a circuit diagram for explaining the first half (mode 6-1) of the circuit operation (mode 6) in the period (6) shown in FIG. Similar to the principle described in (Mode 2), when the switching element S2 is turned off, the S2 drain-source current decays rapidly. At this time, the output capacitance C2 of the switching element S2 is charged, the S2 drain-source voltage rises, the output capacitance C1 of the switching element S1 is discharged, and the S1 drain-source voltage drops. When the primary winding current becomes smaller than the S2 drain-source current, transition to (mode 6-2) is made.

(Mode 6-2)
FIG. 14 is a circuit diagram for explaining the second half (mode 6-2) of the circuit operation (mode 6) in the period (6) shown in FIG. When the primary winding current becomes smaller than the S2 drain-source current, the charging of the output capacitor C2 and the discharging of the output capacitor C1 stop, and conversely, the output capacitor C2 is discharged and the output capacitor C1 is charged. This causes the S2 drain-to-source voltage to drop to near zero and the S1 drain-to-source voltage to rise to be approximately equal to Vin. Eventually, the tail current of the switching element S2 ends and the S2 drain-source current becomes zero, and the energy stored in the leakage inductance L1 also becomes almost zero, so the primary winding current becomes almost zero.

  As described above, the output capacitor C2 is not sufficiently charged in (mode 6-1), and the switching element S2 is turned off while the output capacitor C2 is discharged in (mode 6-2). Thus, although the switching element S2 performs incomplete zero voltage switching, its switching loss is significantly reduced as compared to that during normal switching.

(Mode 7)
FIG. 15 is a circuit diagram for explaining the circuit operation (mode 7) in period (7) shown in FIG. In (mode 7), the switching element S1 is turned on. At this time, the switching elements S1 and S3 are in the on state. Since the primary winding current is already zero in (mode 6-2), the switching element S1 can be regarded as almost zero current switching. Since the output capacitance C1 is discharged by turning on the switching element S1, the drain-to-source voltage of the S1 drops and becomes a zero voltage. Also, since the output capacitance C2 is charged, the S2 drain-source voltage rises and becomes approximately equal to the input voltage Vin. When the discharge of the output capacitance C1 and the charging of the output capacitance C2 are completed, the S1 drain-source current and the S2 drain-source current become zero current.

(Mode 8)
FIG. 16 is a circuit diagram for explaining the circuit operation (mode 8) in period (8) shown in FIG. In (mode 8), the switching element S3 is turned off. S3 is zero current switching since the S3 drain-source current can be considered nearly zero. The slight current circulating in the circuit gradually charges the output capacitance C3 and slightly raises the S3 drain-source voltage. At the same time, the output capacity C4 is gradually discharged and the S4 drain-source voltage is slightly reduced.

  As described above, according to this embodiment, there is no additional passive component such as a capacitor for resonance, a reactor for resonance, a snubber capacitor, etc., which are required in the prior art such as Patent Document 1 and Patent Document 2 above. In either case, switching related to turn-on and turn-off of the switching elements S1 to S4 is any one of zero voltage switching, incomplete zero voltage switching, and zero current switching, thereby suppressing the switching loss of the switching elements S1 to S4 and highly efficient power A conversion system can be provided.

  Furthermore, high efficiency can simplify the heat dissipation means installed to enhance the heat dissipation of the switching elements S1 to S4, so that miniaturization, cost reduction, and high reliability of the power conversion system can be expected. .

  The circuit operation of the power conversion system 1 may operate differently from (Mode 1) to (Mode 8) described above. For example, in (Mode 1) to (Mode 8), no current flows in the diodes D1 to D4 connected in parallel to the switching elements S1 to S4. However, depending on the value of leakage inductance L1, the switching speed of switching elements S1 to S4, the value of output capacitances C1 to C4 of switching elements S1 to S4, the circuit connected to the output side of power conversion system 1, etc. Current may flow to D4. The above-described (mode 1) to (mode 8) is merely an example that briefly describes the features of the present embodiment.

  The driving method (control method) of the switching elements S1 to S4 shown in (Mode 1) to (Mode 8) is generally called a phase shift method. Depending on the value of leakage inductance L1, the switching speed of switching elements S1 to S4, the value of output capacitances C1 to C4 of switching elements S1 to S4, the circuit connected to the output of power conversion system 1, etc. In some cases, the effects of Therefore, the scope of application of the present invention is not limited to the phase shift method.

  A power conversion system according to a second embodiment will be described with reference to FIG. FIG. 17 is a vertical sectional view showing another winding configuration of the high frequency transformer T1 constituting the power conversion system 1 according to the present embodiment.

  The shapes of the cross sections of the primary winding W1 and the secondary winding W2 are respectively rectangular, and the primary winding W1 and the secondary winding W2 are alternately arranged in the order of the primary winding W1 and the secondary winding W2. A winding layer 43 is formed. Next to the winding layer 43, the primary winding W1 and the primary winding W2 are alternately arranged in the order of the secondary winding W2 and the primary winding W1 to form a winding layer 44. That is, in the vertical cross section of the winding layers 43 and 44, the primary windings W1 and W2 are arranged in a staggered manner. A winding layer 43 and a winding layer 44 are paired, and wound around the legs of the iron core 4 to constitute a high frequency transformer T1.

  In other words, the primary winding W1 and the secondary winding W2 are disposed to face each other in the radial direction of the iron core 4, and also disposed to face each other in the circumferential direction of the iron core 4.

  With this configuration, the primary winding W1 and the secondary winding W2 in the winding layer 43 face each other, and the primary winding is also between the winding layer 43 and the winding layer 44. W1 and W2 are oppositely disposed. Thereby, the winding loss can be suppressed as in the principle described in the first embodiment.

  When the power conversion system 1 has a small capacity and the high frequency transformer T1 is formed by a plate-like winding, the winding cross-sectional area can be reduced, but the winding shape becomes a thin strip and the desired mechanical In some cases, the strength can not be secured. In such a case, if the cross-sectional area shape of the winding is made rectangular as shown in FIG. 17, the design freedom of the high frequency transformer T1 is increased, and mechanical strength can be secured.

  Also, the cross-sectional shape of the inexpensive and available winding is generally circular. Even if the cross-sectional shape of the winding is a circle, similar effects can be expected if the arrangement described above is used.

  When the rated power of the power conversion system 1 is, for example, 10 kW or less, the drive frequency of the high frequency transformer T1 often exceeds 10 kHz. In such a case, the primary winding W1 and the secondary winding W2 may each be called a litz wire, and may be constituted by an assembly of thin wires whose surfaces are electrically insulated. By doing this, the skin effect is suppressed and the winding resistance can be further reduced.

  In addition, when the rated power of the power conversion system 1 is, for example, 100 kW or more, the driving frequency of the high frequency transformer T1 may be a high frequency equal to or higher than a commercial frequency, for example, 1 kHz. By doing this, the cross-sectional area of the iron core 4 can be reduced, and the volume of the high frequency transformer T1 can be reduced.

  Furthermore, by making the iron core 4 a wound iron core configured by winding a plurality of thin strip-shaped magnetic materials, generation of eddy current loss that becomes noticeable in a high frequency region can be suppressed, and the volume of the high frequency transformer T1 is further reduced. it can. As such a wound core, although a thin strip alloy having an amorphous structure or a fine crystal structure mainly composed of iron is preferable, it is not limited to this.

  By designing the high frequency transformer T1 as described above, it is possible to suppress the switching loss and provide a highly efficient power conversion system.

  A configuration example of the power conversion system of the third embodiment will be described with reference to FIG. The power conversion system 1 according to the present embodiment is obtained by replacing the rectifier circuit 21 of FIG. 1 (first embodiment) with the full bridge circuit 26. The configuration other than the full bridge circuit 26 is the same as that of FIG. The explanation is omitted.

  The full bridge circuit 26 includes, for example, switching elements S5 to S8 such as four IGBTs, diodes D5 to D8, and a capacitor Cout. The output capacitances C5 to C8 of the switching elements S5 to S8 are expressed in a form connected in parallel to the switching elements S5 to S8.

  More specifically, a third switching leg 24 in which diodes D5 and D6 are connected in antiparallel to two (semiconductor) switching elements S5 and S6 connected in series, and two (semiconductor) switching elements connected in series The fourth switching leg 25 in which the diodes D7 and D8 are connected in antiparallel to each of S7 and S8 is connected in parallel, and the AC terminals C and C 'of both switching legs are respectively the secondary winding W2 of the high frequency transformer T1. Connected to both ends of the

  The drive of the full bridge circuit 26 is controlled by the control means 31 in the same manner as the full bridge circuit 11. The control target can be designed arbitrarily.

  By configuring the power conversion system 1 as in this embodiment (FIG. 18), it is possible to bidirectionally exchange power between the DC voltage Vin side and the DC voltage Vout side. Such a configuration is suitable for, for example, a railway power supply system that may bidirectionally exchange power, or in a microgrid application provided with a power generation device.

  In addition, although a power generation system such as wind power generation or solar power generation is usually unidirectional transmission from the power generation system to a load, power may be required when starting the power generation system. Even in such a case, the power conversion system 1 capable of bidirectional power interchange is preferable.

  Also in the present embodiment (FIG. 18), the high frequency transformer T1 is configured as described in the first embodiment or the second embodiment, and additional passive components such as a resonance capacitor, a resonance reactor, and a snubber capacitor are provided. Even if it does not exist, the switching loss of switching element S1-S4 or switching element S5-S8 can be suppressed, and a highly efficient power conversion system can be provided.

  The configuration example of the power conversion system of the fourth embodiment will be described with reference to FIGS. 19A to 19D. The power conversion system 2 of the present embodiment is configured of a plurality of power conversion systems 1.

  FIG. 19A is an example of a parallel input / parallel output type power conversion system in which a plurality of power conversion systems 1 are connected in parallel to an input Vin and an output Vout of the power conversion system 2, respectively. 19B shows parallel input / serial output type power conversion in which a plurality of power conversion systems 1 are connected in parallel to an input Vin of a power conversion system 2 and a plurality of power conversion systems 1 are connected in series to an output Vout. It is an example of a system. FIG. 19C is an example of a serial input / serial output type power conversion system in which a plurality of power conversion systems 1 are connected in series to the input Vin and the output Vout of the power conversion system 2, respectively. 19D shows serial input / parallel output power conversion in which a plurality of power conversion systems 1 are connected in series to an input Vin of a power conversion system 2 and a plurality of power conversion systems 1 are connected in parallel to an output Vout It is an example of a system.

  As shown in FIGS. 19A to 19D, power conversion system 2 is configured by connecting the input terminals of a plurality of power conversion systems 1 in parallel or in series, and connecting the output terminals of a plurality of power conversion systems 1 in parallel or in series. Can be configured. That is, the method of configuring the power conversion system 2 is classified into four types: parallel input / parallel output type, parallel input / serial output type, serial input / serial output type, and serial input / parallel output type.

  The power conversion system 2 of the present embodiment has a control means 32. The control means 32 can communicate with the control means 31 (not shown) of the plurality of power conversion systems 1, and can also communicate with other power conversion systems, grid-connected devices, a central power supply command station, and the like.

  By setting the configuration of the power conversion system 2 to any of the above four classes according to the application, it is possible to cope with a large current or a high voltage. For example, the parallel input / parallel output type shown in FIG. 19A can cope with a large current input / large current output. The parallel input / serial output type shown in FIG. 19B can handle large current input / high voltage output. The series input / series output type shown in FIG. 19C can handle high voltage input / high voltage output. The series input / parallel output type shown in FIG. 19D can cope with high voltage input / high current output.

  Also in the present embodiment (FIGS. 19A to 19D), the high frequency transformer T1 of each of the plurality of power conversion systems 1 is configured as described in the first embodiment or the second embodiment, and a resonance capacitor, Even if there is no additional passive component such as a resonance reactor or a snubber capacitor, the switching loss of the switching elements S1 to S4 can be suppressed, and the power conversion system 2 as a whole can provide a highly efficient power conversion system.

  The control method of the power conversion system of the fifth embodiment will be described with reference to FIGS. 20A and 20B. FIG. 20A is a gate drive signal waveform diagram of a conventional method (phase shift method) shown for comparison, and FIG. 20B is a gate drive signal waveform diagram showing a control method of the present invention.

  Focusing on mode 4 and mode 8 shown in FIG. 11 and FIG. 16 respectively in the first embodiment, in mode 4 and mode 8, the current contributing to the desired operation of the power conversion system 1 does not flow. Therefore, modes 3 and 4 and modes 7 and 8 can be combined into modes 3 + 4 and 7 + 8, respectively.

  As shown in FIG. 20A, in the conventional phase shift method, after the S2 gate drive signal of the switching element S2 is turned on, the S4 gate drive signal of the switching element S4 is turned off later. Similarly, after the S1 gate drive signal of the switching element S1 is turned on, the S3 gate drive signal of the switching element S3 is turned off later.

  On the other hand, in the control method of the present invention, as shown in FIG. 20B, the S2 gate drive signal of the switching element S2 is turned on, and the S4 gate drive signal of the switching element S4 is synchronously turned off. Similarly, the S1 gate drive signal of the switching element S1 is turned on, and the S3 gate drive signal of the switching element S3 is synchronously turned off. By controlling the power conversion system 1 in this manner, the period of modes 1 and 5 becomes longer compared to the conventional phase shift method, and more power transfer becomes possible.

  In the conventional phase shift method, the duty ratio of the S1 to S4 gate drive signals is fixed at 50%. A desired control operation is realized by adjusting the overlapping condition of the S1 gate drive signal and the S4 gate drive signal (S2 gate drive signal and S3 gate drive signal), that is, the phase. In the phase shift method, the S1 gate drive signal and the S2 gate drive signal and the S2 gate drive signal are simultaneously turned on to prevent the DC terminals from being short-circuited. Dead time (both are off periods) is required between the gate drive signal and the S3 gate drive signal and the S4 gate drive signal.

  Also in the control method of the present embodiment (FIG. 20B), by configuring the high frequency transformer T1 of the power conversion system 1 to be a target as described in the first embodiment or the second embodiment, a resonance capacitor and a resonance reactor Even if there is no additional passive component such as a snubber capacitor, the switching loss of the switching elements S1 to S4 can be suppressed, and the efficiency of the power conversion system 1 can be improved.

  The present invention is not limited to the embodiments described above, but includes various modifications. For example, the above-described embodiments are described in detail in order to explain the present invention in an easy-to-understand manner, and are not necessarily limited to those having all the described configurations. It is possible to replace part of the configuration of one embodiment with the configuration of another embodiment, and it is also possible to add the configuration of another embodiment to the configuration of one embodiment. Moreover, it is also possible to add, delete, and replace other configurations for part of the configuration of each embodiment. For example, the switching elements S1 to S8 include a switching element group configured by connecting a plurality of switching elements in series or in parallel. The same applies to the other components.

  Moreover, in order to make this invention intelligible, each above-described Example does not necessarily describe all the components, and has abbreviate | omitted description of a one part structure. For example, the high-frequency transformer T1 is an insulating paper for enhancing the insulation performance of the winding, a spacer for enhancing the insulation performance of the winding and keeping the distance between the windings constant, a heat medium for cooling the winding and core, Illustration of a support for appropriately supporting the windings and cores is omitted. It may be said that the actual high frequency transformer T1 includes all of these elements.

  Further, in each of the embodiments, the control lines and the information lines indicate what is considered necessary for the description, and not all the control lines and the information lines in the product are necessarily shown. In practice, almost all configurations may be considered to be connected to each other.

1, 2 Power conversion system 4 Iron core 11, 26 Full bridge circuit 12, 13, 24 Switching leg 21 Rectifying circuit 22, 23 Diode leg 31 Control means 41 Winding body 42 Winding pair 43 , 44 ... Wire layer 45, 46 ... Wiring A, A '... DC terminal B, B', C, C '... AC terminal C1 to C8 ... Output capacitance Cin, Cout ... Smoothing capacitor D1 to D8 ... Diode L1 ... Leakage Inductance L2 ... Smoothing reactor S1 ~ S8 ... Switching element T1 ... High frequency transformer Vin ... Input (voltage)
Vout: Output (voltage)
W1 ... primary winding W2 ... secondary winding

Claims (15)

A first bridge circuit for converting an input DC voltage into an AC voltage;
A high frequency transformer for transforming the alternating voltage converted by the first bridge circuit;
A second bridge circuit magnetically coupled to the first bridge circuit via the high frequency transformer and converting an alternating voltage transformed by the high frequency transformer into a direct current voltage;
Control means for controlling the operation of at least one of the first bridge circuit and the second bridge circuit;
The high frequency transformer has a primary winding connected to the first bridge circuit.
A secondary winding connected to the second bridge circuit;
An iron core on which the primary winding and the secondary winding are wound;
The power conversion system, wherein the primary winding and the secondary winding are alternately wound around the iron core so as to be disposed to face each other. The power conversion system according to claim 1,
The power conversion system, wherein the primary winding and the secondary winding are disposed to face each other over the entire range. The power conversion system according to claim 1,
The primary winding and the secondary winding have a rectangular cross-sectional shape in each cross section,
A power conversion system characterized in that the largest area surface of the primary winding and the largest area surface of the secondary winding are opposed to each other. The power conversion system according to claim 3,
The power conversion system according to claim 1, wherein the primary winding and the secondary winding are wound around the iron core such that the vertical direction of the respective largest area faces is a radial direction. The power conversion system according to claim 1,
The power conversion system, wherein the iron core is a wound iron core configured by winding a plurality of thin magnetic material strips. The power conversion system according to claim 1,
The power conversion system characterized in that the iron core is made of a thin strip alloy having an amorphous structure or a fine crystal structure mainly composed of iron. The power conversion system according to claim 1,
The power conversion system, wherein the primary winding and the secondary winding are disposed to face each other in the radial direction of the iron core, and are disposed to face each other in the circumferential direction of the iron core. . The power conversion system according to claim 1,
The first bridge circuit includes a first switching leg in which a first switching element and a second switching element are connected in series;
A third switching element and a fourth switching element connected in series, and configured of a second switching leg connected in parallel with the first switching leg,
Between both ends of the first switching leg and between both ends of the second switching leg are between first DC terminals,
A bridge circuit in which a connection point between the first switching element and the second switching element and a connection point between the third switching element and the fourth switching element are first AC terminals,
A first smoothing capacitor connected between the first DC terminals;
The first switching element, the second switching element, the third switching element, and a diode connected in anti-parallel to an output capacitance connected in parallel to each of the fourth switching element ,
The power conversion system, wherein the primary winding is connected between the first AC terminals.   The power conversion system according to claim 8, wherein the first switching element, the second switching element, the third switching element, and the fourth switching element are IGBTs. Conversion system. The power conversion system according to claim 8,
The second bridge circuit includes a first diode leg in which a first diode and a second diode are connected in series;
A third diode and a fourth diode connected in series, and a second diode leg connected in parallel with the first diode leg,
Between both ends of the first diode leg and between both ends of the second diode leg are between second DC terminals,
A bridge circuit in which a connection point between the first diode and the second diode and a connection point between the third diode and the fourth diode are connected to a second AC terminal,
A second smoothing capacitor connected between the second DC terminals;
The power conversion system, wherein the secondary winding is connected between the second AC terminals. The power conversion system according to claim 8,
The second bridge circuit includes a third switching leg in which a fifth switching element and a sixth switching element are connected in series;
A seventh switching element and an eighth switching element connected in series, and a fourth switching leg connected in parallel with the third switching leg,
Between both ends of the third switching leg and between both ends of the fourth switching leg are between second DC terminals,
A bridge circuit in which a connection point between the fifth switching element and the sixth switching element and a connection point between the seventh switching element and the eighth switching element are between the second AC terminals,
A second smoothing capacitor connected between the second DC terminals;
The fifth switching element, the sixth switching element, the seventh switching element, and a diode connected in anti-parallel to an output capacitance connected in parallel to each of the eighth switching elements ,
The power conversion system, wherein the secondary winding is connected between the second AC terminals. The power conversion system according to any one of claims 1 to 11 is a power conversion module,
An input terminal configured by connecting in series or in parallel between input terminals of at least two or more of the power conversion modules;
An output terminal configured by connecting in series or in parallel between output terminals of at least two or more of the power conversion modules;
And control means for controlling the power conversion module. The power conversion system according to any one of claims 8 to 11, wherein
A power conversion system characterized in that the first switching element, the second switching element, the third switching element, and the fourth switching element are driven by the control means in a phase shift system. A control method of a power conversion system for controlling a power conversion system according to any one of claims 8 to 11,
After turning on the second switching element, the fourth switching element is turned off later,
A control method of a power conversion system, characterized in that the third switching element is turned off after the first switching element is turned on. A control method of a power conversion system for controlling a power conversion system according to any one of claims 8 to 11,
The second switching element is turned on, and the fourth switching element is synchronously turned off,
A control method of a power conversion system, comprising turning on the first switching element and synchronously turning off the third switching element.

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