JP2019071709A - Inverter controller and sensorless drive system - Google Patents

Abstract

To provide an inverter controller for accurately controlling current and a sensorless drive system.SOLUTION: An inverter controller comprises an estimation value calculation section 40 for calculating a rotational phase angle estimation value of a synchronous machine M on the basis of current values detected by current detection sections SU and SW and an output voltage target vector in initial estimation. The estimation value calculation section 40 uses the output voltage target vector when current synchronized with a rotor frequency is fed to the synchronous machine M and the current values detected by the current detection sections SU and SW, calculates an error Δθ of the rotational phase angle estimation value of the synchronous machine M and calculates the rotational phase angle estimation value θso that the error Δθ becomes zero with winding resistance R of the synchronous machine M, d-axis average inductance Land q-axis differential inductance Las setting values when an inverter main circuit section INV is started from a stop state.SELECTED DRAWING: Figure 1

Description

  Embodiments of the present invention relate to an inverter control device and a sensorless drive system.

  In a control device of an inverter for driving a synchronous machine having a magnet and magnetic saliency, it is desired to supply a current as designed and control the output torque of the synchronous machine with high accuracy.

  In addition, a rotation sensorless control method that does not use a rotation sensor such as a resolver or an encoder has been proposed in order to reduce the size and weight of the control device, reduce the cost, and improve the reliability. In rotation sensorless control, it is desirable to be able to accurately calculate estimated values of the rotational phase angle and rotational speed of the rotor of a synchronous machine in a wide speed range from inverter stop to maximum speed drive.

Patent No. 5534935 gazette Patent No. 4984057

Ichikawa, S. Chen, S. Shibata, M. Doki, S. Okuma, "Sensorless Control of Salient-Pole-Type Permanent Magnet Synchronous Motor Based on Extension-Induced Voltage Model," Transactions of the Institute of Electrical Engineers of Japan, D, Vol. 122, No. 12 , P. 1088-p. 1096, 2002.

  For example, in the case of a synchronous machine having magnets and magnetic saliency, when estimating the position of the rotor with the inductance as the set value, the d-axis differential inductance changes significantly depending on whether the d-axis current is positive or negative. The position estimation result may be destabilized because the set value used and the actual value deviate from each other.

  The embodiment of the present invention has been made in view of the above circumstances, and an object thereof is to provide an inverter control device and a sensorless drive system which perform current control with high accuracy.

  The inverter control device according to the embodiment includes a current command generation unit that generates a current command value, a current detection unit that detects a current value of an alternating current output from the inverter main circuit unit, the current command value, and the current detection unit. A gate command generation unit that generates a gate command of the inverter main circuit unit such that the current value detected in the step matches, and calculating an output voltage target vector of the inverter main circuit unit based on the gate command; And an estimated value calculation unit that calculates a rotational phase angle estimated value based on the current value detected by the current detection unit and the output voltage target vector in the initial estimation when the main circuit unit starts. The estimated value calculation unit may be configured to generate the output voltage target vector when a current synchronized with a rotor frequency is supplied when the inverter main circuit unit starts from a stop state. Using the current value detected by the current detection unit, the winding resistance, the d-axis average inductance, and the q-axis differential inductance are set values to calculate the error of the estimated rotational phase angle, and the error is considered to be zero. The rotation phase angle estimated value is calculated so that

FIG. 1 is a block diagram schematically showing an example of the configuration of an inverter control device and a sensorless drive system according to a first embodiment. FIG. 2 is a diagram for explaining the definition of the d-axis, the q-axis, and the estimated rotation coordinate system (dc-axis, qc-axis) in the embodiment. FIG. 3 is a diagram for explaining a configuration example of a part of the synchronous machine shown in FIG. FIG. 4 is a diagram for explaining an example of the relationship between the d-axis current and the d-axis differential inductance of the permanent magnet synchronous motor having the magnetic saliency. FIG. 5 is a diagram for explaining an example of the relationship between the d-axis current and the q-axis differential inductance of a permanent magnet synchronous motor having magnetic saliency. FIG. 6 is a diagram for explaining an example of the operation of the inverter control device of the embodiment. FIG. 7 is a diagram showing an example of a simulation result when the rotational phase angle estimated value is calculated using the extended induced voltage in the d-axis direction. FIG. 8 is a diagram showing an example of a simulation result when the rotational phase angle estimated value is calculated using the extended induced voltage in the d-axis direction. FIG. 9 is a diagram showing an example of a simulation result when the rotational phase angle estimated value is calculated using the extended induced voltage in the q-axis direction. FIG. 10 is a diagram showing an example of a simulation result when the rotational phase angle estimated value is calculated using the extended induced voltage in the q-axis direction. FIG. 11 is a block diagram schematically showing one configuration example of the inverter control device and the sensorless drive system of the second embodiment.

  Hereinafter, the inverter control device and the sensorless drive system according to the embodiment will be described in detail with reference to the drawings.

  FIG. 1 is a block diagram schematically showing an example of the configuration of an inverter control device and a sensorless drive system according to a first embodiment. The inverter control device 1 of the present embodiment is, for example, an inverter control device that controls an inverter main circuit that drives a permanent magnet synchronous motor having magnetic saliency, and is mounted in a sensorless drive system that drives a vehicle.

The sensorless drive system according to this embodiment includes a synchronous machine M, an inverter main circuit unit INV, an inverter control device 1, and a command generation unit CTR. The output torque of the synchronous machine M is transmitted to the wheels via an axle (not shown).
Inverter control device 1 includes: current command generation unit 10, dq / αβ conversion unit 20, current control unit 30, estimated value calculation unit 40, polarity determination unit 50, adder 60, 3-phase / αβ conversion And a current sensor (current detection unit) SU, SW.
The current sensor (current detection unit) detects the current value of the alternating current output from the inverter main circuit unit INV that drives the synchronous machine M.

The current command generation unit 10 receives a current amplitude command idq_ref, a current phase command β_ref, and a current conduction flag Ion from the command generation unit (upper controller) CTR. The current command generation unit 10 calculates the d-axis current command value id_ref and the q-axis current command value iq_ref corresponding to the current supplied to the synchronous machine M based on the current amplitude command idq_ref and the current phase command β_ref. When the energization flag Ion is on (high level), the value is output. The d-axis current command value id_ref and the q-axis current command value iq_ref are obtained by the following equations.
id_ref = −idq_ref · sin β_ref
iq_ref = idq_ref · cos β_ref

FIG. 2 is a diagram for explaining the definition of the d-axis, the q-axis, and the estimated rotation coordinate system (dc-axis, qc-axis) in the embodiment.
The d-axis is a vector axis at which the static inductance is minimized in the rotor of the synchronous machine M, and the q-axis is a vector axis orthogonal to the d-axis in electrical angle. On the other hand, the estimated rotational coordinate system corresponds to the d axis and the q axis at the estimated position of the rotor. That is, the vector axis rotated by the estimation error Δθ from the d axis is the dc axis, and the vector axis rotated by the estimation error Δθ from the q axis is the qc axis. The d-axis current command value id_ref obtained by the above equation is a vector value in the direction rotated 180 degrees from the dc axis, and the q-axis current command value iq_ref is a vector value in the qc axis direction. The α-axis represents the U-phase winding axis of the synchronous machine M, and the β-axis is an axis perpendicular to the α-axis in electrical angle.

  The dq / αβ conversion unit 20 receives the d-axis current command value id_ref, the q-axis current command value iq_ref, and the estimated value θest of the rotational phase angle. The dq / αβ conversion unit 20 combines the d-axis current command value id_ref and the q-axis current command value iq_ref represented in the dq-axis coordinate system with the α-axis current command value iα_ref represented in the αβ-axis fixed coordinate system. It is a vector converter that converts it into the β-axis current command value iβ_ref. The α axis indicates the U-phase winding axis of the synchronous machine M, and the β axis is an axis orthogonal to the α axis. The value represented by the fixed coordinate system of the αβ axis can be calculated without using the rotor phase angle of the synchronous machine M.

The current control unit 30 includes subtractors 32 and 34, an angle calculation unit 36, and a gate command generation unit 38.
The subtractors 32 and 34 are disposed downstream of the dq / αβ conversion unit 20. The α-axis current command value iα_ref output from the dq / αβ conversion unit 20 is input to the subtractor 32, and the β-axis current command value iβ_ref is input to the subtractor 34. In addition, the current sensors SU and SW detect current values of at least two phases output from the inverter main circuit unit INV, and the current value iα_FBK converted to the αβ-axis fixed coordinate system by the three-phase / αβ conversion unit 70 is subtracted Is input to the subtractor 32, and is input to the subtractor 34.

The subtractor 32 outputs the current vector deviation Δi _α between the α-axis current command value iα_ref and the current value iα_FBK output from the inverter main circuit unit INV.
The subtractor 34 outputs the current vector deviation Δi _β between the β-axis current command value iβ_ref and the current value iβ_FBK output from the inverter main circuit unit INV.

The current vector deviation Δiα and the current vector deviation Δiβ output from the subtractors 32 and 34 are input to the angle calculation unit 36. The angle calculator 36 calculates an angle θi of the current vector deviation of the αβ axis (fixed coordinate system) from the input current vector deviations Δiα and Δiβ. The angle θi is obtained by the arctangent (tan ⁻¹ ) of the current vector deviations Δiα and Δiβ.

  Gate command generation unit 38 applies switching elements of U-phase, V-phase, and W-phase of inverter main circuit unit INV such that the current command value and the current value actually output from inverter main circuit unit INV coincide with each other. Output gate command.

  In this embodiment, there are eight combinations of switching states of six (two each phase) switching elements (not shown) of the inverter main circuit unit INV. Therefore, each phase of the output voltage of the inverter main circuit unit INV is different. The eight voltage vectors corresponding to the respective switching states are assumed in consideration of the phase difference of. The eight voltage vectors can be expressed as six basic voltage vectors V1 to V6 that are different in phase and equal in size by π / 3, and two zero voltage vectors V0 and V7.

  Here, eight voltage vectors V0 to V7 correspond to eight switching states. For example, when the switching element on the positive side of each phase is on, it is represented as "1", and the switching element on the negative side of each phase Is represented as "0" when is on. For example, when the U-phase positive switching element is on, the V-phase negative switching element is on, and the W-phase positive switching element is on, the voltage vector is expressed as (101).

  In the present embodiment, non-zero voltage vectors (voltage vectors V1 to V6 other than zero voltage vectors V0 = (000) and V7 = (111)) are based on the current command value and the angle θi of the current vector deviation of the detected current. Will be described as an example of current tracking PWM control that generates a gate command by selecting. The voltage vector V1 corresponds to (001) when it is represented by a gate command of UVW. Similarly, voltage vectors V2 to V7 and V0 are (010), (011), (100), (101), (110), (111) and (000). Among these, the voltage vector V0 and the voltage vector V7 are referred to as a zero voltage vector because the inter phase voltage of UVW is 0 V, and the voltage vectors V2 to V6 are referred to as non-zero voltage vectors. When the inverter main circuit unit INV outputs the zero voltage vector V0 or the zero voltage vector V7, the current changes only by the induced voltage of the rotor, and the amount of change decreases. Therefore, in the present embodiment, only non-zero voltage vectors are selected as voltage vectors in order to increase the current differential term when detecting the rotor position.

  The gate command generation unit 38 includes a table 38A storing gate commands of U phase, V phase and W phase with respect to the range of the angle θi, and a three phase / αβ conversion unit 38B.

  The table 38A sets a voltage vector corresponding to −π / 6 <angle θi ≦ π / 6 to V4 = (100), and a voltage vector corresponding to π / 6 <angle θi ≦ π / 2 to V6 = (110). The voltage vector corresponding to π / 2 <angle θi ≦ 5π / 6 is V2 = (010), and the voltage vector corresponding to 5π / 6 <angle θi ≦ 7π / 6 is V3 = (011), 7π / 6 Store voltage command corresponding to angle θi ≦ 3π / 2 as V1 = (001), voltage vector corresponding to 3π / 6 <angle θi ≦ 11π / 6 as V5 = (101), and store gate command of UVW phase doing.

  The gate command generation unit 38 selects a voltage vector closest to the vector of the angle θi with reference to the voltage vector V4 using the table 38A, and outputs a gate command corresponding to the selected voltage vector.

  The 3-phase / αβ conversion unit 38B receives the gate command output from the table 38A, performs αβ conversion on the gate command corresponding to the UVW phase, and calculates output voltage target vectors Vα and Vβ of the αβ axis fixed coordinate system. Output. The output voltage target vectors Vα and Vβ are αβ converted three-phase AC voltage commands that can be calculated from the gate command of the inverter main circuit unit INV, and the output voltage of the inverter main circuit unit INV that the gate command is to realize It is a vector value.

  The inverter main circuit unit INV includes a DC power supply (DC load) (not shown) and two switching elements of U-phase, V-phase, and W-phase. The two switching elements in each phase are connected in series between the DC line connected to the positive electrode of the DC power supply and the DC line connected to the negative electrode of the DC power supply. The switching elements of the inverter main circuit INV are controlled by the gate command received from the gate command generator 38. The inverter main circuit unit INV is a three-phase AC inverter that outputs the U-phase current Iu, the V-phase current Iv, and the W-phase current Iw to the synchronous machine M which is an AC load. The inverter main circuit unit INV can also charge the power generated by the synchronous machine M to a secondary battery which is a DC power supply.

FIG. 3 is a diagram for explaining a configuration example of a part of the synchronous machine shown in FIG.
Here, only a part of the synchronous machine M is shown, and the stator 100 and the rotor 200 of the synchronous machine M are, for example, a combination of a plurality of configurations shown in FIG. 3.

The synchronous machine M is a permanent magnet synchronous motor having magnetic saliency and is a synchronous machine having different inductance characteristics between the N direction of the magnet and the S direction of the magnet.
The synchronous machine M is, for example, a magnet type synchronous motor provided with a stator 100 and a rotor 200. The rotor 200 has a plurality of air gaps 4, a plurality of outer peripheral bridges BR1, a plurality of center bridges BR2, and a plurality of magnets 3.

  The center bridge BR2 is disposed along a line extending between the air gap 4 and connecting the center of the rotor 200 and the outer periphery. The line on which the center bridge BR2 is disposed is the d axis. The outer circumferential bridge BR1 is located between the outer circumference of the rotor 200 and the air gap 4. In the portion of the synchronous machine M shown in FIG. 3, six air gaps 4 extending from the outer peripheral portion of the rotor 200 toward the central portion are provided. Three air gaps 4 and three air gaps 4 are disposed in line symmetry with respect to the d-axis, and each of the plurality of air gaps 4 extends between the center bridge BR2 and the outer circumferential bridge BR1.

  Each of the plurality of magnets 3 is disposed at an end of each air gap 4 on the center bridge BR 2 side for stabilizing the magnetic path. The plurality of magnets 3 are arranged in such a direction that the outer peripheral side of the rotor 200 is an N pole and the central portion side is an S pole.

The estimated value calculation unit 40 calculates the output voltage target vectors Vα and Vβ when the current synchronized with the rotor frequency is supplied to the synchronous machine M in the initial estimation when the inverter main circuit unit INV starts from the stop state. Using the current values detected by the detection units SU and SW, with the winding resistance R of the synchronous machine M, the d-axis average inductance and the q-axis differential inductance as set values, the error of the estimated rotational phase angle of the synchronous machine M calculates the [Delta] [theta], the error [Delta] [theta] is computed the rotational phase angle estimate theta _est and the rotational speed estimation value omega _est of the synchronous machine M such that the zero. The current synchronized with the rotor frequency is, for example, a current that flows in a direction in which the inductance of the synchronous machine M is minimum.

The estimated value calculator 40 includes an αβ / dq converter 42, an error calculator 44, a PLL (Phase Locked Loop) calculator 46, a low pass filter 48, and an integral calculator 49.
The αβ / dq conversion unit 42 receives the rotational phase angle estimated value θest from the adder 60, receives the output voltage target vectors Vα and Vβ of the αβ axis fixed coordinate system from the gate command generation unit 38, and performs three-phase / αβ conversion The unit 70 receives the current values iα_FBK and iβ_FBK of the αβ axis fixed coordinate system from the unit 70, converts these vector values into the dq axis coordinate system, and outputs them. The values output from the αβ / dq conversion unit 42 are voltage vectors Vdc and Vqc in the dcqc coordinate system including the estimation error Δθ, and current vectors idc and iqc.

The error calculation unit 44 receives the voltage vectors Vdc and Vqc and the current vectors idc and iqc from the αβ / dq conversion unit 42, and calculates an estimation error Δθ based on them.
Below, an example of operation | movement of the difference | error calculating part 44 is demonstrated.

The voltage equation of the magnet type synchronous machine is expressed by [Equation 1].
Here, Vd and Vq are dq axis voltages, Id and Iq are dq axis currents, R is a winding resistance, ωe is an electrical angular velocity, Ld and Lq are dq axis inductances, Φ Is the magnet flux and p is the differential operator (= d / dt).

Furthermore, focusing on the difference between the inductance for the fundamental current and the behavior for the inductance for the harmonic current, the motor inductance is defined as follows.
That is, the dq-axis inductance relative to the fundamental wave current d-axis average inductance _{L da,} and q-axis average inductance _{L qa,} the dq-axis inductance relative to harmonic current, d-axis differential inductance _{L dh,} and q-axis differential inductance _{L qh.}

Based on this, when [Formula 1] is rewritten, the following [Formula 2] is obtained.
Here, L _da and L _qa are dq axis average inductances, and L _dh and L _qh are dq axis differential inductances.

Furthermore, to estimate the rotational phase angle θ, the expression of the induced voltage in the q-axis direction in which the magnet voltage is generated can be rewritten as Expression 3 below.

_Ex0 of the above [Formula 3] is expressed by the following [Formula 4], and in the following description, it is referred to as an extended induced voltage in the q-axis direction.

Furthermore, when the rotational phase angle θ causes an error Δθ with respect to the dq axis, the above [Equation 3] can be transformed into the following [Equation 5].

Furthermore, when the above [Equation 5] is modified to calculate the rotational phase angle error Δθ, the following [Equation 6] is obtained.

Further, when the dc axis component of the above [Equation 6] is divided by the qc axis component and the arc tangent is taken, the rotational phase angle error Δθ becomes the following [Equation 7].

According to the above, the voltages V _dc and V _qc and the detection currents I _dc and I _qc are acquired in the error calculation unit 44, and the winding resistance value R, the q axis average inductance L _qa and the d axis differential inductance L are obtained as parameters. By appropriately setting _dh , it is possible to calculate the rotational phase angle error Δθ in the above [Equation 7].

  However, as shown in FIG. 3, in the synchronous machine M in which a small amount of magnet is embedded in the rotor 200 (magnetic torque is not used for driving), the positive and negative d-axis currents are caused by the influence of magnetic saturation between the outer peripheral bridge BR1 and the center bridge BR2. The d-axis differential inductance characteristic greatly differs depending on whether it is the current in the −d-axis direction or the current in the + d-axis direction.

FIG. 4 is a diagram for explaining an example of the relationship between the d-axis current and the d-axis differential inductance of the permanent magnet synchronous motor having the magnetic saliency.
For example, assuming that the d-axis differential inductance L _dh when a current of 10% of the rating of the synchronous machine M is applied in the −d-axis direction is a set value in the above [Equation 7], the estimated rotation phase angle _θest is NS reversal When it converges to the specified angle, the current is actually conducted in the + d-axis direction, and the actual d-axis differential inductance (actual value) is about 52% of the set d-axis differential inductance (setting value). It breaks away. At this time, there is a possibility that the calculation of the rotational phase angle becomes unstable at the timing when current ripple occurs, for example, and the initial position estimation can not be performed.

Therefore, in the inverter control device and the sensorless drive system of the present embodiment, the error calculation unit 44 does not calculate the rotational phase angle error Δθ using the above-described extended induced voltage _Ex0 in the q-axis direction, but does not calculate the d-axis direction. The rotational phase angle error Δθ is calculated using the extended induced voltage.
Hereinafter, an example of an operation in which the error calculation unit 44 calculates the rotational phase angle error Δθ using the extended induction voltage in the d-axis direction will be described.

If the above [Formula 2] is transformed into the expression by the extended induction voltage in the d axis direction, it becomes [Formula 8].

At this time, the control device of the inverter main circuit INV that drives a synchronous machine in which the magnet magnetic flux が is sufficiently small compared to the voltage drop, that is, a small amount of magnet is embedded in the rotor 200 (magnetic torque is not used for driving). Then, the above [Equation 8] can be expressed as the following [Equation 9].

Here, E _x1 above [Equation 9] can be expressed as follows [Formula 10]. In the following description, E _x1 will be referred to as an extended induced voltage in the d-axis direction.

Furthermore, when the rotational phase angle error Δθ is generated with respect to the dq axis, the above [Equation 9] can be modified as the following [Equation 11].

Further, when the above [Equation 11] is modified in order to calculate the rotational phase angle error Δθ, the following [Equation 12] is obtained.

Furthermore, when the dc axis component of the above [Equation 12] is divided by the qc axis component and the arctangent is taken, the rotational phase angle error Δθ becomes the following [Equation 13].

According to the above, in the error calculating unit 44, dc- and qc-axis voltage _{V dc, V qc, dcqc} axis current _{I dc,} acquires the _{I qc,} as parameters, the winding resistance value R, d-axis average inductance _{L qa,} q-axis By appropriately setting the differential inductance L _qh , it is possible to calculate the rotational phase angle error Δθ in the above [Equation 13].

FIG. 5 is a diagram for explaining an example of the relationship between the d-axis current and the q-axis differential inductance of a permanent magnet synchronous motor having magnetic saliency.
For example, _assuming that the q-axis differential inductance L _qh when a current of 10% of the rating of the synchronous machine M is applied in the −d-axis direction is a set value in the above [Equation 12], the estimated rotation phase angle θ is NS inverted When the angle converges, in actuality, the current flows in the + d axis direction. Even in this case, the actual q-axis differential inductance (actual value) with respect to the set q-axis differential inductance (setting value) only deviates by about 8%, and the desensitization against the parameter setting error is realized. be able to.

The PLL operation unit 46 performs the convergence control (PLL control) on the rotational phase angle error Δθ output from the error operation unit 44 using proportional / integral operation (PI control) to estimate the first rotation speed. Calculate and output the value ω _est '.
Low pass filter 48 outputs the rotational angular velocity estimate omega _est except the high frequency component of the first rotating angular velocity estimate omega _{est 'output} from the PLL operation unit 46.

Integral calculation unit 49 calculates the integral of the rotational angular velocity estimate omega _est outputted from the low-pass filter 48 the rotational phase angle estimate theta _{est ',} and outputs. The rotational phase angle estimated value θ _est 'output from the integral calculation unit 49 is the output of the estimated value calculation unit 40 and is input to the adder 60.

In the initial estimation when starting up the inverter main circuit unit INV, for example, when the current synchronized with the rotor frequency of the synchronous machine M is supplied, the polarity determination unit 50 generates a magnetic flux or voltage synchronized with the rotor frequency generated. The magnetic pole discrimination is performed using both or both of them, and the correction value θ _NS of the estimated value θ _est 'of the rotational phase angle based on the discrimination result is output. The polarity determination unit 50 can determine the magnet magnetic pole using, for example, a d-axis fundamental wave magnetic flux generated when a current in the d-axis direction is applied or a q-axis voltage generated by the fundamental wave magnetic flux.

  In the present embodiment, in the synchronous machine M, a difference occurs in the magnitude of the d-axis linkage flux between the case where the current is supplied to the + d axis and the case where the current is supplied to the -d axis. Therefore, the polarity determination unit 50 determines the magnet polarity of the synchronous machine M based on the difference in the d-axis linkage flux. The difference in the d-axis linkage flux occurs not only in a synchronous machine with a small amount of magnet but also in a synchronous machine with a large amount of magnet.

The polarity determination unit 50 _calculates the q-axis voltage set value _{Vd_FF} with [Equation 14], and the voltage difference ΔV _{q —} NS as a reference for the NS determination with [Equation 15]. The q-axis voltage actual value V _qc can be expressed as [Equation 16].

The d-axis inductance set value _{Ld_FF} may be a value between the d-axis inductance when current is supplied in the + d-axis direction and the d-axis inductance when current is supplied in the -d-axis direction. In the present embodiment, the d-axis inductance setting value _{Ld_FF} is, for example, an average value of the d-axis inductance when current is supplied to the + d axis and the d-axis inductance when current is supplied to the -d axis.

Set the d-axis inductance setting value _{L D_FF,} if the current control is performed accurately, the current command value _{i d_ref} is equal to the d-axis current actual value _{i dc,} the voltage difference [Delta] V _{Q_NS} relationship [Equation 17] It becomes.

The polarity determination unit 50 outputs the correction value θ _NS of the rotational phase and the rotational phase angle estimated by the speed estimation means in accordance with the relationship of the above [Equation 17].
The polarity determination unit 50 receives the q-axis voltage V _qc , the d-axis current command _{id_ref,} and the estimated rotational speed value ω _est . Polarity determination unit 50 calculates the voltage difference [Delta] V _{Q_NS} by the [formula 17], the voltage difference delta _{Vq_NS} determines whether or zero, and outputs the correction value theta _NS according to the determination result. That is, the polarity determination unit 50 sets the correction value θNS to 0 ° when the voltage difference ΔV _{q_NS} is greater than or equal to zero, and _{sets the} correction value θ _NS to π (180 °) when the voltage difference ΔV _{q_NS} is less than zero. The correction value θ _NS output from the polarity determination unit 50 is input to the adder 60.

The adder 60 adds the rotational phase angle estimated value _θest output from the estimated value calculation unit 40 and the correction value θ _NS output from the polarity determination unit 50, and uses the correction value θ _NS to estimate the rotational phase angle _θest is corrected. The rotational phase angle estimated value _θest after correction is supplied to the dq / αβ conversion unit 20 and the αβ / dq conversion unit 62, and is used for vector conversion.

FIG. 6 is a diagram for explaining an example of the operation of the inverter control device of the embodiment.
Here, a coasting restart operation will be described as an example in which the rotational phase angle and the rotational speed are estimated and restarted from the state where the inverter control device 1 is stopped.

  In the inverter control device 1 of the present embodiment, the polarity determination is performed in the initial estimation at the time of startup. That is, the calculation of the rotational phase angle estimated value by the estimated value calculation unit 40 and the magnet magnetic pole determination by the polarity determination unit 50 are executed according to the start command of the inverter main circuit unit INV. The inverter main circuit unit INV is in a stopped state, and the synchronous machine M is in a free run until it is initialized before startup and after initialization is completed.

The command generation unit CTR sets the current commands _{id_ref} and _{iq_ref} for energizing the motor and the current phase _{β_ref} , and sets various flags (current energization flag (Ion), initialization flag, initial estimation flag, NS discrimination calculation flag, The NS determination result reflection flag (normal control flag) is controlled. The command generation unit CTR supplies the initialization value calculation unit 40 with the initialization flag, the initial estimation flag, the normal control flag, and the NS determination calculation flag. The command generation unit CTR supplies the NS determination result reflection flag to the polarity determination unit 50. The command generation unit CTR supplies a current conduction flag (Ion) to the current command generation unit 10.

  When the command generation unit CTR receives the start command, it simultaneously raises the initialization flag. Subsequently, the command generation unit CTR raises the initial estimation flag and the current conduction flag (Ion), and lowers the initialization flag.

When the initialization flag rises, the estimated value calculation unit 40 sets an estimated value of the rotational phase angle and the rotational speed to an initial value and initializes it. Subsequently, when the initial estimation flag rises, calculation of the rotational phase angle estimated value _θest and the rotational speed estimated value _ωest is started.

Subsequently, the command generation unit CTR raises the NS determination calculation flag.
The polarity determination unit 50 calculates the voltage difference ΔV _{q —} NS when the NS determination calculation flag rises.

Subsequently, the command generation unit CTR lowers the initial estimation flag and the NS discrimination calculation flag, and raises the NS discrimination result reflection flag.
Polarity determination unit 50, when NS determination result reflection flag rises, and outputs the correction value theta _NS rotational phase angle according to the value of the voltage difference [Delta] V _{Q_NS} as shown in [Equation 17].

Subsequently, the command generation unit CTR lowers the NS determination result reflection flag and raises the initialization flag.
When the initialization flag rises, the estimated value calculation unit 40 sets the estimated values θ _est and ω _est of the rotational phase angle and the rotational speed to initial values and initializes them.

Subsequently, the command generation unit CTR lowers the initialization flag and raises the normal control flag. When the normal control flag rises, the estimated value calculation unit 40 ends the initial estimation process, and starts an operation of power running drive or regenerative drive.
Then, an example of a result of having performed simulation about inverter control device 1 of the above-mentioned embodiment is explained.

FIGS. 7 and 8 are diagrams showing an example of simulation results when the rotational phase angle estimated value is calculated using the extended induced voltage in the d-axis direction.
In the example shown in FIGS. 7 and 8, the d-axis differential inductance L _dh when a current of 10% of the rating of the synchronous machine M is applied in the −d-axis direction is set as the set value in the above [Equation 13]. Shows d-axis current and q-axis current passing through machine M and their estimated values dc-axis current and qc-axis current, and using the actual rotation phase angle θ and extended induced voltage in the d-axis direction in the middle stage The calculated rotational phase angle estimated value _θest is shown, and the output torque of the synchronous machine M is shown in the lower part.

  7 and 8 show simulation results of a period including a period from the timing when the current conduction flag Ion shown in FIG. 6 rises to the timing when the NS determination calculation flag rises. For example, in this simulation, the inverter controller starts current conduction at 0 seconds, starts calculation of estimated values of rotational phase angle and rotational speed, and starts calculation of polarity determination at 0.1 seconds. doing.

  The example shown in FIG. 7 shows the case where the actual value of the rotational phase angle and the estimated value do not deviate. The example shown in FIG. 8 shows the case where the actual value of the rotational phase angle and the estimated value deviate by 180 °. According to this simulation result, the calculation of the rotational phase angle estimated value was stably performed in both cases of FIG. 7 and FIG.

Then, the result of having performed simulation about the comparative example of the inverter control apparatus 1 of the above-mentioned embodiment is demonstrated.
FIG. 9 and FIG. 10 are diagrams showing an example of simulation results when the rotational phase angle estimated value is calculated using the extended induced voltage in the q-axis direction.

In the example shown in FIGS. 9 and 10, the d-axis differential inductance L _dh when a current of 10% of the rating of the synchronous machine M is supplied in the −d-axis direction is set as the set value in the above [Equation 7]. Shows d-axis current and q-axis current passing through machine M and their estimated values dc-axis current and qc-axis current, and using the actual rotation phase angle θ and extended induced voltage in the d-axis direction in the middle stage The calculated rotational phase angle estimated value _θest is shown, and the output torque of the synchronous machine M is shown in the lower part.

  FIGS. 9 and 10 show simulation results of a period including a period from the timing when the current conduction flag Ion shown in FIG. 6 rises to the timing when the NS determination calculation flag rises. For example, in this simulation, the inverter controller starts current conduction at 0 seconds, starts calculation of estimated values of rotational phase angle and rotational speed, and starts calculation of polarity determination at 0.1 seconds. doing.

  The example shown in FIG. 9 shows the case where the actual value of the rotational phase angle and the estimated value do not deviate. The example shown in FIG. 10 shows the case where the actual value of the rotational phase angle and the estimated value deviate by 180 °.

In this example, when the rotational phase angle and the actual value deviate by 180 °, that is, the rotational phase angle estimated value _θest converges to the NS inverted angle, and the d-axis differential inductance set by the error calculation unit 44 (set value ) Is different from the actual d-axis differential inductance (actual value). According to the result shown in FIG. 10, calculation of the rotational phase angle estimated value is destabilized at the timing when the current ripple occurs, etc., and then the rotational phase angle estimated value converges to a state deviated from the actual value by 180 °. It was not.

  As described above, in the inverter control device and the sensorless drive system according to the present embodiment, current tracking PWM control is used as a current control method, and the inverter main device is selected according to the phase of the difference vector between the current command value and the current detection value. A gate command to the circuit unit INV is generated. The gate command is a non-zero voltage vector, and the harmonic current can be increased by applying the maximum voltage that the inverter main circuit INV can output to the synchronous machine M.

At this time, if rotational phase angle estimation based on the extended induction voltage in the d-axis direction is performed as described above, the harmonic current can be increased while performing current control, so low speed including the state where synchronous machine M is stopped For the state of operating at the maximum speed, it is possible to calculate the estimated value of the rotational phase angle with one equation ([Equation 13]) to control the inverter main circuit INV.
That is, according to the present embodiment, it is possible to provide an inverter control device and a sensorless drive system that perform current control with high accuracy.

Further, in the above embodiment, the drive system adopting the current tracking type PWM control which determines the gate command of the inverter main circuit unit INV according to the current vector deviation (Δi _α , Δi _β ) has been described. Even when other methods are adopted, the same effect can be obtained. For example, the same effect can be obtained by a method such as PI control in which the inverter voltage command is calculated based on the current vector deviation (Δi _α , Δi _β ).

In the above embodiment, the gate command of the inverter main circuit INV is directly determined from the angle θi of the current vector deviation (Δi _α , Δi _β ). However, the gate command for controlling the current is determined. If possible, the same effect as the above embodiment can be obtained even if a method such as triangular wave comparison modulation or space vector modulation is adopted.

In the above embodiment, the method of calculating the rotational phase angle error Δθ on the rotational coordinate system and calculating the rotational phase angle estimated value _θest and the rotational speed estimated value _ωest from the information has been described as an example. Similar effects can be obtained even with the method of directly calculating the rotational phase angle as disclosed in Patent Document 1. When the rotational phase angle is directly calculated, the rotational phase angle can be directly calculated without PLL control and integration of the rotational phase angle error Δθ. Also in this case, since the motor angle is used to calculate the phase angle, the same effect as the above embodiment can be obtained.

Next, an inverter control device and a sensorless drive system according to a second embodiment will be described below with reference to the drawings. In the following description, the same components as those in the first embodiment described above are denoted by the same reference numerals and description thereof is omitted.
FIG. 11 is a block diagram schematically showing one configuration example of the inverter control device and the sensorless drive system of the second embodiment.

The inverter control device of the present embodiment further includes an estimation method switching unit 80.
The estimation method switching unit 80 outputs an estimation method switching signal to the error calculation unit 44. The estimation method switching unit 80 reads a switch signal such as a dip switch provided outside the inverter control device or the sensorless drive system, or the value of the estimation method switching signal according to the value of the d-axis differential inductance L _dh. Switch to “1” or “0” and output.

  The estimation method switching signal is, for example, that a synchronous machine whose magnet voltage is sufficiently small compared to the voltage drop at rated operation is electrically connected to the inverter main circuit unit INV, that is, by the output of the inverter main circuit unit INV. If driven, it is '0', otherwise it is '1'.

In addition, the estimation method switching unit 80 may perform auto tuning based on the d-axis differential inductance L _dh . The estimation method switching unit 80 may obtain the d-axis differential inductance L _dh from the outside in order to determine the estimation method switching signal, and calculates the d-axis differential inductance L _dh by the voltage equation of the above [Equation 2] May be When the d-axis differential inductance L _dh is calculated by Equation 2, the estimation method switching unit 80 determines the voltages V _d and V _q , the detection currents I _d and I _q , the winding resistance R, the electrical angular velocity ωe, The d axis differential inductance L _dh can be calculated by acquiring the dq axis average inductances L _da and L _qa and the magnet magnetic flux Φ.

For example, the estimation method switching unit 80 calculates the ratio of the respective d-axis differential inductances L _dh when the current is supplied in the −d-axis direction and the + d-axis direction (the actual value of the d-axis differential inductance with respect to the setting value of the d-axis differential inductance When the ratio of the d-axis differential inductance L _dh is less than 30%, the estimation method switching signal is “1 (second) when the ratio of the d-axis differential inductance L _dh is less than 30%. Level).
For example, the d-axis differential inductance L _dh when a current of 10% of the rating of the synchronous machine M is applied in the −d-axis direction is the set value in the above [Equation 7], and the rotation phase angle estimated value _θest is NS inverted When the angle converges, in actuality, the current flows in the + d axis direction. At this time, when the actual d-axis differential inductance (actual value) is apart from the set d-axis differential inductance (set value) by 30% or more, the estimation method switching unit 80 outputs “0 If the actual d-axis differential inductance (actual value) is less than 30% with respect to the set d-axis differential inductance (setting value), the estimation method switching unit 80 sets the estimation method switching signal to "1". Do.

  The error calculating unit 44 switches the method of calculating the rotational phase angle error Δθ in accordance with the value of the estimation method switching signal. For example, when the value of the estimation method switching signal is “1”, the error calculation unit 44 calculates and outputs the rotational phase angle error Δθ by [Equation 7] (first method), and the value of the estimation method switching signal Is 0, the rotational phase angle error .DELTA..theta. Is calculated and output by [Equation 13] (second method).

  Note that the error calculation unit 44 is configured to calculate the rotational phase angle error Δθ by switching between another method not using Equation 7 and a method using Equation 13 according to the estimation method switching signal. It does not matter.

  Further, in the present embodiment, the method of calculating the rotational phase angle error Δθ using the equation 7 or the equation 13 has been described. However, another method or another one can be used as long as it can calculate the estimated rotational phase angle. A plurality of schemes may be further adopted to switch three or more operation schemes based on the value of the estimation scheme switching signal.

As another calculation method, for example, a method of calculating the rotational phase angle error Δθ as in the following [Expression 18] may be adopted using the d-axis component of [Expression 12].
For example, as described in Patent Document 1, a method may be employed in which the rotational phase angle estimated value is directly calculated in the αβ coordinate system (fixed coordinate system).

According to the inverter control device and the sensorless drive system of the present embodiment, the same effects as those of the first embodiment described above can be obtained. Furthermore, according to the inverter control device and the sensorless drive system of the present embodiment, the synchronous machine in which the d-axis differential inductance changes largely depending on the direction of the d-axis current (whether it is the −d-axis direction or the + d-axis direction) is The rotation phase angle estimated value and the rotation speed estimated value are calculated by the method using [], and the rotation phase angle is calculated by the method using [Expression 7] for a synchronous machine that can not expand the expression of [Expression 9] Estimates and rotational speed estimates can be computed. This makes it possible to stably calculate the rotational phase angle estimated value regardless of the characteristics of the synchronous machine.
That is, according to the present embodiment, it is possible to provide an inverter control device and a sensorless drive system that perform current control with high accuracy.

  In the first and second embodiments, each configuration of inverter control device 1 and sensorless drive system may be realized by hardware or software, and hardware and software may be used. It may be realized by a combination. The inverter control device 1 includes at least one processor such as a CPU or an MPU and a memory storing a program to be executed by the processor, and the processor executes the program for performing the operation of the inverter control device 1 described above. It may be configured.

  While certain embodiments of the present invention have been described, these embodiments have been presented by way of example only, and are not intended to limit the scope of the invention. These novel embodiments can be implemented in various other forms, and various omissions, substitutions, and modifications can be made without departing from the scope of the invention. These embodiments and modifications thereof are included in the scope and the gist of the invention, and are included in the invention described in the claims and the equivalent scope thereof.

DESCRIPTION OF SYMBOLS 1 ... Inverter control apparatus, 10 ... Current command generation part, 20 ... dq / (alpha) beta conversion part, 30 ... Current control part, 32 ... Subtractor, 34 ... Subtractor, 36 ... Angle operation part, 38 ... Gate command generation part, 38A: Table 38B: Three-phase / αβ conversion unit 40: Estimated value operation unit 42: αβ / dq conversion unit 44: Error operation unit 46: PLL operation unit 48: Low pass filter 49: Integral operation unit , 50: polarity determination unit, 60: addition unit, 62: αβ / dq conversion unit, 70: 3 phase / αβ conversion unit, 80: estimation method switching unit, M: synchronous machine, 100: stator, 200: rotor , 3 ... Magnets, 4 ... Air gaps, BR1 ... Outer peripheral bridges, BR2 ... Center bridges, V0 to V7 ... Voltage vectors, V0, V7 ... Zero voltage vectors, V1 to V6 ... Basic voltage vectors.

Claims (6)

A current command generation unit that generates a current command value;
A current detection unit that detects the current value of the alternating current output from the inverter main circuit unit;
A gate command of the inverter main circuit unit is generated so that the current command value matches the current value detected by the current detection unit, and an output voltage target vector of the inverter main circuit unit is calculated based on the gate command. A gate command generation unit to
An estimated value calculation unit for calculating a rotational phase angle estimated value based on the current value detected by the current detection unit and the output voltage target vector in initial estimation when the inverter main circuit unit starts up; Have
The estimated value calculation unit, when the inverter main circuit unit starts from a stop state, the output voltage target vector when a current synchronized with the rotor frequency is supplied, and a current value detected by the current detection unit Using the winding resistance, the d-axis average inductance, and the q-axis differential inductance as set values, the error of the estimated rotational phase angle is calculated, and the estimated rotational phase angle is calculated so that the error is zero. An inverter control device characterized by:   The inverter control device according to claim 1, wherein the current synchronized with the rotor frequency is a current which is energized in a direction in which an inductance is minimized. The estimated value calculator calculates a rotational angular velocity estimated value at which the error is zero,
A polarity determination unit that determines the rotor polarity based on the rotational angular velocity estimated value and outputs a correction value of the rotational phase angle estimated value;
An adder for adding the rotation phase angle estimated value and the correction value and outputting the sum;
The inverter control device according to claim 1 or 2, characterized in that: It further comprises an estimation method switching unit for outputting an estimation method switching signal to the estimated value calculation unit,
The estimated value calculation unit calculates the error using the winding resistance, the q-axis average inductance, and the d-axis differential inductance as set values according to the estimation method switching signal, and the winding The rotation phase angle estimated value is calculated by switching between the resistance, the d-axis average inductance, and the second method of calculating the error with the q-axis differential inductance as a set value. The inverter control device according to any one of items 1 to 3.   The estimation method switching unit is configured such that the ratio of the actual value of the d-axis differential inductance to the set value of the d-axis differential inductance when energizing in the −d-axis direction and the + d-axis direction is 30% or more The inverter control device according to claim 4, wherein the signal is a signal that is at a level, and is at a second level when the ratio is less than 30%. The inverter control device according to any one of claims 1 to 5.
Synchronous machine,
A sensorless drive system comprising: the inverter main circuit unit for driving the synchronous machine.