JP2021022965A - Driving device of induction motor, driving method, and electric vehicle - Google Patents

Abstract

To estimate a rotation direction and a speed of an induction motor being an AC motor in a short time regardless of a rotation speed while obviating the need for enhancing response of a control system.SOLUTION: A drive system 1S of an induction motor 1 comprises an inverter 3 for driving the induction motor 1, and a controller 4 for controlling the inverter 3. When the inverter 3 outputs a power to the induction motor 1 after it is restarted, the controller 4 switches a control mode to transfer to either one of an AC voltage application mode for applying an AC voltage to the induction motor 1 or a DC voltage application mode for applying a DC voltage, on the basis of a short-circuit current detection value detected while motor terminals of the induction motor 1 are short-circuited.SELECTED DRAWING: Figure 1

Description

The present invention relates to a drive device, a drive method, and an electric vehicle for an induction motor.

For example, Patent Document 1 discloses a method of estimating the rotation direction and speed of an AC motor in a free-run state from a voltage command for controlling the current flowing through the AC motor to zero by an induced voltage due to residual magnetic flux. If the amplitude of the voltage command is less than a predetermined threshold, the control to make the current zero is stopped, and by applying a DC voltage, the rotation direction and speed of the AC motor are changed from the frequency of the current flowing through the AC motor. presume.

Japanese Patent No. 4407151

However, in the above-mentioned conventional technique, since the speed is estimated by controlling the three-phase AC current flowing through the AC motor to zero, there is a problem that it is necessary to improve the response of the control system. Further, when estimating the speed from the frequency of the current flowing through the AC motor by applying a DC voltage, there is a problem that it takes a lot of time to estimate the speed in the low speed range of the AC motor.

An object of the present invention has been made in consideration of the above points, and the rotation direction and speed of the induction motor, which is an AC motor, can be shortened regardless of the rotation speed while eliminating the need for high response of the control system. One purpose is to be able to estimate in time.

In order to solve such a problem, in the present invention, as one means for achieving one object, the drive device of the induction motor includes a power conversion device for driving the induction motor and a control device for controlling the power conversion device. Then, when the control device outputs power to the induction motor after restarting the power conversion device, the control device ACs to the induction motor based on a short-circuit current detection value detected by short-circuiting the motor terminals of the induction motor. It is characterized in that it switches between the control mode of applying an AC voltage to which a voltage is applied and the mode of applying a DC voltage to which a DC voltage is applied.

According to the present invention, for example, the rotation direction and speed of the induction motor can be estimated in a short time regardless of the rotation speed while eliminating the need for increasing the response of the control system.

The functional block diagram which shows the whole structure example of the drive system of Embodiment 1. FIG. The flowchart which shows the speed estimation processing example of the induction motor of Embodiment 1. FIG. 6 is a functional block diagram showing a configuration example of a DC voltage calculation unit according to the first embodiment. FIG. 6 is a functional block diagram showing a configuration example of an AC voltage calculation unit according to the first embodiment. FIG. 6 is a functional block diagram showing a configuration example of a short-circuit current frequency calculation unit according to the first embodiment. FIG. 5 is a flowchart showing an example of AC voltage calculation processing by a controller including an AC voltage calculation unit of the first embodiment. The figure which shows an example of the speed estimation waveform when DC voltage application is selected after the amplitude determination of a short circuit current. The figure which shows an example of the speed estimation waveform at the time of selecting the AC voltage application after the amplitude determination of a short circuit current. The functional block diagram which shows the structural example of the AC voltage calculation part of Embodiment 2. Vector diagram at short circuit. The vector diagram after the phase adjustment of Embodiment 2. The functional block diagram which shows the structural example of the AC voltage calculation part of Embodiment 3. The vector diagram after the phase adjustment of Embodiment 3. The figure which shows an example of the velocity estimation waveform of Embodiment 3. The functional block diagram which shows the configuration example of the drive system of Embodiment 4. FIG. 5 is a functional block diagram showing a configuration example of an AC voltage calculation unit according to the fourth embodiment. The schematic diagram which shows an example of the electric vehicle of Embodiment 5.

Hereinafter, embodiments of the present invention will be described with reference to the drawings. In the following description, the same or similar elements and processes are, in principle, given the same reference numerals. In addition, duplicate explanations will be omitted for the same functions and processes. It should be noted that the configuration and processing described below are merely examples, and it is not intended that the embodiment according to the present invention is limited to the following specific modes. In addition, each embodiment and modification can be combined in part or in whole within the scope of the technical idea of the present invention and within the range consistent with it.

Prior to the description of the following embodiments, the prior art and its problems will be described. In an electric vehicle, after accelerating by starting the inverter from a stopped state, the inverter may be stopped to move to a coasting state in which the vehicle coasts, and the inverter may be started again from the coasting state to accelerate or decelerate.

Since the inverter is stopped in the coasting state, no voltage is applied from the inverter to the induction motor, but in principle, the magnetic flux does not disappear immediately even if the inverter is stopped. The magnetic flux remaining inside the rotor when the inverter is stopped in this way is called the residual magnetic flux. When the residual magnetic flux is present, an induced voltage proportional to the product of the residual magnetic flux and the rotation speed is generated.

When the inverter is stopped, the energy of the residual magnetic flux is consumed by the resistance of the induction motor, and is attenuated according to the time constant determined by the resistance value and the inductance value of the induction motor.

On the other hand, in recent years, due to the growing awareness of energy conservation, induction motors are designed with low slip, so resistance is small. Therefore, the time constant at which the residual magnetic flux is attenuated tends to be long. As a result, when the inverter is restarted from the stopped state or the coasting state, the residual magnetic flux tends to remain without disappearing, and the induced voltage due to the residual magnetic flux becomes a disturbance, and there is a concern that the speed estimation fails.

From the above, it is desirable that the speed of the induction motor can be estimated regardless of the amount of residual magnetic flux. A speed estimation method considering the residual magnetic flux has been proposed as in Patent Document 1 described above, but it is necessary to improve the response of the control system, and it takes a lot of time to estimate the speed in the low speed range of the AC motor. There is a problem such as doing. The following embodiments show an example of means for solving this problem.

[Embodiment 1]
First, the components of the first embodiment will be described. FIG. 1 is a functional block diagram showing an overall configuration example of the drive system 1S of the first embodiment. The drive system 1S of the present embodiment includes an induction motor 1, a phase current detection circuit 2, an inverter 3, and a controller 4.

The phase current detection circuit 2 is composed of a Hall CT (Current Transformer) or the like, and detects three-phase AC currents Iu, Iv, and Iw of U phase, V phase, and W phase. However, it is not always necessary for the phase current detection circuit 2 to detect the currents of all three phases, and one of the two phases is detected and the other one phase is calculated by assuming that the three-phase current is in an equilibrium state. It may be configured.

The inverter 3 is composed of a DC voltage power supply 5, a gate driver 6, and a plurality of switching elements of IGBT (Insulated Gate Bipolar Transistor) Q1 to Q6 and diodes D1 to D6. The effect of this embodiment is not limited to the type of semiconductor switching element.

The controller 4 uses an αβ coordinate conversion unit 7 that converts a three-phase AC current into an αβ coordinate system current, a dq coordinate conversion unit 8 that converts an αβ coordinate system current into a dq coordinate system current, and an αβ coordinate system current. Short-circuit current calculation unit 9 that calculates the current when the inverter is short-circuited, AC current command calculation unit 10 that outputs the AC current command, and AC voltage calculation unit 11 that calculates the AC voltage command from the AC current command and the current of the dq coordinate system. Has.

Further, the controller 4 selects an AC voltage command or a DC voltage command using a DC current command calculation unit 12 that outputs a DC current command, a DC voltage calculation unit 13 that calculates a DC voltage command from the DC current command, and a short-circuit current. It has a voltage mode switching unit 14 and a PWM signal generation unit 15 that generates a PWM signal from a voltage command output by the voltage mode switching unit 14 and outputs the PWM signal to the gate driver 6.

Next, the operation of the controller 4 in this embodiment will be described with reference to FIGS. 1 and 2.
FIG. 2 is a flowchart showing an example of speed estimation processing executed by the controller 4 of the induction motor 1 of the first embodiment.

First, in step S11, when the induction motor 1 is restarted from the stopped state or the coasting state, the controller 4 continues to short-circuit the motor terminals of the induction motor 1 for a predetermined short-circuit period from the start of the short-circuit. For example, this predetermined short-circuit period is a period equal to or greater than the primary conversion time constant Tσ obtained by dividing the leakage inductance Lσ of the induction motor 1 by the primary conversion resistance Rσ. At this time, the controller 4 outputs a signal for short-circuiting either the upper or lower arm of the inverter 3. During the short circuit, the three-phase AC voltage commands Vu ^* , Vv ^* , and Vw ^* , which are the outputs of the voltage mode switching unit 14, are zero.

Next, in step S12, the controller 4 determines whether or not the amplitude | I1 | of the short-circuit current I1 is less than the short-circuit current amplitude threshold Th when a predetermined short-circuit period elapses from the start of the short-circuit. At this time, the short-circuit current calculation unit 9 calculates the amplitude | I1 | of the short-circuit current I1 using the following equation (1).

The controller 4 shifts the process to step S13 (DC application mode) when the amplitude | I1 | of the short-circuit current I1 is less than the short-circuit current amplitude threshold Th (step S12YES), and when the amplitude | I1 | is greater than or equal to the short-circuit current amplitude threshold. The process shifts to step S14 (AC application mode) in (step S12NO). The voltage mode switching unit 14 switches between the DC voltage application mode and the AC voltage application mode.

In step S13, the controller 4 shifts to the DC voltage application mode, and applies the DC voltage to the induction motor 1 from the transition time to a predetermined DC voltage application period. For example, this DC voltage application period is the same period as the half cycle of the maximum rotor frequency at the time of retreat. The controller 4 shifts the process to step S14 after the DC voltage application period has elapsed.

In step S14, the controller 4 shifts to the AC voltage application mode, and estimates the speed of the induction motor 1 for a predetermined AC voltage application period (speed estimation period). For example, this speed estimation period is a period equal to or longer than the current control response time constant. The controller 4 ends the speed estimation when the speed estimation period elapses.

The details of each functional block will be described below.

The DC voltage calculation unit 13 will be described with reference to FIG. FIG. 3 is a functional block diagram showing a configuration example of the DC voltage calculation unit 13 of the first embodiment. The DC voltage calculation unit 13 includes an LPF (Low-Pass Filter) 16, a voltage command calculation unit 17, and a dq coordinate inverse conversion unit 18. The LPF 16 calculates the d-axis magnetic flux command Φ _2d ^* based on the d-axis current command Id ^* output from the DC current command calculation unit 12. The voltage command calculation unit 17 calculates the d-axis voltage command Vd ^* and the q-axis voltage command Vq ^* by the following equations (2) and (3).

However, in the above equations (2) and (3), R1 ^* is the primary resistance set value, Iq ^* is the q-axis current command, ω1 is the inverter frequency, Lσ ^* is the control side leakage inductance, and M ^* is the control side mutual inductance. , L2 ^* is the control side secondary inductance, and T2 ^* is the control side secondary time constant. Since the DC voltage calculation unit 13 applies a DC voltage in the d-axis direction, ω1 = 0 and Iq ^* = 0, so that the d-axis voltage command Vd ^* and the q-axis voltage command Vq output from the voltage command calculation unit 17 ^* Is as shown in the following equations (4) and (5).

The dq coordinate inverse conversion unit 18 converts the d-axis voltage command Vd ^* and the q-axis voltage command Vq ^* output from the voltage command calculation unit 17 into three-phase coordinates to convert the U-phase voltage command Vu ^* and the V-phase voltage command Vv. ^* , W-phase voltage command Vw ^* is generated and output.

The AC voltage calculation unit 11 will be described with reference to FIG. FIG. 4 is a functional block diagram showing a configuration example of the AC voltage calculation unit 11 of the first embodiment. The AC voltage calculation unit 11 includes an LPF 16b, a voltage command calculation unit 17b, a dq coordinate inverse conversion unit 18b, a speed estimation unit 19, a speed estimation initial value switching unit 20, a speed estimation initial value calculation unit 21, and a short-circuit current frequency calculation unit 22. Has.

Hereinafter, the speed estimation unit 19, the speed estimation initial value switching unit 20, the speed estimation initial value calculation unit 21, and the short-circuit current frequency calculation unit 22 will be described in detail. Since the LPF 16b, the voltage command calculation unit 17b, and the dq coordinate inverse conversion unit 18b have the same configuration as the LPF 16, the voltage command calculation unit 17, and the dq coordinate inverse conversion unit 18, respectively, the description thereof will be omitted.

The speed estimation unit 19 outputs the speed estimation values calculated by the following equations (6) and (7). From the voltage equation of the induction motor, the q-axis current detection value Iq and the q-axis current command value Iq ^* are expressed by the following equations.

However, in the above equations (6) and (7), ωr is the rotor velocity, ωr ^ is the velocity estimate, Rσ ^* is the linear conversion resistance set value, and s is the Laplace operator. Here, assuming that the constant of the induction motor 1 and the constant of the controller 4 match, the d-axis current detection value Id, the d-axis magnetic flux Φ _2d , and the q-axis magnetic flux command Φ 2q ^* each match each command value. Assuming that there is, the following equation (8) is obtained.

From the above equation (8), if Iq ^* −Iq = 0, ωr−ωr ^ = 0⇔ωr ^ = ωr, and the velocity is estimated when the q-axis current detection value Iq converges to the q-axis current command value Iq ^*. Complete. However, in the above equation (8), ωacr is the current control response angular frequency.

The voltage command calculation unit 17b substitutes the velocity estimation value ωr ^ obtained by the above equation (8) into ω1 of the above equations (2) and (3), and substitutes the d-axis voltage command Vd ^* and the q-axis voltage command Vq ^*. To calculate.

The speed estimation initial value calculation unit 21 operates when the amplitude | I1 | of the short-circuit current I1 is less than the short-circuit current amplitude threshold Th, and the DC voltage calculation unit 13 applies a DC voltage to the induction motor 1. If the q-axis current detection value Iq> 0, the speed estimation initial value calculation unit 21 determines that the rotation direction of the induction motor 1 is backward, and if Iq ≦ 0, it is forward.

Hereinafter, the principle that the rotation direction can be determined from the polarity of the q-axis current will be described. The relationship between the q-axis current and the rotation direction is as follows.

However, in the above equation (9), Id is the d-axis current detection value, M is the mutual inductance, L2 is the secondary inductance, and T2 is the secondary time constant. The DC voltage calculation unit 13 applies a voltage in the positive direction of the d-axis. Therefore, since the d-axis current detection value Id is positive, the signs of ωr and Iq are opposite. Therefore, the rotation direction of the induction motor 1 can be determined from the positive and negative of the q-axis current detection value Iq.

The speed estimation initial value calculation unit 21 outputs the initial value ω _{ini_Iq} of the speed estimation according to the determined rotation direction of the induction motor 1. For example, the initial value ω _{ini_Iq} of speed estimation is the maximum speed or half of the maximum speed in the forward direction when it is determined to be forward, and the maximum speed or maximum speed in the backward direction when it is determined to be backward. Half the speed is set. Further, in the speed estimation initial value calculation unit 21, the timing at which the voltage mode switching unit 14 selects the U-phase voltage command Vu ^* , the V-phase voltage command Vv ^* , and the W-phase voltage command Vw ^* output by the AC voltage calculation unit 11. The initial value ω _{ini_Iq} is output at (hereinafter referred to as AC voltage application start timing), and 0 is output at other timings.

The short-circuit current frequency calculation unit 22 will be described with reference to FIG. FIG. 5 is a functional block diagram showing a configuration example of the short-circuit current frequency calculation unit 22 of the first embodiment. The short-circuit current frequency calculation unit 22 includes an inverse tangent calculation unit 24, a PI controller 25, and an integrator 26. The inverse tangent calculation unit 24 can obtain the phase θi of the short-circuit current by calculating the inverse tangent arctan (Iβ / Iα) of the currents Iα and Iβ in the αβ coordinate system. PI control is performed by the PI controller 25 so that the θi and the d-axis phase θdc on the control side match. The PI control is called a PLL control (Phase Locked Loop control), and the output of the PI control can be used as the speed estimation value ω _PLL ^.

The speed estimation initial value switching unit 20 inputs the speed estimation initial value ω _PLL ^ calculated by the short-circuit current frequency calculation unit 22 or the speed estimation initial value ω _{ini_Iq} calculated by the speed estimation initial value calculation unit 21. The speed estimation initial value switching unit 20 switches this input depending on whether or not the amplitude | I1 | of the short-circuit current I1 is equal to or greater than a predetermined threshold value Th. When the amplitude | I1 | is equal to or more than the threshold Th, the speed estimation initial value ω _PLL ^ calculated by the short-circuit current frequency calculation unit 22 is selected as an input, and when it is less than the threshold Th, the speed estimated initial value calculation unit 21 calculates the speed. The estimated initial value ω _{ini_Iq} is selected as the input. Thereby, the initial value of the velocity estimation according to the residual magnetic flux amount can be set as the initial value of the integral of the above equation (8) in the velocity estimation unit 19.

The explanation is returned to FIG. The short-circuit current calculation unit 9 calculates the amplitude | I1 | of the short-circuit current I1 using the above equation (1).

The voltage mode switching unit 14 inputs a voltage command calculated by the AC voltage calculation unit 11 or a voltage command calculated by the DC voltage calculation unit 13. The voltage mode switching unit 14 switches depending on whether or not the amplitude | I1 | of the short-circuit current I1 is equal to or greater than a predetermined threshold value. If the amplitude | I1 | is equal to or greater than the threshold value Th, the voltage command calculated by the AC voltage calculation unit 11 is selected as an input, and if it is less than the threshold value Th, the voltage command calculated by the DC voltage calculation unit 13 is selected as an input. When the voltage command calculated by the DC voltage calculation unit 13 is selected, the voltage mode switching unit 14 selects the voltage command calculated by the AC voltage calculation unit 11 after the DC voltage application period has elapsed.

The above is the description of the components of the first embodiment.

With reference to FIG. 6, the speed estimation process of the induction motor 1 by the controller 4 including the AC voltage calculation unit 11 of the first embodiment will be described. FIG. 6 is a flowchart showing an example of speed estimation processing of the induction motor 1 by the controller 4 including the AC voltage calculation unit 11 of the first embodiment.

First, in step S21, the controller 4 short-circuits the motor terminals of the induction motor 1. The process of step S21 is the same as the process of step S11 of FIG. Next, in step S22, the short-circuit current frequency calculation unit 22 calculates and outputs the speed estimation value ω _PLL ^.

Next, in step S23, the controller 4 determines whether or not to shift to the DC application mode. The process of step S23 is the same as the process of step S12 of FIG. The controller 4 shifts the process to step S24 when shifting to the DC application mode (step S23YES). On the other hand, the controller 4 shifts the process to step S26 when shifting to the AC application mode (step S23NO).

In step S24, the controller 4 shifts to the DC voltage application mode and applies DC to the induction motor 1. The process of step S24 is the same as the process of step S13 of FIG. Next, in step S25, the speed estimation initial value calculation unit 21 calculates and outputs the initial value ω _{ini_Iq} of the speed estimation.

In step S23NO following step S23NO or step S25, the speed estimation initial value switching unit 20 determines whether or not the amplitude | I1 | of the short-circuit current I1 is equal to or greater than the short-circuit current amplitude threshold Th. The speed estimation initial value switching unit 20 uses the above equation (the speed estimation value ω _PLL ^ calculated by the short-circuit current frequency calculation unit 22 when the amplitude | I1 | of the short-circuit current I1 is equal to or greater than the short-circuit current amplitude threshold Th (YES). It is set to the initial value of the integration of 8) (step S27). On the other hand, in the speed estimation initial value switching unit 20, when the amplitude | I1 | of the short-circuit current I1 is less than the short-circuit current amplitude threshold Th (step S26NO), the speed estimation initial value calculation unit 21 calculates the initial speed estimation value ω _{ini_Iq.} Is set to the initial value of the integration of the above equation (8) (step S28).

In step S29 following step S27 or step S28, the controller 4 shifts to the AC voltage application mode, applies AC to the induction motor 1, and estimates the speed by the speed estimation unit 19. The process of step S29 is the same as that of step S14 of FIG.

Next, referring to FIG. 7, as a result of the threshold determination of the amplitude | I1 | of the short-circuit current I1 in step S12 of FIG. 2 and step S23 of FIG. 6, the short-circuit current I1 and three-phase when shifting to the DC voltage application mode The waveforms of the AC voltages Vu, Vv, Vw, the q-axis current detection value Iq, and the speed estimation value ωr ^ will be described. FIG. 7 is a diagram showing an example of a speed estimation waveform when DC voltage application is selected after determining the amplitude of the short-circuit current.

As shown in FIG. 7A, the amplitude | I1 | is less than the threshold Th at the time of the amplitude | I1 | determination timing t12 at which the short-circuit period T11 from the inverter restart timing t11 ends (| I1 | <Th). Therefore, the DC voltage is applied to the induction motor 1 during the DC voltage application period T12 at the timings t12 to t13. Specifically, as shown in FIG. 7B, for example, a positive voltage is applied to the U phase of the induction motor 1, and the same negative voltage is applied to the V phase and the W phase.

As a result, as shown in FIG. 7C, the q-axis current detection value Iq flows through the induction motor 1 during the DC voltage application period T12. Then, as shown in FIG. 7D, at the DC voltage application end timing t13 when the DC voltage application period T12 ends and the AC voltage application period (speed estimation period) T13 starts, the speed estimation initial value calculation unit 21 , _{Positive /} negative of the q-axis current detection value Iq of the induction motor 1, that is, the initial value ω _{ini_Iq} of the speed estimation according to the rotation direction is obtained (see reference numeral P1 in FIG. 7).

Therefore, in order to shift to the speed estimation by applying the AC voltage with the initial value ω _{ini_Iq} of the speed estimation according to the rotation direction of the induction motor 1 set, the AC voltage application period (speed estimation period) T13 at the timings t13 to t14. It is possible to estimate the speed in a shorter period of time.

Next, referring to FIG. 8, as a result of the threshold determination of the amplitude | I1 | of the short-circuit current I1 in step S12 of FIG. 2, the short-circuit current I1, the three-phase AC voltage Vu, Vv, when shifting to the AC voltage application mode, The waveforms of Vw, the q-axis current detection value Iq, and the velocity estimation value ωr ^ will be described. FIG. 8 is a diagram showing an example of a speed estimation waveform when AC voltage application is selected after determining the amplitude of the short-circuit current.

As shown in FIG. 8A, the amplitude | I1 | is equal to or higher than the threshold value Th at the time of the amplitude | I1 | determination timing t22 at which the short-circuit period T21 from the inverter restart timing t21 ends (| I1 | ≧ Th). Therefore, as shown in FIG. 8B, the AC voltage is applied to the induction motor 1 during the AC voltage application period (speed estimation period) T22 at the timings t22 to t23.

At this time, the voltage mode switching unit 14 selects the U-phase voltage command Vu ^* , V-phase voltage command Vv ^* , and W-phase voltage command Vw ^* output by the AC voltage calculation unit 11, and outputs the short-circuit current frequency calculation unit 22. The initial value ω _PLL ^ of the speed estimation is used as the initial value ω _ini of the speed estimation unit 19 (see FIG. 8C). ω _PLL ^ ≒ ωr, and by using the initial speed estimation value ω _PLL ^ as the initial value ω _ini of the speed estimation unit 19, the speed estimation can be started with the approximate value of the true speed value obtained, and the speed estimation can be started. The accuracy can be improved. Further, when the short circuit is completed, the amplitude | I1 | of the short circuit current I1 is equal to or higher than the threshold value Th. Therefore, since the speed is estimated by applying the AC voltage in the state where ω _ini = ω _PLL ^ ≈ωr is set, the speed can be estimated in a shorter period of time.

In the present embodiment, when there is residual magnetic flux, the short-circuit current frequency calculation unit 22 can calculate the frequency of the short-circuit current and set the initial speed estimation value. As a result, the difference between the initial speed estimation value and the true speed value becomes small. In this case, since the motor terminals are only short-circuited, it is not necessary to increase the response of the controller 4. In addition, when switching from the short-circuited state to the AC voltage application, the speed estimation by applying the AC voltage can be started when the difference between the initial speed estimation value and the true speed value is small, so that it is not necessary to increase the response of the AC voltage calculation unit 11. Is.

On the other hand, when the residual magnetic flux is zero, the DC voltage calculation unit 13 applies a DC voltage to the induction motor 1. At this time, the rotation direction can be determined from the positive and negative of the q-axis current detected value flowing by applying the DC voltage, and the speed estimation by applying the AC voltage can be started in the state where the initial speed estimation value according to the rotation direction is obtained. As a result, the speed can be estimated quickly regardless of the rotation speed. Therefore, according to the present embodiment, it is not necessary to increase the response of the controller 4, and the speed can be estimated in a short period of time.

[Embodiment 2]
The second embodiment will be described below with reference to FIGS. 9 to 11.

FIG. 9 is a functional block diagram showing a configuration example of the AC voltage calculation unit 11B of the second embodiment. The AC voltage calculation unit 11B has a configuration that replaces the AC voltage calculation unit 11 of the first embodiment in the drive system 2S of the second embodiment, and in addition to the components of the AC voltage calculation unit 11, the first phase adjustment unit 23 is further added. Have.

As shown in FIG. 9, the first phase adjusting unit 23 sets the input as the speed estimation value ω _PLL ^ by the short-circuit current frequency calculation unit 22, and sets the output as the compensation amount θcmp-1. The first phase adjusting unit 23 outputs the compensation amount θcmp-1 according to the speed estimation value ω _PLL ^ at the AC voltage application start timing (timing t13 in FIG. 7 and timing t22 in FIG. 8), and at other timings. Output 0. The compensation amount θcmp-1 calculated by the first phase adjusting unit 23 is input to the dq coordinate inverse conversion unit 18b and the dq coordinate conversion unit 8, and θcmp-1 is added to each current phase.

Next, the problems when the phase adjustment is not performed will be described. FIG. 10 is a vector diagram at the time of a short circuit, and FIG. 11 is a vector diagram after phase adjustment of the second embodiment. When the d-axis (hereinafter referred to as the dc-axis) on the control side is taken in the direction of the residual magnetic flux Φ1 at the time of a short circuit, the following equations (10) and (11) are established from the voltage equation of the induction motor.

However, since it is short-circuited, Vdq = 0. Further, F (s) = R _σ + L _σ s, and Φ _2d ≧ 0. Here, if the phase difference between the short-circuit current I1 and the residual magnetic flux Φ1 is defined as Δθ, the phase difference Δθ is expressed by the following equation (12).

Now, since the dc axis is taken in the residual magnetic flux Φ1 direction, the condition for | Δθ |> 90 deg is Id <0. Therefore, from the above equation (10), when ωr> F (s) / (ω1LσT ₂ ) = (Rσ + Lσs) / (ω1LσT ₂ ), | Δθ |> 90deg, and as shown in FIG. 10, the negative of the dc axis. The Φ _2d component of the residual magnetic flux Φ1 exists in the direction of. As a result, the rise of Φ _2d becomes slow, and it takes time to estimate the speed.

Therefore, in the present embodiment, the first phase adjusting unit 23 adjusts the phase so that Φ _2d > 0, and shortens the speed estimation time. When the first phase adjusting unit 23 adds the compensation amount θcmp-1 = Δθ to the current phases of the dq coordinate inverse conversion unit 18b and the dq coordinate conversion unit 8, the dc axis is rotated and the dc axis is rotated as shown in FIG. And the direction of the residual magnetic flux Φ1 coincide with each other. Further, regarding the value of the compensation amount θcmp-1, the relationship between Δθ and the rotation speed may be obtained in advance and tabulated. The compensation amount θcmp-1 does not necessarily have to match Δθ, and may be, for example, an approximate value of the above equation (12).

The above is the description of the second embodiment. According to the present embodiment, the first phase adjusting unit 23 adjusts Δθ, and after the phase adjustment, Φ _2d > 0, so that the speed estimation period can be shortened.

[Embodiment 3]
The third embodiment will be described below with reference to FIGS. 12 to 14.

FIG. 12 is a functional block diagram showing a configuration example of the AC voltage calculation unit 11C of the third embodiment. The AC voltage calculation unit 11C has a configuration that replaces the AC voltage calculation unit 11 of the first embodiment in the drive system 3S of the third embodiment, and in addition to the components of the AC voltage calculation unit 11, the second phase adjustment unit 27 and Further, it has a d-axis magnetic flux calculation unit 28 that estimates the amount of d-axis magnetic flux.

The second phase adjusting unit 27 inputs the speed estimation value ω _PLL ^ by the short-circuit current frequency calculation unit 22, and outputs the compensation amount θcmp-2. The second phase adjusting unit 27 outputs the compensation amount θcmp-2 according to the speed estimation value ω _PLL ^ at the AC voltage application start timing, and outputs 0 at other timings. Further, the compensation amount θcmp-2 at this time is such that the product of the d-axis current detection value Id after the phase adjustment and the mutual inductance M is equal to the Φ _2d component of the residual magnetic flux Φ1 at the AC voltage application start timing. , Set according to the speed.

The d-axis magnetic flux calculation unit 28 calculates the d-axis magnetic flux estimated value Φ _2d ^ by multiplying the d-axis current detection value Id obtained from the dq coordinate conversion unit 8 with the mutual inductance M. Further, the value of the d-axis magnetic flux estimated value Φ _2d ^ is set to the initial value of the integrator in the LPF 16b of the AC voltage calculation unit 11C. The initial value of the integrator in LPF16b is the initial value [Phi _{2d * _INI} of [Phi _2d ^*.

FIG. 13 is a vector diagram of the third embodiment after phase adjustment. FIG. 13 is a vector diagram when the compensation amount θcmp-2 is set so that the d-axis current detection value Id after the phase adjustment by the second phase adjustment unit 27 establishes M · Id = Φ _2d .

Next, the problem when the initial value of the magnetic flux command is not set will be described. Φ _2d = Φ _2d ^* is a precondition of the above equation (8), but in the presence of residual magnetic flux, Φ _2d ≠ Φ _2d ^* , and the above equation (8) does not hold. Under the condition of Φ _2d ≠ Φ _2d ^* in the above equation (8), even if Iq ^* −Iq = 0, ωrΦ _2d −ωr ^ Φ _2d ^* = 0⇔ωr ^ = Φ _2d / (Φ _2d ^* ) ωr Therefore, an error remains in the speed estimate.

Here, if α is a real number and Φ _2d = αΦ _2d ^* , then ωrΦ _2d −ωr ^ Φ _2d ^* = 0⇔ (αωr−ωr ^) Φ _2d ^* = 0. If Φ _2d ^* ≠ 0, then αωr−ωr ^ = 0. When Φ _2d is assumed to converge to Φ _2d ^* α converges to 1, ωr ^ = becomes a ωr, take the time to Φ2d converges to Φ2d ^*.

From these, but if satisfied Φ _2d = Φ _2d ^* Iq is completed speed estimation at the time of convergence to Iq ^*, also Φ _2d ≠ Φ _2d ^* when Iq is converged to Iq ^*, further Φ _2d must wait until it converges to Φ _2d ^*. Therefore, when Φ _2d ≠ Φ _2d ^* , it takes time for the estimated velocity value to converge to the true value.

Therefore, in the present embodiment, the d-axis magnetic flux calculation unit 28 sets Φ _2d ^* _{_ini} . Due to the phase adjustment of the second phase adjusting unit 27, M · Id = Φ _2d at the AC voltage application start timing. Further, in the d-axis magnetic flux calculation unit 28, M · Id = Φ _2d ^ is established. Since Φ _2d ^ is set to the initial value of the integrator in LPF16b, Φ _2d ^ = Φ _2d ^* _{_ini} . Thus the ^{_{M · Id ^ = Φ 2d *}} _ini ⇔Φ 2d = Φ 2d * _ini.

FIG. 14 is a diagram showing an example of the velocity estimation waveform of the third embodiment. With reference to FIG. 14, the d-axis current detection value Id, the d-axis magnetic flux Φ2d, the three-phase AC voltage Vu, Vv, Vw, and the speed estimation value ωr ^ when the phase is adjusted when shifting to the AC voltage application mode. The waveform will be described. In the waveform of the d-axis current detection value Id and the waveform of the d-axis magnetic flux Φ2d, the broken line in the horizontal axis direction indicates the command value, and the solid line indicates the detected value.

The short-circuit period T31 from the inverter restart timing t31 to the phase adjustment timing t32 ends, and the second phase adjustment unit 27 outputs the compensation amount θcmp-2, so that the dc axis rotates. As a result, the value of Id changes (see reference numeral P31 in FIG. 14A). At this time, since M · Id = Φ _2d as shown in FIG. 13, the d-axis magnetic flux estimated value Φ _2d ^ estimated from the d-axis current by the d-axis magnetic flux calculation unit 28 is set in the LPF 16b of the AC voltage calculation unit 11. By setting the initial value of the integrator, as shown by the reference numeral P32 in FIG. 14B, Φ _2d and Φ _2d ^* at the start of speed estimation by applying the AC voltage are matched, and the amplitude of the AC voltage is adjusted. be able to.

The above is the description of the third embodiment. According to this embodiment, the initial value of the d-axis magnetic flux command can be set according to the magnitude of the residual magnetic flux, and the effect that the speed estimation time is shortened is obtained.

[Embodiment 4]
The components of the fourth embodiment will be described. As shown in FIG. 15, the present embodiment further includes a speed detecting means 29 for detecting the speed of the induction motor 1 in addition to the components of the first embodiment in the drive system 4S of the fourth embodiment. FIG. 15 is a functional block diagram showing a configuration example of the drive system of the fourth embodiment.

Further, in the present embodiment, the AC voltage calculation unit 11D shown in FIG. 16 has a configuration that replaces the AC voltage calculation unit 11 of the first embodiment in the drive system 4S, and in addition to the components of the AC voltage calculation unit 11, it rotates. A direction determining unit 30 and a multiplier 31 are further provided. FIG. 16 is a functional block diagram showing a configuration example of the AC voltage calculation unit of the fourth embodiment. In the present embodiment, the amplitude at the start of application of the AC voltage to the induction motor 1 is adjusted according to the speed detection value obtained from the speed detection means 29.

The speed detecting means 29 is a resolver or the like, detects the absolute value | ωr | of the speed of the induction motor 1, and inputs it to the multiplier 31 in the AC voltage calculation unit 11D.

The rotation direction determination unit 30 determines the rotation direction of the induction motor 1 from the positive / negative of ω _ini , which is the output of the speed estimation initial value switching unit 20, and outputs the rotation direction information Rd. When ω _ini is positive, Rd = 1, and when ω _ini is negative, Rd = -1.

The multiplier 31 multiplies | ωr | obtained by the speed detecting means 29 and Rd, which is the output of the rotation direction determining unit 30, and outputs the result to the speed estimating unit 19.

Issues when this embodiment is not implemented will be described. When this embodiment is not implemented, ω _ini becomes ω _{ini_Iq} which is the output of the speed estimation initial value calculation unit 21 or ω _PLL ^ which is the output of the short-circuit current frequency calculation unit 22. Since one value of ω _{ini_Iq} is set in advance for each of forward and backward, a torque shock occurs at the start of speed estimation if ωr is other than that value. Further, since ω _PLL ^ is an approximate value of ωr, a torque shock occurs at the start of speed estimation due to the speed difference.

On the other hand, in the present embodiment, the rotation direction can be known by assigning a code corresponding to the rotation direction to | ωr | detected by the speed detection means 29, and the true value of the rotation speed is calculated by the above equation (8). ) Can be set to the initial value of the integrator. As a result, the speed estimation error at the start of speed estimation can be made zero, and the torque shock during the speed estimation period can be reduced.

[Embodiment 5]
Hereinafter, the drive systems 1S, 2S, 3S, and 4S including the induction motor 1 and the controllers 4, 4B, 4C, and 4D of the above-described first to fourth embodiments are, for example, electric vehicles such as electric railway vehicles and electric vehicles, and industrial machines. It is suitable for application to a driving device or the like. In this embodiment, the electric vehicle 100 to which the drive systems 1S to 4S of the above-described first to fourth embodiments are applied will be described. FIG. 17 is a schematic view showing an example of the electric vehicle 100 of the fifth embodiment.

Electric power is supplied from the overhead wire 103 to the drive systems 1S to 4S of the electric vehicle 100 via the current collector 101, and the drive systems 1S to 4S drive the induction motor 1 for rotating the wheels 102 on the rail 200. ..

In the low-slip induction motor 1, residual magnetic flux tends to remain, and the velocity estimation may fail due to the disturbance of the induced voltage generated in proportion to the residual magnetic flux and the rotation speed. Therefore, when the low-slip induction motor 1 is applied to the electric vehicle 100, the riding comfort may be deteriorated due to the occurrence of torque shock due to the failure of speed estimation. Therefore, by applying any of the drive systems 1S to 4S including the controllers 4 to 4D of the first to fourth embodiments to the electric vehicle 100, it is possible to estimate the speed in a short period of time regardless of the residual magnetic flux amount. Moreover, it is possible to realize the electric vehicle 100 which does not require high response of the control system.

The present invention is not limited to the above-described embodiment, and includes various modifications. For example, the above-described embodiment has been described in detail in order to explain the present invention in an easy-to-understand manner, and is not necessarily limited to the one including all the described configurations. Further, it is possible to replace a part of the configuration of one embodiment with the configuration of another embodiment, and it is also possible to add the configuration of another embodiment to the configuration of one embodiment. Further, it is possible to add / delete / replace / integrate / distribute other configurations for a part of the configurations of each embodiment. Further, each process shown in the embodiment may be appropriately distributed or integrated based on the processing efficiency or the mounting efficiency.

1S, 2S, 3S, 4S: Drive system, 1: Induction motor, 3: Inverter, 4,4B, 4C, 4D: Controller, 6: Gate driver, 7: αβ coordinate conversion unit, 8: dq coordinate conversion unit, 9: Short-circuit current calculation unit, 10: AC current command calculation unit, 11, 11B, 11C, 11D: AC voltage calculation unit, 12: DC current command calculation unit, 13: DC voltage calculation unit, 14: Voltage mode switching unit, 15: PWM signal generator, 100: electric vehicle

Claims (12)

It has a power conversion device that drives an induction motor and a control device that controls the power conversion device.
The control device is
When power is output to the induction motor after the power conversion device is restarted, an AC voltage is applied to the induction motor based on a short-circuit current detection value detected by short-circuiting the motor terminals of the induction motor. A drive device for an induction motor, which is characterized by switching between a mode and a control mode of a DC voltage application mode in which a DC voltage is applied. The control device is
The claim is characterized in that when the amplitude of the short-circuit current detection value is less than a predetermined threshold value, the mode is switched to the DC voltage application mode, and when the amplitude is greater than or equal to the predetermined threshold value, the mode is switched to the AC voltage application mode. The drive device for an induction motor according to 1. The control device is
After shifting to the DC voltage application mode, a DC voltage is applied to the induction motor over the DC voltage application period, and after the DC voltage application period ends, the mode is switched to the AC voltage application mode.
The drive device for an induction motor according to claim 1 or 2, wherein the speed of the induction motor is estimated over a speed estimation period after shifting to the AC voltage application mode. The control device is
When shifting from the DC voltage application mode to the AC voltage application mode, it is determined from the q-axis current flowing through the induction motor whether the induction motor is in the forward or backward rotation direction, and according to the determined rotation direction. The drive device for an induction motor according to claim 3, wherein an initial value for estimating the speed of the induction motor in the AC voltage application mode is set. The control device is
The drive device for an induction motor according to any one of claims 1 to 4, wherein the phase of the residual magnetic flux of the induction motor is adjusted according to the speed of the induction motor. The control device is
Adjusting the phase at the start of application of the AC voltage to the induction motor in the AC voltage application mode so that the phase difference between the phase of the short-circuit current detection value and the phase of the residual magnetic flux of the induction motor becomes zero. The drive device for an induction motor according to claim 5. The control device is
The induction motor in the AC voltage application mode so that the estimated magnetic flux obtained by multiplying the mutual inductance of the induction motor and the d-axis component amplitude of the short-circuit current detection value and the amplitude of the residual magnetic flux of the induction motor match. The drive device for an induction motor according to claim 5, wherein the phase at the start of application of an AC voltage to the device is adjusted. The control device is
7. The invention is characterized in that the amplitude at the start of application of the AC voltage to the induction motor is adjusted by setting the estimated magnetic flux value to the initial value of the magnetic flux command of the induction motor in the AC voltage application mode. The drive device for an induction motor according to. The control device is
Any one of claims 1 to 8, wherein the amplitude at the start of application of the AC voltage to the induction motor is adjusted according to the speed detection value obtained from the speed detection means for detecting the speed of the induction motor. The drive device for an induction motor according to the section. The control device is
Estimate the rotation direction of the induction motor
It is characterized in that the rotation speed based on the estimated rotation direction and the amplitude of the speed detection value obtained from the speed detection means is set as the initial value of the speed estimation at the start of application of the AC voltage to the induction motor. The drive device for an induction motor according to claim 9. It is a method of driving an induction motor,
The drive device of the induction motor is
It has a power conversion device for driving the induction motor and a control device for controlling the power conversion device.
The control device
When power is output to the induction motor after the power conversion device is restarted, an AC voltage is applied to the induction motor based on a short-circuit current detection value detected by short-circuiting the motor terminals of the induction motor. A method for driving an induction motor, which comprises switching between a mode and a control mode of a DC voltage application mode in which a DC voltage is applied. An electric vehicle provided with the drive device for the induction motor according to any one of claims 1 to 10.

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