JP2019097341A - Motor controller and motor system - Google Patents

Abstract

To reduce an axial error while suppressing a wasteful current before changeover into a position sensorless control mode in a motor control device for controlling a motor by vector control.SOLUTION: A motor controller performs changeover from a synchronous control mode to a position sensorless control mode during a sensorless changeover period. In the relevant period, the motor controller calculates a q-axis current "Iq0" based on an axial error "Δθ0" of a real axis (dq axis) and a control axis (γδ axis) at a start time of the relevant period and a γ axis target current "Iγo" to specify "Iq0×(1+Kg)" as a target value with a prescribed coefficient being "Kg" (Kg>0). Then, the motor controller repeats processing to acquire the axial error "Δθ0" and processing to increase a δ axis target current Iδso that while decreasing a y-axis target current Iγan addition value of its q axis component and a q-axis component of a δ-axis target current becomes equal to a target value until the axial error Δθ becomes equal to or smaller than a prescribed threshold value.SELECTED DRAWING: Figure 3

Description

  The present invention relates to a motor control device and a motor system, and, for example, to switching technology from a synchronous control mode to a position sensorless control mode.

  Patent Document 1 shows a method for reducing switching shock when switching from the synchronous control mode to the position sensorless control mode. In this method, during the synchronous control mode, the current vector phase θs is controlled to reduce the axis error Δθc between the rotor axis of the motor and the control system axis of dq vector control, and the axis error Δθc is reduced. Switching to the position sensorless control mode is performed.

JP, 2010-29016, A

  In recent years, as a control method of a three-phase motor represented by a permanent magnet synchronous motor (PMSM), a vector control method without position sensor is widely used. In this method, an induced voltage is calculated from the terminal voltage and terminal current of a three-phase motor, and the position and speed of the rotor are estimated based on the induced voltage. However, at the time of stop or low speed rotation of the rotor, the induced voltage proportional to the rotational speed is very small, so it is difficult to correctly estimate the position and speed of the rotor and to control the motor correctly. Therefore, during stop to low speed rotation of the rotor, the rotational speed is usually raised using open loop control (synchronous control mode) with the rotor drawn into the d-axis, and then switching to the position sensorless control mode Is done.

  However, in the synchronous control mode, as shown in Patent Document 1, an axis error occurs depending on the state of the load. Therefore, when the synchronous control mode is switched to the position sensorless control mode as it is, a switching shock such as a sudden jump of the current or vibration of the rotational speed may occur. On the other hand, it is possible to use the system of patent document 1 as the countermeasure. However, in the method of Patent Document 1, when switching from the synchronous control mode to the position sensorless control mode, the axis error is reduced by controlling the phase while the amplitude value of the current vector is fixed. A large amount of unnecessary current (d-axis current) may flow.

  The embodiment to be described later is made in view of the foregoing, and other problems and novel features will be apparent from the description of the present specification and the accompanying drawings.

  The motor control device according to one embodiment controls the three-phase motor by vector control without position sensor, and switches from synchronous control mode with open loop control to position sensorless control mode with closed loop control. It carries out in the switching period in synchronous control mode using processing of 2. In the first process, the motor control device calculates the q-axis current based on the axis error between the real axis (dq axis) and the control axis (γδ axis) at the start time of the switching period and the γ axis target current. Assuming that the q-axis current is “Iq0” and the predetermined coefficient is “Kg” (Kg> 0), “Iq0 × (1 + Kg)” is set as the target value. In the second process, the motor control device performs a process A for acquiring an axis error, and decreases the γ-axis target current, and reduces the q-axis component of the decreased γ-axis target current and the q-axis component of the δ-axis target current. A process B of increasing the δ-axis target current so that the addition value becomes equal to the target value is repeated until the axis error becomes equal to or less than a predetermined threshold value, and then the mode is switched to the position sensorless control mode.

  According to the one embodiment, it is possible to reduce the axis error while suppressing unnecessary current before switching to the position sensorless control mode.

FIG. 1 is a schematic view showing a configuration example of a motor system according to a first embodiment of the present invention. It is a schematic diagram which shows the structural example and coordinate axis of a three-phase motor in FIG. It is a timing chart which shows the general operation example of the motor system of FIG. FIG. 7 is a diagram for explaining an operation state in a period of positioning control mode to synchronous control mode of FIG. 3 in the motor system of FIG. FIG. 6 is a diagram for explaining an operation state in a position sensorless control mode of FIG. 3 in the motor system of FIG. 1. It is a vector diagram corresponding to the operation state in synchronous control mode of FIG. It is a vector diagram corresponding to the operation state in the position sensorless control mode of FIG. It is the schematic which shows the structural example of the control mode switching control part in FIG. It is a flowchart which shows the operation example in the sensorless switching period in the control mode switching control part of FIG. It is a vector diagram which shows an example of the change of the various operation states accompanying the process in the sensorless switching period of FIG. The motor control apparatus by Embodiment 2 of this invention WHEREIN: It is the schematic which shows the structural example of the control mode switching control part of FIG. It is the schematic which shows the structural example of the coefficient calculating part in FIG. FIG. 13 is a schematic view showing a configuration example in which the coefficient calculation unit of FIG. 12 is modified in the motor control device according to Embodiment 3 of the present invention. (A) And (b) is a figure explaining the axial difference | error reduction system in the motor control apparatus which becomes a comparative example of this invention.

  In the following embodiments, when it is necessary for the sake of convenience, it will be described by dividing into a plurality of sections or embodiments, but unless specifically stated otherwise, they are not mutually unrelated, one is the other Some or all of the variations, details, supplementary explanations, etc. are in a relation. Further, in the following embodiments, when referring to the number of elements (including the number, numerical value, quantity, range, etc.), it is particularly pronounced and clearly limited to a specific number in principle. Except for the specific number, it is not limited to the specific number, and may be more or less than the specific number.

  Furthermore, in the following embodiments, the constituent elements (including element steps and the like) are not necessarily essential unless explicitly stated or considered to be obviously essential in principle. Needless to say. Similarly, in the following embodiments, when referring to the shapes, positional relationships and the like of components etc., the shapes thereof are substantially the same unless particularly clearly stated and where it is apparently clearly not so in principle. It is assumed that it includes things that are similar or similar to etc. The same applies to the above numerical values and ranges.

  Hereinafter, embodiments of the present invention will be described in detail based on the drawings. In all the drawings for describing the embodiments, the same reference numeral is attached to the same member in principle, and the repetitive description thereof will be omitted.

Embodiment 1
<< Configuration of motor system >>
FIG. 1 is a schematic view showing a configuration example of a motor system according to a first embodiment of the present invention. FIG. 2 is a schematic view showing a configuration example of the three-phase motor in FIG. 1 and coordinate axes. The motor system shown in FIG. 1 includes a motor control device MCD, an inverter INV, a current sensor ISEN, and a three-phase motor MT. The three-phase motor MT is, for example, a permanent magnet synchronous motor (PMSM) (in other words, a brushless DC motor).

  Specifically, as shown in FIG. 2, the three-phase motor MT includes external terminals PNu, PNv, PNw of three phases (u phase, v phase, w phase), and a rotor RT as a permanent magnet. And the stator can be represented by an equivalent circuit. When viewed on a three-phase axis (u-axis, v-axis, w-axis) as fixed coordinates, the three-phase motor MT has three-phase resistance components Ru, Rv, Rw, inductance components Lu, Lv, Lw and an induced voltage Eu , Ev, Ew. Taking the u phase as an example, the resistance component Ru, the inductance component Lu and the induced voltage Eu are coupled in series between the middle point and the u-phase external terminal PNu with the rotor RT as the middle point. The u-axis, v-axis and w-axis are arranged in the current-flowing direction (current direction) of the u-phase, v-phase and w-phase, respectively, and are arranged in order at an electrical angle of 120 ° with respect to the rotation direction of the rotor RT.

  On the other hand, the fixed coordinates (three-phase axes) can be converted into two-phase axes (d-axis, q-axis) serving as rotational coordinates. The d-axis is disposed in the magnetic flux direction of the rotor RT, and the q-axis is disposed in a direction orthogonal to the d-axis. The rotational phase θ usually represents the phases of the u-axis and the d-axis, and is information representing the positional relationship between the three phases (u-phase, v-phase, w-phase) and the rotor RT. When viewed on a two-phase axis as rotational coordinates, the three-phase motor MT is represented by a resistance component Rs, a d-axis inductance component Ld and a q-axis inductance component Lq, and a d-axis induced voltage Ed and a q-axis induced voltage Eq. Be done. However, the d-axis induced voltage Ed is zero. When a current in the q-axis direction (q-axis current Iq) flows by vector combination of three-phase currents, torque in the rotational direction is generated with respect to the rotor RT based on Fleming's law. On the other hand, the current in the d-axis direction (d-axis current Id) usually does not contribute to the torque.

  Also, as a structure of the three-phase motor MT (specifically, the rotor RT), an SPM (Surface Permanent Magnet) structure (non salient pole structure) with Lq = Ld and an IPM (Interior Permanent Magnet) structure with Lq ≠ Ld (Salted electrode structure) is known. In the first embodiment, the three-phase motor MT has an SPM structure, but may have an IPM structure. In the case of the IPM structure, torque (reactance torque) is also generated by the d-axis current Id. The resistance component Rs has the same value in the d axis and the q axis regardless of the SPM structure and the IPM structure.

  Returning to FIG. 1, the inverter INV functions as a motor driver and drives voltages Vu and Vv based on PWM (Pulse Width Modulation) signals PWMu, PWMv and PWMw on the external terminals PN (u, v, w) of the three-phase motor MT. , Vw are applied respectively. The inverter INV is not shown, but a three-phase high side switching element provided between the three-phase external terminal PN (u, v, w) and the high potential side power supply voltage, and a three-phase external terminal A three-phase low-side switching element provided between PN (u, v, w) and the low potential side power supply voltage is provided.

  The current sensor ISEN detects a u-phase current Iu, a v-phase current Iv, and a w-phase current Iw flowing through the three-phase motor MT. Specifically, for example, the current sensor ISEN is a three-shunt method in which a three-phase current is detected by inserting a resistor on the low side of each phase in the inverter INV, or a low potential power supply voltage of the inverter INV. This is realized by a one-shunt method or the like in which one resistor is inserted and a three-phase current is detected by devising sampling timing.

  The motor control device MCD is configured of, for example, a microcontroller, etc. equipped with a processor, an analog / digital converter, a PWM modulator, etc., and controls the three-phase motor MT via the inverter INV by vector control. That is, the internal unit constituting the motor control device MCD is mainly implemented by program processing by the microcontroller. However, as a matter of course, the present invention is not limited to this, and part or all of the internal units can be configured by dedicated hardware. Hereinafter, each internal unit of the motor control device MCD will be described.

  The three-phase / two-phase converter AXCC functions as a current detection unit that detects the γ-axis current Iγ and the δ-axis current Iδ based on each phase current (Iu, Iv, Iw) from the current sensor ISEN. Specifically, the three-phase / two-phase converter AXCC sequentially uses each phase current from the current sensor ISEN every predetermined control period (for example, several tens to several hundreds of μs, etc.) using an analog to digital converter. To detect. The three-phase / two-phase converter AXCC rotates each phase current (Iu, Iv, Iw) of the fixed coordinates (u, v, w coordinates) on the basis of the rotation phase θ from the phase calculator PHC. The γ-axis current Iγ and the δ-axis current Iδ are detected by converting to dq coordinates). The γ axis and the δ axis are control axes of the three-phase motor MT respectively corresponding to the d axis and the q axis (that is, the real axis of the three phase motor MT) shown in FIG. The control axis is equal to the real axis in the ideal state with no axis error.

  The speed command generator RSG generates a rotational speed command ωref which is a target rotational speed of the three-phase motor MT. The subtractor SBrs detects a speed error between the rotation speed command ωref and the rotation speed ωest from the PLL controller PLLC. The speed controller PICrs performs PI (P: Proportional, I: Integral) control or the like based on the speed error from the subtractor SBrs, and generates a δ-axis target current Iδ_r for bringing the speed error closer to zero.

  The control mode switching control unit MDCT controls switching from the synchronous control mode to the position sensorless control mode, which will be described later in detail, and performs the switching in a switching period in the synchronous control mode. The synchronous control mode is used at the start stage of the three-phase motor MT, sets the γ-axis target current Iγ_c to a predetermined value, and increases the target rotational speed by open loop control without using the speed controller PICrs etc. In this mode, the phase motor MT is driven. The position sensorless control mode is a mode in which the three-phase motor MT is driven while controlling the target rotational speed by closed loop control using the speed controller PICrs or the like. . The control mode switching control unit MDCT generates the γ axis target current Iγ_c and the δ axis target current Iδ_c in the period of the synchronous control mode.

  The d-axis current command generator IDG generates the γ-axis target current Iγ_r during the position sensorless control mode. The selector SELid selects the γ-axis target current Iγ_r from the d-axis current command generator IDG or the γ-axis target current Iγ_c from the control mode switching control unit MDCT. The selector SELiq selects the δ-axis target current Iδ_r from the speed controller PICrs or the δ-axis target current Iδ_c from the control mode switching control unit MDCT.

The PWM signal generation unit PWMG generates respective currents of the γ-axis target currents Iγ ^* and δ-axis target currents Iδ ^* from the selectors SELid and SELiq and the γ-axis currents Iγ and δ-axis currents Iδ from the three-phase / two-phase converter AXCC. Based on the error and the rotational phase θ, PWM signals PWMu, PWMv and PWMw for each phase of the three-phase motor MT are generated. Specifically, the PWM signal generation unit PWMG includes subtractors SBid and SBiq, a d-axis current controller PICid, a q-axis current controller PICiq, a two-phase / three-phase converter AXCR, and a PWM modulator PWMMD. Equipped with

The subtractors SBid and SBiq respectively generate the γ-axis target currents Iγ ^* and δ-axis target currents Iδ ^* from the selectors SELid and SELiq and the γ-axis currents Iγ and δ-axis currents Iδ from the 3-phase / 2-phase converter AXCC, respectively. Each current error is detected. The d-axis current controller PICid performs PI control based on the γ-axis current error from the subtractor SBid to generate a γ-axis target voltage Vγ ^* for bringing the γ-axis current error closer to zero. Similarly, the q-axis current controller PICiq performs PI control based on the δ-axis current error from the subtractor SBiq, and generates a δ-axis target voltage Vδ ^* for bringing the δ-axis current error closer to zero.

The two-phase / three-phase converter AXCR performs conversion from rotational coordinates (dq coordinates) to fixed coordinates (u, v, w coordinates) based on the rotational phase θ from the phase calculator PHC, to obtain the γ-axis target voltage Vγ ^* And δ-axis target voltage Vδ ^* is converted into u-phase target voltage Vu ^* , v-phase target voltage Vv ^* and w-phase target voltage Vw ^* . The PWM modulator PWMMD determines three-phase duty ratios for generating three-phase target voltages (Vu ^* , Vv ^* , Vw ^* ), and the three-phase PWM signal PWMu having the three-phase duty ratios. , PWMv and PWMw respectively.

The induced voltage observer BEOB includes the γ-axis target voltage Vγ ^* and the δ-axis target voltage Vδ ^* from each current controller (PICid, PICiq), and the γ-axis current Iγ and the δ-axis current from the three-phase / two-phase converter AXCC. The γ-axis induction voltage Eγ and the δ-axis induction voltage Eδ are calculated based on Iδ and the target rotational speed ω ^* input to the phase calculator PHC. Specifically, the induced voltage observer BEOB calculates the γ axis induced voltage Eγ and the δ axis induced voltage Eδ, respectively, based on the equations (1) and (2). In the equations (1) and (2), “Rs” is the resistance component shown in FIG. 2, and “Lq” is the q-axis inductance component (= d-axis inductance component) shown in FIG. These motor constants (Rs, Lq) are fixedly set in advance or variably set using an identifier (not shown) or the like.

Eγ = Vγ ^* −Rs · Iγ + ω ^* · Lq · Iδ (1)
Eδ = Vδ ^* -Rs · Iδ-ω ^* · Lq · Iγ (2)
The axis error calculator AERC calculates an axis error Δθ between the dq axis as the real axis of the three phase motor MT and the γδ axis as the control axis of the three phase motor MT using the γ axis induced voltage Eγ. Specifically, the axis error calculator AERC calculates the axis error Δθ based on, for example, the equation (3).

Δθ = tan ⁻¹ (Eγ / Eδ) (3)
The PLL controller (timing controller) PLLC generates a rotational speed ωest that causes the axis error Δθ from the axis error calculator AERC to converge to zero, using PLL (Phase Locked Loop) control. The selector SELrs selects the rotation speed command ωref from the speed command generator RSG or the rotation speed ωest from the PLL controller PLLC, and outputs the selected rotation speed as the target rotation speed ω ^* . The phase calculator PHC generates a rotational phase θ by integrating the target rotational speed ω ^* .

<< Schematic operation of motor system >>
FIG. 3 is a timing chart showing a schematic operation example of the motor system of FIG. FIG. 4 is a diagram for explaining the operation state in the period from the positioning control mode to the synchronous control mode of FIG. 3 in the motor system of FIG. FIG. 5 is a diagram for explaining an operation state in the position sensorless control mode of FIG. 3 in the motor system of FIG. FIG. 6 is a vector diagram corresponding to the operation state in the synchronous control mode of FIG. FIG. 7 is a vector diagram corresponding to the operation state in the position sensorless control mode of FIG.

  In the positioning control mode (time t0 to t1) of FIG. 3 and the synchronous control mode (time t1 to t3), as shown in FIG. 4, the d-axis current command generator IDG, the speed controller PICrs and the PLL controller PLLC Is controlled to the off state (inactive state). Further, the selectors SELid, SELiq, and SELrs both select the “S2” side (that is, the outputs of the control mode switching control unit MDCT and the speed command generator RSG).

When the three-phase motor MT is activated, the position of the rotor RT is unknown. Therefore, in the positioning control mode (time t0 to t1), the speed command generator RSG fixes the rotation speed command ωref (target rotation speed ω ^* ) to zero, and the rotation phase θ also becomes zero accordingly. The control mode switching control unit MDCT increases the γ-axis target current Iγ_c (Iγ ^* ) from zero to a predetermined value in a short time with the δ-axis target current Iδ_c (Iδ ^* ) set at zero. As a result, the rotor RT is drawn into the γ-axis, and the γ-axis and the d-axis become substantially coincident (that is, the axis error Δθ is substantially zero).

Subsequently, in a period (time t1 to t2) excluding the sensorless switching period in the synchronous control mode, the speed command generator RSG increases the rotational speed command ωref (target rotational speed ω ^* ) at a constant acceleration rate. Further, the control mode switching control unit MDCT maintains the γ-axis target current Iγ_c (Iγ ^* ) and the δ-axis target current Iδ_c (Iδ ^* ) (here, zero) at time t1. As a result, the rotor RT rotates in synchronization with the target rotational speed ω ^* (a rotational magnetic field due to the rotational phase θ) in a state of being drawn into the γ axis.

However, in fact, in such a synchronous control mode, in order to accelerate the rotor RT at a constant acceleration rate, a torque current corresponding to the inertia, friction, and load of the three-phase motor MT is required. Therefore, as shown in FIG. 6, an axis error Δθ occurs between the control axis (γδ axis) and the real axis (dq axis). The torque current is a q-axis current, and assuming that the necessary q-axis current (torque current) is “Iq0”, the axis error Δθ maintains the relationship of equation (4) with respect to the γ-axis target current Iγ ^* . Occurs on Here, as the acceleration rate becomes higher or the load becomes heavier, the required q-axis current (torque current) Iq0 increases, and along with this, the axis error Δθ also increases.

Iq0 = Iγ ^* × sin (Δθ) (4)
Further, as shown in FIG. 6, although the induced voltage Eq originally occurs on the q axis, the γ-axis induced voltage of 筈 should ideally be zero by the induced voltage observer BEOB if the axis error Δθ exists. Eγ is detected. Therefore, the axis error calculator AERC calculates the axis error Δθ based on the γ axis induced voltage Eγ (specifically, the ratio to the δ axis induced voltage Eδ) as shown in the equation (3). The PLL controller PLLC generates the rotational speed ωest so that the axis error Δθ converges to zero, thereby matching the real axis (dq axis) with the control axis (γδ axis). generates a timing signal synchronized with the rotation of the dq axis).

Returning to FIG. 3, in the sensorless switching period (time t2 to t3) in the synchronous control mode, the control mode switching control unit MDCT reduces the γ-axis target current Iγ_c (Iγ ^* ), while the details will be described later. By increasing the axis target current Iδ_c (Iδ ^* ), the axis error Δθ is reduced. Then, the control mode switching control unit MDCT switches to the position sensorless control mode (time t3) when the axis error Δθ becomes less than or equal to a predetermined threshold value (for example, near zero).

  In the position sensorless control mode (time t3), as shown in FIG. 5, the control mode switching control unit MDCT is in the off state (inactive state) not generating the γ axis target current Iγ_c and the δ axis target current Iδ_c. . Further, the selectors SELid, SELiq, and SELrs both select the “S1” side (ie, the respective outputs of the d-axis current command generator IDG, the speed controller PICrs, and the PLL controller PLLC).

As described above, in the position sensorless control mode (time t3), the motor control device MCD controls the three-phase motor MT by the closed loop control using the PLL controller PLLC, the speed controller PICrs, and the like. As a result, as shown in FIGS. 3 and 7, the axis error Δθ is controlled by the PLL controller PLLC to keep zero, and the δ-axis target current Iδ_r (Iδ ^* ) from the speed controller PICrs is necessary. The magnitude is controlled according to the torque current. Further, the γ-axis target current Iγ_r (Iγ ^* ) from the d-axis current command generator IDG is normally set to zero.

  Here, if the sensorless switching period (time t2 to t3) is not provided in FIG. 3, switching to the position sensorless control mode is performed in a state in which the axis error Δθ is large. In this case, the PLL controller PLLC changes the rotational speed ωest so as to reduce the axis error Δθ, and accordingly, the δ-axis target current Iδ_r from the speed controller PICrs also largely changes. A sensorless switching period (time t2 to t3) is provided to reduce such switching shock.

<< Configuration and operation of control mode switching control unit >>
FIG. 8 is a schematic view showing a configuration example of the control mode switching control unit in FIG. FIG. 9 is a flowchart showing an operation example in a sensorless switching period in the control mode switching control unit of FIG. The control mode switching control unit MDCT shown in FIG. 8 includes a γ current generation unit GCAL, a δ current calculation unit DCAL, a switching sequencer SEQ, and a holding unit MEM. The switching sequencer SEQ switches the positioning control mode, the synchronous control mode, and the position sensorless control mode shown in FIG. 3 based on the target rotational speed ω ^* and the axis error Δθ, and issues an instruction according to each control mode.

Specifically, when the three-phase motor MT is activated, the switching sequencer SEQ sets the control mode to the positioning control mode, and causes the selectors SELid, SELiq, and SELrs to select the “S2” side as shown in FIG. , D-axis current command generator IDG, speed controller PICrs, and PLL controller PLLC are controlled to the off state. In this state, the γ current generation unit GCAL generates and outputs the γ axis target current Iγ_c (Iγ ^* ) as shown in FIG. 3 according to the instruction from the switching sequencer SEQ.

Next, when the γ-axis target current Iγ_c (Iγ ^* ) reaches a predetermined value, the switching sequencer SEQ switches the control mode to the synchronous control mode and notifies the speed command generator RSG to that effect. In response to this, the speed command generator RSG raises the rotational speed command ωref (target rotational speed ω ^* ) at a predetermined acceleration rate. In addition, the γ current generation unit GCAL maintains the γ axis target current Iγ_c (Iγ ^* ) at the time when the positioning control mode is finished (time t1 in FIG. 3).

After that, when the rotational speed command ωref (the target rotational speed ω ^* ) reaches a predetermined value in the synchronous control mode, the switching sequencer SEQ executes the process of the sensorless switching period as shown in FIG. In FIG. 9, the switching sequencer SEQ starts processing of the sensorless switching period (step S101). First, the γ-axis target current Iγ_c (Iγ ^* ) and axis error at the start time of the sensorless switching period (time t2 in FIG. 3) Δθ is held in the holding unit MEM as “Iγ0” and “Δθ0” (step S102).

  Next, the δ current operation unit DCAL calculates the q-axis current Iq0 of the equation (5) in accordance with the instruction from the switching sequencer SEQ (step S103). The q-axis current Iq0 is a torque current required at the start time of the sensorless switching period (time t2 in FIG. 3). Further, the δ current calculation unit DCAL determines a target value Itg shown in the equation (6) using a predetermined coefficient Kg (Kg> 0) (step S103). The target value Itg is set, for example, to a value slightly larger than the required torque current.

Iq0 = Iγ0 × sin (Δθ0) (5)
Itg = Iq0 × (1 + Kg) (6)
Subsequently, the switching sequencer SEQ acquires the axis error Δθ from the axis error calculator AERC (step S104), and determines whether the axis error Δθ is less than or equal to a predetermined threshold value Δθth (for example, a value near zero). (Step S105). If the axis error Δθ is larger than the threshold value Δθth, the γ current generation unit GCAL decreases the γ axis target current Iγ_c (Iγ ^* ) by a predetermined value (ΔIγ) according to an instruction from the switching sequencer SEQ Step S106).

Next, the δ current calculation unit DCAL calculates the δ axis target current I δ — c (I δ ^* ) of Expression (7) in accordance with the instruction from the switching sequencer SEQ (step S107). The δ-axis target current Iδ_c in equation (7) is the q-axis component (sin (Δθ)) of the γ-axis target current Iγ_c in step S106, and the q-axis component (cos (Δθ)) of the δ-axis target current Iδ_c. Is determined to be equal to the target value Itg of equation (6). In other words, the δ current calculation unit DCAL increases the δ axis target current Iδ_c so as to compensate for the decrease of the target value Itg according to the decrease of the γ axis target current Iγ_c in step S106.

Iδ_c = Itg / cos (Δθ) −Iγ_c · tan (Δθ) (7)
Subsequently, the γ current generation unit GCAL and the δ current calculation unit DCAL output the γ axis target current Iγ_c in step S106 and the δ axis target current Iδ_c in step S107, respectively, in accordance with the instruction from the switching sequencer SEQ. (Step S108). Then, the switching sequencer SEQ repeats the processing of steps S104 to S108 every predetermined control cycle (for example, several tens to several hundreds of μs) until the axis error Δθ in step S104 becomes equal to or less than the predetermined threshold value Δθth. .

Thereafter, when the axis error Δθ becomes equal to or less than the predetermined threshold value Δθth, the switching sequencer SEQ switches to the position sensorless control mode (step S109). Specifically, as shown in FIG. 5, the switching sequencer SEQ controls the d-axis current command generator IDG, the speed controller PICrs, and the PLL controller PLLC to the on state. At this time, the switching sequencer SEQ outputs the δ-axis target current Iδ_c (Iδ ^* ) and the target rotational speed ω ^* at the end time of the sensorless switching period (time t3 in FIG. 3) and the output of the speed controller PICrs (Iδ_r And the initial value of the output (ωest) of the PLL controller PLLC.

Then, the switching sequencer SEQ causes the selectors SELid, SELiq, SELrs to select the “S1” side. As a result, when switching from the sensorless switching period to the position sensorless control mode, the δ-axis target current Iδ ^* and the target rotational speed ω ^* can maintain continuity. In the position sensorless control mode, the switching sequencer SEQ controls the γ current generation unit GCAL and the δ current calculation unit DCAL to the off state (inactive state).

  The decrease amount (ΔIr) of the γ-axis target current Iγ_c in step S106 may be a constant value or a value that changes sequentially. When the amount of decrease (ΔIr) is a constant value, the calculations in steps S106 and S107 can be simplified. The sensorless switching period is, for example, several tens of ms to several hundreds of ms.

  FIG. 10 is a vector diagram showing an example of changes in various operation states accompanying the processing in the sensorless switching period of FIG. In FIG. 10, time t2a is a state at the start time of the sensorless switching period (time t2 in FIG. 3). In this state, an axis error occurs so that the γ axis current Iγ flows in the γ axis, and the q axis component (sin (Δθ)) of the γ axis current Iγ becomes equal to the required torque current (q axis current) Iq0. Δθ is occurring.

  At time t2b, the state immediately after execution of steps S106 and S107 is shown. That is, in step S106, the γ-axis current Iγ decreases, and in step S107, the δ-axis current Iδ increases so as to compensate for the decrease in the target value Itg (= Iq0 × (1 + Kg)) associated with step S106. In this state, the sum of the q-axis component (sin (Δθ)) of the γ-axis current Iγ and the q-axis component (cos (Δθ)) of the δ-axis current Iδ becomes equal to the target value Itg. In other words, the q-axis component of the combined current Ia which is the vector addition of the γ-axis current Iγ and the δ-axis current Iδ becomes equal to the target value Itg.

  However, the torque current actually required is "Iq0". Therefore, as shown at time t2c, the control axis (γδ axis) rotates so that the q-axis component of the combined current Ia and the necessary torque current (q-axis current) Iq0 become equal. As a result, the axis error Δθ decreases. State changes such as time t2b and time t2c are repeated until the axis error Δθ becomes equal to or less than the threshold value Δθth.

<< Main effects of Embodiment 1 >>
FIGS. 14 (a) and 14 (b) are diagrams for explaining an axis error reduction method in a motor control device according to a comparative example of the present invention. In the method of the comparative example, a current command calculator ICMD is used. The current command computing unit ICMD uses the phase generator PHG in a period (time t1 'to t2') excluding the control on period in the synchronous control mode, and generates a synthesized current Ia 'having a constant amplitude value (Isync). The current vector phase θs is changed at a predetermined rate. The combined current Ia ′ is a current obtained by vector addition of the d-axis (γ-axis) current Idc ^* and the q-axis (δ-axis) current Iqc ^* .

Thereafter, the current command computing unit ICMD converges the current vector phase θs of the combined current Ia ′ to zero in the axial error Δθ using the current phase controller IPHC in the control on period (time t2 ′ to t3 ′). Control in the direction. When the axis error Δθ converges to zero, switching to the position sensorless control mode (from time t3 ′) is performed, and in this mode, the d-axis (γ-axis) current Idc ^* is controlled to zero. .

  In the method of the comparative example, conceptually, for example, at time t2a in FIG. 10, the γ-axis current Iγ (synthesized current Ia ′ in the comparative example) is rotated while maintaining the amplitude value, and as a result, rotation By rotating the control axis (γδ axis) so that the q-axis component of the later combined current Ia ′ and the required torque current Iq0 become equal, the axis error Δθ is reduced. However, in the method of the comparative example, since the combined current Ia 'is rotated while maintaining the amplitude value, a large amount of unnecessary current (that is, d-axis current) which is not required for generation of torque flows. In particular, in the case of a light load or the like, the γ-axis current Iγ (synthesized current Ia ′) becomes much larger than the required torque current Iq0, so that the wasted current also increases.

  On the other hand, in the method of the first embodiment, unlike the case of FIG. 14B, as shown in FIG. 3, the amplitude value of the combined current Ia is decreased in the sensorless switching period (time t2 to t3). A system is adopted in which the amplitude value of the combined current Ia is brought close to the required amplitude value of the torque current Iq0. As a specific example, the amplitude value of the combined current Ia at time t2b in FIG. 10 is smaller than the amplitude value of the γ-axis current Iγ at time t2a. In response to this, the useless current (d-axis current) also decreases sequentially, and the γ-axis current (equal to the d-axis current) at the end of the sensorless switching period (time t3 in FIG. 3) Unlike (b), it is ideally zero.

  As described above, by using the method of the first embodiment, it is possible to reduce the axis error Δθ while suppressing unnecessary current before the switching to the position sensorless control mode is performed. As a result, high efficiency of the three-phase motor MT can be achieved. Further, by reducing the axis error Δθ, it is possible to suppress a switching shock such as a sudden jump of the current or a vibration of the rotational speed.

Second Embodiment
<< Configuration and operation of control mode switching control unit >>
FIG. 11 is a schematic view showing a configuration example of the control mode switching control unit of FIG. 1 in the motor control device according to Embodiment 2 of the present invention. The control mode switching control unit MDCTa shown in FIG. 11 is added with a coefficient calculation unit KCAL for calculating a predetermined coefficient Kg, in the configuration example shown in FIG. The coefficient calculation unit KCAL is controlled to be in the on state (active state) in the sensorless switching period in response to an instruction from the switching sequencer SEQ, and calculates a predetermined coefficient Kg according to the axis error Δθ.

  FIG. 12 is a schematic view showing a configuration example of the coefficient calculation unit in FIG. The coefficient calculation unit KCAL shown in FIG. 12 detects a time-series change of the axis error Δθ, and receives an output of the filter FLT and a filter FLT that smoothes the increase for the change in the increase direction. And a compensator PICkg for calculating a predetermined coefficient Kg by performing control. Specifically, in the filter FLT, a high-pass filter (in other words, a differentiator) HPF that detects a fluctuation value of the axis error Δθ in each control cycle and a fluctuation value detected by the high-pass filter HPF is in a positive direction (increasing direction). A low pass filter LPF is provided which outputs the result of smoothing the fluctuation value as a control value Δθ hpf for the case.

  The compensator PICkg calculates a predetermined coefficient Kg by performing PI control based on the equation (8) with the control value Δθhpf as an input. In equation (8), “Kp” is a proportional (P) constant, “Ki” is an integral (I) constant, and “s” is a Laplace operator. Also, in this example, the compensator PICkg performs PI control, but at least I control may be performed.

Kg = (Kp + Ki / s) × Δθhpf (8)
When such a coefficient calculation unit KCAL is used, when the axis error Δθ increases, the value of the predetermined coefficient Kg also increases according to the increase. On the other hand, when the axis error Δθ decreases, the value of the predetermined coefficient Kg is maintained.

<< Main effects of Embodiment 2 >>
As described above, by using the method of the second embodiment, the following effects can be obtained in addition to the various effects described in the first embodiment. First, the value of the torque current Iq0 calculated by the equation (5) may actually deviate from the value that is truly required. Specifically, for example, when a change in load torque occurs within the sensorless switching period, the value of the necessary torque current Iq0 also changes accordingly. The axis error calculator AERC estimates the axis error Δθ using the motor constants (resistance component Rs, q axis inductance component Lq) as shown in equations (1) to (3), but the motor constants If “ば ら” varies, “Δθ 0” in equation (5) may also be inaccurate.

  If, for example, the target value Itg calculated by the equation (6) is insufficient for the value that is truly required due to such a factor, the axial error Δθ does not converge (for example, increases) to zero, or In some cases, out of step may cause an uncontrollable situation. That is, for example, at time t2b in FIG. 10, a situation may occur in which the torque current (Iq0) actually required is larger than the q-axis component of the combined current Ia.

Further, in order to simplify the switching process from the synchronous control mode to the position sensorless control mode, it is desirable to increase the target rotational speed ω ^* even during the sensorless switching period, as shown in FIG. However, in this case, since the required torque current Iq0 increases in the sensorless switching period with acceleration, this may cause the target value Itg to be insufficient for the value that is truly required.

Therefore, when the method of the second embodiment is used, when the target value Itg is insufficient, the target value Itg can be increased by a necessary amount through the predetermined coefficient Kg. In other words, the target value Itg can be increased by the necessary amount so as to decrease the axis error Δθ. As a result, various effects described in the first embodiment can be obtained even when variations in motor constants, fluctuations in load torque, and the like occur. Furthermore, it becomes possible to increase the target rotational speed ω ^* during the sensorless switching period, and the processing can be simplified.

Third Embodiment
<< Configuration and operation of control mode switching control unit >>
FIG. 13 is a schematic diagram showing a configuration example in which the coefficient calculation unit of FIG. 12 is modified in the motor control device according to Embodiment 3 of the present invention. The coefficient operation unit KCALa shown in FIG. 13 has the input of the filter FLT from the axis error Δθ, which is the output of the axis error calculator AERC, compared to the configuration example of FIG. It is replaced by the induced voltage Eγ. Along with this, the compensator PICkg calculates a predetermined coefficient Kg by performing PI control based on the equation (9) with the control value Eγ_hpf from the filter FLT as an input.

Kg = (Kp + Ki / s) × Eγ_hpf (9)
Since the axis error Δθ and the γ-axis induction voltage Eγ are in a proportional relationship as shown in the equation (3) or as described in FIG. 6 and the like, such a configuration is used instead of the configuration example of FIG. It is also possible to use an example. By using the method of the third embodiment, the same effect as that of the second embodiment can be obtained.

  As mentioned above, although the invention made by the present inventor was concretely explained based on an embodiment, the present invention is not limited to the above-mentioned embodiment, and can be variously changed in the range which does not deviate from the gist. For example, the above-described embodiments are described in detail in order to explain the present invention in an easy-to-understand manner, and are not necessarily limited to those having all the described configurations. Also, part of the configuration of one embodiment can be replaced with the configuration of another embodiment, and the configuration of another embodiment can be added to the configuration of one embodiment. . In addition, with respect to a part of the configuration of each embodiment, it is possible to add, delete, and replace other configurations.

AERC axis error calculator AXCC 3 phase / 2 phase converter AXCR 2 phase / 3 phase converter BEOB induced voltage observer DCAL δ current operation unit Eγ, Eδ induction voltage FLT filter GCAL γ current generation unit IDG d axis current command generator INV Inverter ISEN Current sensor Ia Combined current Iγ ^* γ-axis target current Iδ ^* δ-axis target current KCAL coefficient calculation unit Kg coefficient MCD motor controller MDCT control mode switching control unit MEM holding unit MT 3-phase motor PHC phase calculator PICid d-axis current Controller PICiq q-axis current controller PICkg compensator PICrs speed controller PLLC PLL controller PWMG PWM signal generator PWMMD PWM modulator RSG speed command generator RT rotor SB subtractor SEL selector SEQ switch sequencer Δθ axis error θ times Phase ω ^* target rotational speed

Claims (13)

A motor control device for controlling a three-phase motor by vector control, comprising:
An axis error calculator which calculates an axis error between a dq axis which is a real axis of the three-phase motor and a γδ axis which is a control axis corresponding to the dq axis using a γ axis induced voltage of the three phase motor ,
A current detection unit that detects a γ-axis current and a δ-axis current based on the current flowing through the three-phase motor;
Each of the three-phase motors based on the respective errors between the γ-axis target current and the δ-axis target current, the γ-axis current from the current detection unit and the δ-axis current, and a rotational phase based on a predetermined rotational speed A PWM signal generator that generates a PWM (Pulse Width Modulation) signal for the phase
From the synchronous control mode in which the γ-axis target current is set to a predetermined value and the three-phase motor is driven while the predetermined rotational speed is increased by the open loop control, the predetermined rotational speed is controlled by the closed loop control A control mode switching control unit for switching to the position sensorless control mode for driving the three-phase motor in a switching period in the synchronous control mode;
Have
The control mode switching control unit is configured to:
The q-axis current is calculated based on the γ-axis target current at the start of the switching period and the axis error, and the q-axis current is “Iq0”, and a predetermined coefficient is “Kg” (Kg> 0). A first process of setting × (1 + Kg) to a target value,
Process A for acquiring the axis error from the axis error calculator, and the added value of the q axis component of the decreased γ axis target current and the q axis component of the δ axis target current by decreasing the γ axis target current Processing B to increase the δ-axis target current so that the target value becomes equal to the target value until the axis error in the processing A becomes equal to or less than a predetermined threshold value, and then to the position sensorless control mode A second process of switching
To perform
Motor controller. In the motor control device according to claim 1,
The control mode switching control unit decreases an amplitude value of a combined current of the γ-axis target current and the δ-axis target current in the second process.
Motor controller. In the motor control device according to claim 2,
In the second process, the control mode switching control unit sets the γ-axis target current to “Iγ”, the axis error in the process A to “Δθ”, and the δ-axis target current to “Iq0 × Calculated using an arithmetic expression of (1 + Kg) / cos (Δθ) −Iγ × tan (Δθ) ′ ′,
Motor controller. In the motor control device according to claim 1,
The control mode switching control unit
A filter that detects a time-series change in the axis error in the processing A or the γ-axis induced voltage used in the axis error calculator, and smoothes the increase for the change in the increase direction;
A compensator that receives the output of the filter and performs at least I (Integral) control to calculate the predetermined coefficient in the first process;
Have
Motor controller. In the motor control device according to claim 4,
The predetermined rotational speed is controlled to increase within the switching period,
Motor controller. In the motor control device according to claim 1, further,
A timing controller that generates a first rotational speed that causes the axis error to converge to zero;
A first selector which selects a second rotation speed as the target rotation speed or the first rotation speed, and outputs the predetermined rotation speed;
A speed controller generating a first δ-axis target current based on an error between the second rotation speed and the first rotation speed;
A second selector for selecting the first δ-axis target current or the second δ-axis target current;
Have
The PWM signal generation unit is configured to generate the first γ-axis target current and the δ-axis target current from the second selector, and the respective errors between the γ-axis current from the current detection unit and the δ-axis current. Generating the PWM signal for each phase of the three-phase motor based on a rotational phase based on the predetermined rotational speed from a selector;
The control mode switching control unit causes the first selector and the second selector to select the second rotational speed and the second δ-axis target current, respectively, during the period of the synchronous control mode. The second rotation speed is increased, and during the position sensorless control mode, the first selector and the second selector select the first rotation speed and the first δ-axis target current, respectively. Control the timing controller and the speed controller in the off state,
Motor controller. Three-phase motor,
An inverter for applying a drive voltage based on a PWM (Pulse Width Modulation) signal to each phase of the three-phase motor;
A motor control device for controlling the three-phase motor via the inverter by vector control;
A motor system having
The motor control device
An axis error calculator which calculates an axis error between a dq axis which is a real axis of the three-phase motor and a γδ axis which is a control axis corresponding to the dq axis using a γ axis induced voltage of the three phase motor ,
A current detection unit that detects a γ-axis current and a δ-axis current based on the current flowing through the three-phase motor;
Each of the three-phase motors based on the respective errors between the γ-axis target current and the δ-axis target current, the γ-axis current from the current detection unit and the δ-axis current, and a rotational phase based on a predetermined rotational speed A PWM signal generator for generating a PWM signal for the phase;
From the synchronous control mode in which the γ-axis target current is set to a predetermined value and the three-phase motor is driven while the predetermined rotational speed is increased by the open loop control, the predetermined rotational speed is controlled by the closed loop control A control mode switching control unit for switching to the position sensorless control mode for driving the three-phase motor in a switching period in the synchronous control mode;
Have
The control mode switching control unit is configured to:
The q-axis current is calculated based on the γ-axis target current at the start of the switching period and the axis error, and the q-axis current is “Iq0”, and a predetermined coefficient is “Kg” (Kg> 0). A first process of setting × (1 + Kg) to a target value,
Process A for acquiring the axis error from the axis error calculator, and the added value of the q axis component of the decreased γ axis target current and the q axis component of the δ axis target current by decreasing the γ axis target current Processing B to increase the δ-axis target current so that the target value becomes equal to the target value until the axis error in the processing A becomes equal to or less than a predetermined threshold value, and then to the position sensorless control mode A second process of switching
To perform
Motor system. In the motor system according to claim 7,
The control mode switching control unit decreases an amplitude value of a combined current of the γ-axis target current and the δ-axis target current in the second process.
Motor system. In the motor system according to claim 8,
In the second process, the control mode switching control unit sets the γ-axis target current to “Iγ”, the axis error in the process A to “Δθ”, and the δ-axis target current to “Iq0 × Calculated using an arithmetic expression of (1 + Kg) / cos (Δθ) −Iγ × tan (Δθ) ′ ′,
Motor system. In the motor system according to claim 7,
The control mode switching control unit
A filter that detects a time-series change in the axis error in the processing A or the γ-axis induced voltage used in the axis error calculator, and smoothes the increase for the change in the increase direction;
A compensator that receives the output of the filter and performs at least I (Integral) control to calculate the predetermined coefficient in the first process;
Have
Motor system. The motor system according to claim 10,
The predetermined rotational speed is controlled to increase within the switching period,
Motor system. In the motor system according to claim 7,
The motor control device further comprises:
A timing controller that generates a first rotational speed that causes the axis error to converge to zero;
A first selector which selects a second rotation speed as the target rotation speed or the first rotation speed, and outputs the predetermined rotation speed;
A speed controller generating a first δ-axis target current based on an error between the second rotation speed and the first rotation speed;
A second selector for selecting the first δ-axis target current or the second δ-axis target current;
Have
The PWM signal generation unit is configured to generate the first γ-axis target current and the δ-axis target current from the second selector, and the respective errors between the γ-axis current from the current detection unit and the δ-axis current. Generating the PWM signal for each phase of the three-phase motor based on a rotational phase based on the predetermined rotational speed from a selector;
The control mode switching control unit causes the first selector and the second selector to select the second rotational speed and the second δ-axis target current, respectively, during the period of the synchronous control mode. The second rotation speed is increased, and during the position sensorless control mode, the first selector and the second selector select the first rotation speed and the first δ-axis target current, respectively. Control the timing controller and the speed controller in the off state,
Motor system. In the motor system according to claim 7,
The motor control device comprises a microcontroller
Motor system.

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