JP2009017706A - Motor controller and motor control method - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide a motor controller which can perform stable current control even when the parameters of a three-phase motor fluctuate. <P>SOLUTION: A motor driver 100, which converts voltage commands for d axis in the direction of a field generated by a permanent magnet arranged in the rotor of a three-phase motor 10 and for q axis orthogonal to the d axis into three-phase voltage commands of each phase of three phases and drives the three-phase motor 10 by control based on the three-phase voltage command, is equipped with a PWM control unit 6 which controls the motor current of each phase of three phases by PWM control, and a sliding mode control unit 5 which determines the equivalent input at the time of making it follow a desired standard model and the control input for restricting it to a switching plane and performs sliding mode control synchronously with the cycle of PWM control. <P>COPYRIGHT: (C)2009,JPO&INPIT

Description

  The present invention relates to a motor control device capable of performing stable current control when driving a three-phase motor, and such a motor control method.

  Conventionally, in drive control of a three-phase motor, the motor current is coordinate-converted into a vector component of the d-axis which is the direction of the magnetic field generated by the permanent magnet of the motor rotor and the q-axis vector component orthogonal to the d-axis. There are some which perform rotation control of a phase motor (for example, nonpatent literature 1). In the drive control of a three-phase motor similar to Non-Patent Document 1, the rotation control is stabilized by performing PI control based on the result of coordinate conversion.

  However, in PI control, there is a problem in the control response time, and a certain time (time constant) is required to control the target value. If this time constant is large, the response performance when there is a disturbance deteriorates, so that there is a problem that it cannot react quickly to the disturbance and cannot be immediately returned to the original target value. Therefore, when PI control is used for rotation control of a three-phase motor, there is a problem that current control is not stable when various characteristics (parameters) of the three-phase motor fluctuate, for example.

Sugimoto Hidehiko, “Theory and Design of AC Servo Systems, From Basics to Software Servo”, General Electronic Publishing, p. 86-92

  In view of the above problems, an object of the present invention is to provide a motor control device capable of performing stable current control even when parameters of a three-phase motor fluctuate.

  In order to achieve the above object, the motor control device according to the present invention is characterized by a d-axis that is a direction of a magnetic field generated by a permanent magnet disposed in a rotor of a three-phase motor and a q-axis that is orthogonal to the d-axis. Is converted into a three-phase voltage command value for each of the three phases, the three-phase motor is driven by the control based on the three-phase voltage command value, and the motor current of each three-phase is controlled by the PWM control A PWM control unit that determines an equivalent input for following the intended reference model and a control input for constraining to the switching surface, and the motor current of each of the three phases in synchronization with the period of the PWM control And a sliding mode control unit that controls the sliding mode by sliding mode control.

  With such a configuration, for example, even when the output of the controlled object fluctuates, the sliding mode control unit having a certain tolerance for the output of the controlled object can quickly change the motor current as the input signal. Can be determined. Specifically, by determining the dead zone at the time of switching from the equivalent input and performing the PWM control including the switching plane and the dead zone, the instantaneous value of the motor current can always be constrained near the reference model. Therefore, stable current control can be performed for each of the three phases.

  In the motor control device, the rise of the PWM control signal used for the PWM control is determined according to a signal output from the PWM control unit at a constant period, and the fall of the PWM control signal is It is preferable that it is determined according to a signal output from the sliding mode control unit.

  With such a configuration, PWM control can be performed by synchronizing the rise of the PWM control signal performed by the PWM control unit and the fall of the PWM control signal performed by the sliding mode control unit. Therefore, the cycle by PWM control and the cycle by sliding mode control do not become asynchronous, so that stable current control can be performed.

  Further, the motor control device outputs a voltage command value for the d-axis, which is a direction of a magnetic field generated by a permanent magnet disposed in a rotor of a three-phase motor, and a q-axis voltage command value orthogonal to the d-axis for each of the three phases. A motor control method for a motor drive device that converts to a three-phase voltage command value and drives the three-phase motor by PWM control based on the three-phase voltage command value is also within the scope of rights. The motor current of each phase is controlled by PWM control, an equivalent input to be input to follow the intended reference model and a control input to be constrained by the switching surface are determined, and in synchronization with the PWM control cycle Sliding mode control is performed.

  The motor control method for the motor control device configured as described above is not substantially different from the motor control device as the object of the present invention described above in terms of the substantial characteristic configuration, and the above-described operational effects can be obtained. Is possible.

  Embodiments of the present invention will be described below with reference to the drawings. FIG. 1 is a schematic diagram showing a configuration of a motor control device 100 of the present invention. The motor control device 100 includes a target current setting unit 1, an integration control unit 2, a first 2-phase / 3-phase conversion unit 3, a second 2-phase / 3-phase conversion unit 4, a sliding mode control unit 5, and a PWM control. Unit 6, inverter 7, three-phase / two-phase conversion unit 8, current calculation unit 9, and three-phase motor 10 (details will be described later).

  FIG. 2 is a diagram illustrating the configuration of the PWM control unit 6, the inverter 7, and the three-phase motor 10. Although not shown, the three-phase motor 10 includes a rotor including a permanent magnet and a stator that generates a magnetic field for applying a rotational force to the rotor. This stator includes three-phase stator coils 10u, 10v, and 10w of U phase, V phase, and W phase. One end of each stator coil is connected in common at an electrically neutral point and Y-connected. The other end of each stator coil is connected to the inverter 7.

  As shown in FIG. 2, the inverter 7 includes high-side transistors Q1, Q3, Q5 connected to the positive voltage side of the power source 20, and low-side transistors Q2, Q4 connected to the negative voltage side of the power source 20. Q6 and a total of six transistors Q1 to Q6. For example, when only the transistor Q1 and the transistor Q4 are simultaneously turned on, a current flows from the power supply 20 to the second power supply line 22 via the first power supply line 21, the transistor Q1, the stator coil 10v, the stator coil 10w, and the transistor Q4. On the other hand, when only the transistor Q3 and the transistor Q2 are simultaneously turned on, a current flows from the power supply 20 to the second power supply line 22 via the first power supply line 21, the transistor Q3, the stator coil 10w, the stator coil 10v, and the transistor Q2.

  The direction of the current flowing through the stator coil 10v and the stator coil 10w differs between when only the transistor Q1 and the transistor Q4 are turned on and when only the transistor Q3 and the transistor Q2 are turned on. Therefore, an electromagnetic force corresponding to the direction in which the current flows acts on each stator coil, and an attractive force and a repulsive force are generated between the electromagnetic force and a permanent magnet provided in the rotor. Therefore, the rotor can obtain rotational force by sequentially turning on the upper and lower pair transistors formed by the high-side transistor and the low-side transistor selected from the transistors Q1 to Q6.

  The transistors Q1 to Q6 are provided with diodes D1 to D6 so that the cathode terminal is connected to the collector terminal and the anode terminal is connected to the emitter terminal. Here, energy is stored in each stator coil during energization, but these diodes D1 to D6 are applied to peripheral components by back electromotive force generated due to the energy when each stator coil is de-energized. It is provided in order to prevent adverse effects. A series of controls for the transistors Q1 to Q6 are performed by the PWM controller 6 (details will be described later).

  Returning to FIG. 1, the target current setting unit 1 sets the target current from the total torque required to rotate the three-phase motor 10. The target current set by the target current setting unit 1 is transmitted to the integration control unit 2 and the second 2-phase / 3-phase conversion unit 4. Is also transmitted as a feedback signal.

  Here, the motor control apparatus 100 uses the motor currents iu, iv, and iw as vectors of the d axis that is the direction of the magnetic field generated by the permanent magnets of the rotor of the three-phase motor 10 and the q axis that is orthogonal to the d axis. Coordinate conversion is performed on the components Id and Iq to control the rotation of the three-phase motor 10. FIG. 3 is a diagram showing the principle of this coordinate transformation. The three-phase motor 10 shown in FIG. 3 includes a rotor 10r having a two-pole permanent magnet m, and the rotation angle of the rotor 10r and the electrical angle θ coincide with each other. FIG. 3A is a diagram showing the relationship between the three-phase alternating current waveform and the electrical angle θ, and FIG. 3B is the positional relationship between the rotor 10r and the stator 10s at time t1 in FIG. It is a figure which shows the electric current vector before and behind coordinate transformation. FIG. 3B shows the electrical angle θ that is the magnetic pole position of the rotor 10r with reference to the U-phase magnetic pole position of the stator 10s.

  As shown in FIG. 3B, the direction of the magnetic field generated by the permanent magnet m is defined as the d axis, and the direction orthogonal to the d axis is defined as the q axis. As shown in FIG. 3A, torque is generated by flowing three-phase alternating currents iu, iv, iw through the stator coils 10u, 10v, 10w according to the magnetic pole position of the rotor 10r. The vector ia (Ia) indicating the sum of the armature currents at the electrical angle θ at time t1 in FIG. 3A is zero in the W-phase current iw from FIG. It is a vector sum with the phase current iv. When the current vector ia at the electrical angle θ is decomposed with respect to the d axis and the q axis, a d axis current Id and a q axis current Iq are obtained. In this way, the three-phase motor currents iu, iv, iw are coordinate-converted into the d-axis current Id and the q-axis current Iq.

  Here, in particular, in the permanent magnet embedded type synchronous motor, the inductance viewed from the stator coils 10u, 10v, and 10w changes depending on the relationship with the rotor 10r, that is, the relationship with the magnetic pole position. In the d-axis direction, which is the direction of the magnetic pole, the magnetic path is obstructed because it has a magnetic resistance proportional to the inverse of the permeability of the permanent magnet. On the other hand, in the q-axis direction, since the magnetic material such as silicon steel having a high permeability is passed, the value of the magnetic resistance is significantly smaller than that of the permanent magnet, and the magnetic path is not easily disturbed. For this reason, the q-axis inductance Lq is larger than the d-axis inductance Ld. Since the d-axis and the q-axis change in relation to the magnetic pole position when viewed from the stator coils 10u, 10v, and 10w, the inductance viewed from the stator coils 10u, 10v, and 10w changes.

  Therefore, in addition to the magnet torque (main torque) by the permanent magnet, reluctance torque is also generated due to the difference between the q-axis inductance Lq and the d-axis inductance Ld. When reluctance torque is not actively used, such as a surface magnet type synchronous motor, it is efficient to perform control with Id = 0. However, when using reluctance torque in a permanent magnet embedded synchronous motor or the like, it is more efficient to perform control with Id ≠ 0. In the permanent magnet embedded type synchronous motor, the operating point at which the maximum efficiency is obtained varies depending on the current phase angle β between the d-axis current Id and the q-axis current Iq shown in FIG. 4 (see equation (1)).

  The total torque T of the three-phase motor 10 is Pn: pole pair number, ψa: armature linkage flux, ia: armature current, Ld: d-axis inductance, Lq: q-axis inductance, β: current phase angle. It is represented by the torque equation shown in equation (2).

  In the formula (2), the first term in the braces indicates the magnet torque, and the second term indicates the reluctance torque. Moreover, since it is clear from FIG. 4 that the following equations (3) to (5) are obtained, the torque equation of the equation (2) can also be expressed as the following equation (6).

  As described above, the armature current Ia includes the d-axis current Id and the q-axis current Iq. Therefore, the torque equations shown in the equations (2) and (6) represent the torque of the three-phase motor 10 using the flux linkage, the d-axis and q-axis inductances, and the d-axis and q-axis currents. It can be said that it is an expression.

  Returning to FIG. 1, the integration control unit 2 is based on the target current set by the target current setting unit 1 and the motor current flowing through the three-phase motor 10 detected by the current calculation unit 9. 8 calculates a d-axis current command value Idr and a q-axis current command value Iqr from the d-axis current Id and the q-axis current Iq obtained by performing coordinate conversion. For example, the torque equation shown in the above equation (2) can be transformed into the equation of the armature current Ia. The integral control unit 2 can calculate the current command values Idr and Iqr by substituting the target torque and other parameters to solve the deformed torque equation and performing vector decomposition with the phase angle β. Alternatively, it is naturally possible to calculate the current command values Idr and Iqr from the equation (6). Further, the integration control unit 2 calculates the d-axis voltage command value Vdr and the q-axis voltage command value Vqr from the current command values Idr and Iqr calculated by the target current setting unit 1 based on the voltage equation. . The voltage equations representing the d-axis voltage Vd and the q-axis voltage Vq are: ψa: armature linkage flux, ω: angular velocity, Id: d-axis current, Iq: q-axis current, Ld: d-axis inductance, Lq: The q-axis inductance, Ra: armature resistance, p: differential operator is expressed as the following equation (7).

  Equation (7) is a voltage for driving the three-phase motor 10 using the interlinkage magnetic flux, the impedance of the stator coil of the three-phase motor 10 including the d-axis and q-axis inductances, and the d-axis and q-axis currents. It is clear that the voltage equation represents The integral control unit 2 calculates the d-axis voltage Vd and the q-axis voltage Vq by substituting the current command values Idr and Iqr and other parameters into the voltage equation shown in the equation (7). The calculated d-axis voltage Vd and q-axis voltage Vq are output as a d-axis voltage command value Vdr and a q-axis voltage command value Vqr.

  The first two-phase / three-phase conversion unit 3 converts the d-axis voltage command value Vdr and the q-axis voltage command value Vqr calculated by the integration control unit 2 by reversing the coordinate conversion described above. It is converted into a three-phase voltage command value vu, vv, vw. Further, the second 2-phase / 3-phase converter 4 also converts the target current calculated by the target current setting unit 1 in the reverse of the coordinate conversion described above, so that the three-phase voltage command values vu, vv, Convert to vw. Since this reverse conversion is the reverse conversion of the coordinate conversion described above with reference to FIGS. 3 and 4, detailed description of the conversion method is omitted.

  The calculation results obtained by the first two-phase / three-phase converter 3 and the second two-phase / three-phase converter 4 are the sliding mode control together with the current flowing through the three-phase motor 10 detected by the current calculator 9. Input to part 5. The sliding mode control unit 5 determines an equivalent input in each phase based on the input calculation result, and determines a dead zone in consideration of a chattering frequency synchronized with the PWM control cycle from the equivalent input. When the control of the three-phase motor deviates from the dead zone centering on the switching surface in the sliding mode control, the output of the PWM control of each phase is switched.

This switching will be briefly described with reference to FIG. A broken line (hereinafter referred to as A line) shown by A in FIG. 5 is a line indicating intended control as the motor control apparatus 100, that is, σ _u (x _s ) = 0. The motor control device 100 tries to perform motor control along the A line in order to obtain a necessary torque calculated by each functional unit included in the motor control unit 100 such as the target current setting unit 1 (for example, a point), in particular, the control is affected by various characteristics of the three-phase motor 10, and may deviate from the intended control (control on the A line) (for example, b point).

Here, in the sliding mode control, a dead zone having a predetermined width Δσ _u is determined in advance. In FIG. 5, a dead zone determined by the B line and the C line having a width of Δσ _u from the A line is determined. The B line and the C line are switching surfaces that become a MAX threshold value and a MIN threshold value, respectively, indicating a dead zone. As described above, when the motor control deviates from the motor control on the A line and becomes the motor control indicated on the C line, the sliding mode control unit 5 performs control so that the motor control falls within the dead zone indicated by the B line. (For example, point c).

As the motor control is performed again, when the result of the motor control becomes control along the C line (for example, point d), the sliding mode control unit 5 is placed within the dead zone (for example, point e). The motor is controlled along the line. As described above, the sliding mode control unit 5 performs the PWM control of the three-phase motor 10 so that the desired motor control can be performed within the dead zone having the predetermined width Δσ _u, and the PWM control signal is particularly generated by the sliding mode control. Controls the falling edge.

  As described above, in the sliding mode control, the motor control is performed within the dead zone. However, when control is attempted to deviate from the dead zone, the control cannot be performed so that the cycle is constant because the control is switched on the switching surface. However, in motor control, more stable torque can be obtained by rotating the three-phase motor 10 at a constant period. Therefore, the PWM controller 6 controls the rise of the PWM control signal. As described above, the motor control apparatus 100 has various characteristics of the three-phase motor 10 because the PWM control unit 6 controls the rise of the PWM control signal and the sliding mode control unit 5 controls the fall of the PWM control signal. Even if the fluctuation occurs, stable motor control can be performed.

Next, generation of a PWM control signal performed by the motor control apparatus 100 will be described with reference to the drawings. FIG. 6 is a system block diagram for explaining the embodiment. As shown in FIG. 6, the first output x ₁ is obtained from the input v, and the second output x ₂ , ie, the output i is obtained from the first output x ₁ . Here, L: average inductance of d-axis and q-axis, r: coil resistance per phase, τ: time constant of circuit and sensor.

From the system block diagram of FIG. 6, this system can be expressed as shown in equation (8).

Here, the switching function S indicating the switching surface in the sliding mode control is determined as shown in Equation (9) so as to have an ideal response.

The equivalent input in the sliding mode control is expressed as in equation (10).

ここで、ｕ_max：バッテリ電圧Ｖｂ、ｕ_min：ダイオードの順方向電圧Ｖｆである。 When the switching frequency is finite, the state of the system cannot reach the switching surface, but can reach the vicinity of the switching surface. In this case, if the limit values u _lim are u _min and u _max , the equation (11) is obtained.

Here, u _{max is the} battery voltage Vb, u _{min is} the forward voltage Vf of the diode.

よって、（１１）及び（１２）式より、

が求まる。 On the other hand, in the case of ideal sliding mode control, if u = u _equ , the equation (12) is obtained.

Therefore, from the equations (11) and (12),

Is obtained.

The deviation of σ _u (x) is shown in FIG. As shown in FIG. 7, σ _u (x) varies within the dead zone ± Δσ _u due to the switching delay. When u _lim = u _max , the time corresponding to this is given by equation (14).

一方、チャタリング周波数ｆｃは（１６）式のようになる。

During this time control, u _equ is assumed to be a constant. Further, when u _lim = u _min , the time corresponding to this is given by equation (15).

On the other hand, the chattering frequency fc is expressed by equation (16).

よって、不感帯Δσ_uは（１８）式のように表される。

From the equations (14) to (16), the equation (17) is obtained.

Therefore, the dead zone Δσ _u is expressed as in equation (18).

The setting of the switching timing of the PWM control signal performed by the sliding mode control unit 5 using the dead zone Δσ _u will be described using a flowchart. FIG. 8 shows the calculation of the switching timing in the U phase. In addition, since it is possible to calculate similarly about V phase and W phase, description is abbreviate | omitted.

  First, the coil current of the three-phase motor 10 calculated by the current calculation unit 9 is input to the sliding mode control unit 5 (step # 01). Further, the three-phase command current obtained by converting the target current set by the target current setting unit 1 by the first two-phase / three-phase conversion unit 3 is also input to the sliding mode control unit 5 (step # 02). The sliding mode control unit 5 calculates a deviation from the coil current and the command current (step # 03).

  Subsequently, calculation of equivalent input, dead zone, and switching surface in sliding mode control is performed (steps # 04 to # 06). Here, KI in Step # 06 indicates a gain in the integral control unit 2 for compensating for the deviation. Then, a control input is calculated from the calculated dead zone and the switching surface (step # 07). If the control input obtained in step # 07 is greater than 0 (step # 08: Yes), the set value A is set to 1 (step # 09). On the other hand, when the obtained control input is 0 or less (step # 08: No), the set value A is set to 0 (step # 10).

ここで、Ｚ（ｎ−１）：前回の出力Ｚである。（１９）式で求められたＺがＰＷＭ制御信号の立ち下がりのトリガとなり、出力Ｚが１となった場合にトリガ信号が出力される。 If the switching surface obtained in step # 06 is greater than 0 (step # 11: Yes), the set value B is set to 1 (step # 12). On the other hand, when the switching surface is 0 or less (step # 11: No), the set value B is set to 0 (step 13). Using the set value A and the set value B set in this way, the logical operation of the output Z output from the calculation unit included in the sliding mode control unit 5 is performed (step # 14) (see formula (19)). ).

Here, Z (n-1) is the previous output Z. Z obtained by the equation (19) becomes a trigger for falling of the PWM control signal, and when the output Z becomes 1, a trigger signal is output.

  A PWM control signal is calculated from the logic circuit shown in FIG. 9 using the output Z, the coil current, the command current, the d-axis and q-axis inductance average values obtained by the flowchart shown in FIG. Specifically, in the logic circuit shown in FIG. 9, the pulse generator 5a outputs a pulse having an amplitude of 10, a frequency of 10 kHz, and a duty of 50%. Therefore, 1 is output from the pulse generator 5a every 100 μsec. Thus, if the cycle of the PWM control unit is set to 10 kHz, the signal output from the sliding mode control unit 5 and the PWM control signal of the PWM control unit 6 can be synchronized.

In the above, τ: 5 × 10 ⁻⁶ [sec], L = 3.27 × 10 ⁻⁴ [H], r = 0.15 [Ω], Vb = 200 [V], Vf = 2.8 [V] ] FIG. 10 shows an example of a three-phase current response when KI = 3.9478 × 10 ⁷ . As shown in FIG. 10, according to the motor control apparatus 100, stable current control can be performed.

[Other Embodiments]
In the above embodiment, the frequency of the PWM control signal has been described as 10 kHz, but the present invention is not limited to this. If the period of the signal generated from the pulse generator 5a included in the sliding mode control unit 5 is synchronized with the rising period of the PWM control signal output from the PWM control unit 6, the sliding mode control according to the present invention stabilizes. Of course, it is possible to control the three-phase motor 10 by current control.

The figure which shows the structure of the motor control device typically The figure which shows the structure of a PWM control part, an inverter, and a three-phase motor Diagram showing the principle of coordinate transformation Vector diagram explaining the phase angle of the armature current Diagram showing the locus of sliding mode control Diagram showing an example of system block diagram The figure which shows the deviation of the locus of sliding mode control Flowchart for computing sliding mode control output Logic circuit that calculates the output of the sliding mode controller Diagram showing an example of three-phase current response

Explanation of symbols

1: target current setting unit 2: integration control unit 3: first 2-phase / 3-phase conversion unit 4: second 2-phase / 3-phase conversion unit 5: sliding mode control unit 6: PWM control unit 7: inverter 8 : 3-phase / 2-phase converter 9: Current calculator 10: 3-phase motor

Claims (3)

The voltage command value of the d axis that is the direction of the magnetic field generated by the permanent magnet disposed in the rotor of the three phase motor and the q axis that is orthogonal to the d axis is converted into a three phase voltage command value for each of the three phases. In the motor drive device that drives the three-phase motor by control based on the three-phase voltage command value,
A PWM controller that controls the motor current of each of the three phases by PWM control;
An equivalent input for following the desired reference model and a control input for constraining to the switching surface are determined, and the motor current of each phase of the three phases is controlled by sliding mode control in synchronization with the period of the PWM control. And a sliding mode control unit. The rise of the PWM control signal used for the PWM control is determined according to a signal output from the PWM control unit at a constant period, and
The motor control device according to claim 1, wherein the falling edge of the PWM control signal is determined according to a signal output from the sliding mode control unit. The voltage command value of the d axis that is the direction of the magnetic field generated by the permanent magnet disposed in the rotor of the three phase motor and the q axis that is orthogonal to the d axis is converted into a three phase voltage command value for each of the three phases. In the motor control method for a motor driving device that drives the three-phase motor by PWM control based on the three-phase voltage command value,
Control the motor current of each of the three phases by PWM control,
A motor control method for determining an equivalent input as an input to follow an intended reference model and a control input for constraining to a switching surface, and performing a sliding mode control in synchronization with a period of the PWM control.

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