CN103236814A - Fractional integral sliding mode-based speed control method and device for permanent magnet synchronous motor - Google Patents

Abstract

The invention discloses a speed control method of a permanent magnet synchronous motor based on a fractional integral sliding mode, which includes constructing a fractional integral sliding mode surface S, and designing a fractional integral sliding mode speed controller for speed adjustment. The invention also designs A permanent magnet synchronous motor speed control device based on fractional integral sliding mode; the permanent magnet synchronous motor speed control method and device based on fractional integral sliding mode designed by the present invention can be used in the presence of parameter perturbation in the permanent magnet synchronous motor control system 1. In the case of load disturbance, realize high-performance speed following control of the motor, and overcome the integral saturation effect and transient performance degradation caused by large initial errors or actuator saturation in the existing integral sliding mode control.

Description

Method and device for speed control of permanent magnet synchronous motor based on fractional integral sliding mode

technical field

The invention relates to a method and device for controlling the speed of a permanent magnet synchronous motor based on a fractional-order integral sliding mode, and belongs to the technical field of AC servo motor control.

Background technique

Permanent magnet synchronous motor has the advantages of compact structure, high power density, high energy conversion efficiency, wide speed range, light weight, etc., and has been widely used in industrial, civil, military and other fields.

Since the permanent magnet synchronous motor is a typical nonlinear, multivariable coupling system, its performance is affected by uncertainties such as external load disturbance, internal parameter changes, unmodeled objects, and nonlinear dynamic characteristics. In order to obtain a good dynamic response, some robust control strategies such as nonlinear control, adaptive control, H _∞ control, sliding mode control, etc. have been introduced into the research of AC servo system. Among them, the sliding mode control has attracted the attention of researchers because of its strong robustness, fast response, and simple physical implementation.

Sliding mode control is a variable structure control method. When the system phase trajectory moves on the designed sliding mode surface, it has strong robustness to uncertain items of system parameters and external disturbances. Since the mid-1990s, there have been researches on applying sliding mode variable structure control to the control of AC servo system. In order to further solve the problem of steady-state error in the traditional sliding mode control, an integral term can be introduced in the design of the sliding mode surface. Adding the integral term to the sliding surface can reduce the steady-state error of the system, effectively weaken the chattering effect, and enhance the stability of the controller.

For the speed control of permanent magnet synchronous motor, the integral sliding mode surface S designed in the prior art is defined as follows:

$\mathrm{<span\; class="notranslate">\; S</span>}<span\; class="notranslate">\; =</span>\mathrm{<span\; class="notranslate">\; e</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; t</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; +</span>{\mathrm{<span\; class="notranslate">\; <figure-callout\; id="1"\; label="C"\; filenames="130427101550.png,130427101610.png"\; state="\{\{state\}\}">C</figure-callout></span>}}_{\mathrm{<span\; class="notranslate">\; 1</span>}}{<span\; class="notranslate">\; \&Integral;</span>}_{\mathrm{<span\; class="notranslate">\; 0</span>}}^{\mathrm{<span\; class="notranslate">\; t</span>}}\mathrm{<span\; class="notranslate">\; e</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; \&tau;</span>}<span\; class="notranslate">\; )</span>\mathrm{<span\; class="notranslate">\; d\&tau;</span>}<span\; class="notranslate">\; ,</span>$

It can be seen that the integral sliding mode surface S designed in the prior art is based on the integer-order integral of the velocity error, so the obtained integral sliding mode controller is also of integer order, which will lead to integral saturation when the initial error is large or the actuator is saturated. effects and transient performance degradation.

In the present invention, based on the fractional calculus of the state variables, a nonlinear fractional integral sliding mode surface is constructed, and a new fractional integral sliding mode controller is designed, which is applied to the speed following control system of the permanent magnet synchronous motor , while maintaining the traditional integer-order integral sliding mode controller, it effectively overcomes the problems of integral saturation effect and transient performance degradation.

Contents of the invention

The technical problem to be solved by the present invention is to provide a method that can realize high-performance speed following control of the motor under the condition of parameter perturbation and load disturbance in the permanent magnet synchronous motor control system, and overcome the large A permanent magnet synchronous motor speed control method and device based on a fractional integral sliding mode based on the integral saturation effect caused by the initial error of the initial error or the saturation of the actuator and the transient performance degradation.

The present invention adopts the following technical solutions to solve the above-mentioned technical problems: the present invention designs a permanent magnet synchronous motor speed control method based on fractional integral sliding mode, including the following specific steps:

Step (1): Collect the three-phase currents i _a , i _b , and i _c of the permanent magnet synchronous motor, and perform Clark transformation to convert them into equivalent currents i _a and i _β in the α-β coordinate system;

Step (2): Obtain the actual running speed ω and rotor position θ of the permanent magnet synchronous motor through the position sensor;

Step (3): According to the motor rotor position θ in step (2), perform PARK transformation on the motor rotor position θ and the equivalent current i _a and i _β in the α-β coordinate system in step (1) to generate dq coordinates The actual output current values i _d and i _q under the system;

Step (4): Use the difference e(t) between the actual running speed ω of the motor in step (2) and the given speed ω _ref to construct the fractional-order integral sliding mode surface S;

；Step (5): According to the fractional-order integral sliding mode surface S in step (4), design a fractional-order integral sliding-mode speed controller to adjust the speed, so as to obtain the given value of the q-axis current in the dq coordinate system

;

与q轴实际输出电流值i_q比较后的差值进行q轴电流调节处理，产生q轴电压输出值u_q，对d轴电流给定值与d轴实际输出电流值i_d比较后的差值进行d轴电流调节处理，产生d轴电压输出值u_d，将上述d-q坐标系下的电压输出值u_d和u_q进行PARK逆变换处理，从而得到α-β坐标系下的等效电压控制给定值u_α和u_β，对所述等效电压控制给定值u_α和u_β进行空间矢量脉宽调制，产生PWM信号，利用所述PWM信号控制三相逆变器产生三相电压信号，并利用所述三相电压信号控制永磁同步电机运行。Step (6): Set the value of the q-axis current

The difference after comparison with the actual output current value i _q of the q-axis is adjusted for the q-axis current to generate the q-axis voltage output value u _q , and the given value of the d-axis current The difference between the d-axis actual output current value i _d is compared with the d-axis current adjustment process to generate the d-axis voltage output value u _d , and the above-mentioned voltage output values u _d and u _q in the dq coordinate system are subjected to PARK inverse transformation processing , so as to obtain the equivalent voltage control given values u _α and u _β in the α-β coordinate system, and perform space vector pulse width modulation on the equivalent voltage control given values u _α and u _β to generate a PWM signal, using The PWM signal controls the three-phase inverter to generate a three-phase voltage signal, and uses the three-phase voltage signal to control the operation of the permanent magnet synchronous motor.

As an optimization method of the present invention: the Clark transformation in the step (1) is expressed as the following matrix:

$\left[\begin{array}{c}{\mathrm{<span\; class="notranslate">\; i</span>}}_{\mathrm{<span\; class="notranslate">\; \&alpha;</span>}}\\ {\mathrm{<span\; class="notranslate">\; i</span>}}_{\mathrm{<span\; class="notranslate">\; \&beta;</span>}}\end{array}\right]<span\; class="notranslate">\; =</span>\frac{\mathrm{<span\; class="notranslate">\; 2</span>}}{\mathrm{<span\; class="notranslate">\; 3</span>}}\left[\begin{array}{ccc}\mathrm{<span\; class="notranslate">\; 1</span>}& <span\; class="notranslate">\; -</span>\frac{\mathrm{<span\; class="notranslate">\; 1</span>}}{\mathrm{<span\; class="notranslate">\; 2</span>}}& <span\; class="notranslate">\; -</span>\frac{\mathrm{<span\; class="notranslate">\; 1</span>}}{\mathrm{<span\; class="notranslate">\; 2</span>}}\\ \mathrm{<span\; class="notranslate">\; 0</span>}& \frac{\sqrt{\mathrm{<span\; class="notranslate">\; 3</span>}}}{\mathrm{<span\; class="notranslate">\; 2</span>}}& <span\; class="notranslate">\; -</span>\frac{\sqrt{\mathrm{<span\; class="notranslate">\; 3</span>}}}{\mathrm{<span\; class="notranslate">\; 2</span>}}\end{array}\right]\left[\begin{array}{c}{\mathrm{<span\; class="notranslate">\; i</span>}}_{\mathrm{<span\; class="notranslate">\; a</span>}}\\ {\mathrm{<span\; class="notranslate">\; i</span>}}_{\mathrm{<span\; class="notranslate">\; b</span>}}\\ {\mathrm{<span\; class="notranslate">\; i</span>}}_{\mathrm{<span\; class="notranslate">\; c</span>}}\end{array}\right]<span\; class="notranslate">\; .</span>$

As an optimization method of the present invention: the PARK transformation in the step (3) is expressed as the following matrix:

$\left[\begin{array}{c}{\mathrm{<span\; class="notranslate">\; i</span>}}_{\mathrm{<span\; class="notranslate">\; d</span>}}\\ {\mathrm{<span\; class="notranslate">\; i</span>}}_{\mathrm{<span\; class="notranslate">\; q</span>}}\end{array}\right]<span\; class="notranslate">\; =</span>\left[\begin{array}{cc}\mathrm{<span\; class="notranslate">\; cos</span>}\mathrm{<span\; class="notranslate">\; \&theta;</span>}& \mathrm{<span\; class="notranslate">\; sin</span>}\mathrm{<span\; class="notranslate">\; \&theta;</span>}\\ <span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; sin</span>}\mathrm{<span\; class="notranslate">\; \&theta;</span>}& \mathrm{<span\; class="notranslate">\; cos</span>}\mathrm{<span\; class="notranslate">\; \&theta;</span>}\end{array}\right]\left[\begin{array}{c}{\mathrm{<span\; class="notranslate">\; i</span>}}_{\mathrm{<span\; class="notranslate">\; \&alpha;</span>}}\\ {\mathrm{<span\; class="notranslate">\; i</span>}}_{\mathrm{<span\; class="notranslate">\; \&beta;</span>}}\end{array}\right]<span\; class="notranslate">\; .</span>$

As an optimization method of the present invention: the fractional-order integral sliding mode surface S constructed in the step (4) is expressed by the following formula:

$\mathrm{<span\; class="notranslate">\; S</span>}<span\; class="notranslate">\; =</span>\mathrm{<span\; class="notranslate">\; e</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; t</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; +</span>{\mathrm{<span\; class="notranslate">\; <figure-callout\; id="1"\; label="C"\; filenames="130427101550.png,130427101610.png"\; state="\{\{state\}\}">C</figure-callout></span>}}_{\mathrm{<span\; class="notranslate">\; 1</span>}}<span\; class="notranslate">\; \&Center\; Dot;</span>\mathrm{<span\; class="notranslate">\; D.</span>}_{\mathrm{<span\; class="notranslate">\; t</span>}}^{<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; u</span>}}{}_{\mathrm{<span\; class="notranslate">\; 0</span>}}\mathrm{<span\; class="notranslate">\; e</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; t</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; \&phi;</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; t</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; =</span>\mathrm{<span\; class="notranslate">\; e</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; t</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; +</span>{\mathrm{<span\; class="notranslate">\; C</span>}}_{\mathrm{<span\; class="notranslate">\; 1</span>}}{\mathrm{<span\; class="notranslate">\; C</span>}}^{<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; u</span>}}\mathrm{<span\; class="notranslate">\; e</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; t</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; \&phi;</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; t</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; ,</span>$

定义为差值e(t)的分数阶积分值，u定义为分数阶积分滑模面的阶次，且0<u<1；Among them, e(t) is defined as the difference between the speed reference value ω _ref and the actual motor speed ω, t is defined as a time variable, and the sliding mode coefficient C ₁ >0,

Defined as the fractional integral value of the difference e(t), u is defined as the order of the fractional integral sliding mode surface, and 0<u<1;

和

分别为差值e(t)在t=0时的初始值和分数阶积分值。φ(t) is defined as the decay function, φ(t)=φ(0)e ^-t/n , n is the decay rate constant, and n>0, and

and

are the initial value and fractional integral value of the difference e(t) at t=0, respectively.

As an optimization method of the present invention: the model of the fractional-order integral sliding mode speed controller designed in the step (5) is:

${\mathrm{<span\; class="notranslate">\; i</span>}}_{\mathrm{<span\; class="notranslate">\; q</span>}}^{<span\; class="notranslate">\; *</span>}<span\; class="notranslate">\; =</span>\frac{\mathrm{<span\; class="notranslate">\; 1</span>}}{\mathrm{<span\; class="notranslate">\; h</span>}}<span\; class="notranslate">\; (</span>{\stackrel{<span\; class="notranslate">\; \&Center\; Dot;</span>}{\mathrm{<span\; class="notranslate">\; \&omega;</span>}}}_{\mathrm{<span\; class="notranslate">\; ref</span>}}<span\; class="notranslate">\; +</span>{\mathrm{<span\; class="notranslate">\; g\&omega;</span>}}_{\mathrm{<span\; class="notranslate">\; ref</span>}}<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; the\; ge</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; t</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; +</span>{\mathrm{<span\; class="notranslate">\; <figure-callout\; id="1"\; label="C"\; filenames="130427101550.png,130427101610.png"\; state="\{\{state\}\}">C</figure-callout></span>}}_{\mathrm{<span\; class="notranslate">\; 1</span>}}{\mathrm{<span\; class="notranslate">\; D.</span>}}^{\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; u</span>}}\mathrm{<span\; class="notranslate">\; e</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; t</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; +</span>\stackrel{<span\; class="notranslate">\; \&Center\; Dot;</span>}{\mathrm{<span\; class="notranslate">\; \&phi;</span>}}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; t</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; \&epsiv;sgn</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; S</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; )</span><span\; class="notranslate">\; ,</span>$

电机第二特征系数P定义为永磁同步电机的极对数，J定义为永磁同步电机的转动惯量，ψ_f定义为永磁体与定子交链的磁链，B定义为阻尼系数；

定义为速度给定值的导数，sgn(·)为符号函数，ε定义为切换增益，且ε>0。Among them, the first characteristic coefficient of the motor

The second characteristic coefficient of the motor P is defined as the number of pole pairs of the permanent magnet synchronous motor, J is defined as the moment of inertia of the permanent magnet synchronous motor, ψ _f is defined as the flux linkage between the permanent magnet and the stator, and B is defined as the damping coefficient;

Defined as the derivative of the speed given value, sgn(·) is a sign function, ε is defined as switching gain, and ε>0.

As an optimization method of the present invention: the PARK inverse transformation in the step (6) is expressed as the following matrix:

$\left[\begin{array}{c}{\mathrm{<span\; class="notranslate">\; i</span>}}_{\mathrm{<span\; class="notranslate">\; \&alpha;</span>}}\\ {\mathrm{<span\; class="notranslate">\; i</span>}}_{\mathrm{<span\; class="notranslate">\; \&beta;</span>}}\end{array}\right]<span\; class="notranslate">\; =</span>\left[\begin{array}{cc}\mathrm{<span\; class="notranslate">\; cos</span>}\mathrm{<span\; class="notranslate">\; \&theta;</span>}& <span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; sin</span>}\mathrm{<span\; class="notranslate">\; \&theta;</span>}\\ \mathrm{<span\; class="notranslate">\; sin</span>}\mathrm{<span\; class="notranslate">\; \&theta;</span>}& \mathrm{<span\; class="notranslate">\; cos</span>}\mathrm{<span\; class="notranslate">\; \&theta;</span>}\end{array}\right]\left[\begin{array}{c}{\mathrm{<span\; class="notranslate">\; i</span>}}_{\mathrm{<span\; class="notranslate">\; d</span>}}\\ {\mathrm{<span\; class="notranslate">\; i</span>}}_{\mathrm{<span\; class="notranslate">\; q</span>}}\end{array}\right]<span\; class="notranslate">\; .</span>$

The present invention also designs a permanent magnet synchronous motor speed control device based on fractional-order integral sliding mode, including a current acquisition module, a position sensor, a Clark transformation module, a first comparator, a second comparator, a third comparator, a PARK Transformation module, q-axis current controller, d-axis current controller, PARK inverse transformation module, space vector pulse width modulation module, three-phase inverter and permanent magnet synchronous motor, also including fractional order integral sliding mode speed controller, where :

The current collection module is used to collect the three-phase stator currents _ia , _ib and _ic of the motor, and transmit them to the Clark transformation module;

The Clark transformation module is used to transform the motor three-phase stator currents i _a , i _b and i _c into equivalent currents i _a and i _β in the α-β coordinate system, and transmit i _a and i _β to the PARK transformation module;

The position sensor obtains the motor actual running speed ω and the rotor position θ of the permanent magnet synchronous motor, and transmits the motor rotor position θ to the PARK transformation module and the PARK inverse transformation module respectively, and transmits the motor running speed ω to the first a comparator;

The PARK transformation module performs PARK transformation on the received rotor position θ and the equivalent currents i _a and i _β in the α-β coordinate system to generate actual output current values i _d and i _q in the dq coordinate system, and The actual output current value i _d of the d-axis is transmitted to the third comparator, and the actual output current value i _q of the q-axis is transmitted to the second comparator;

与q轴实际输出电流值i_q经过第二比较器比较后的差值输入至q轴电流控制器，所述q轴电流控制器产生q轴电压输出值u_q，并传输至PARK逆变换模块；The first comparator transmits the difference e(t) between the given speed value ω _ref and the actual operating speed ω of the motor to the fractional-order integral sliding mode speed controller, and the fractional-order integral sliding mode speed controller performs Speed adjustment processing, output q-axis current given value in dq coordinate system and transmitted to the second comparator, the q-axis current given value

The difference between the actual output current value i _q of the q-axis and the second comparator is input to the q-axis current controller, and the q-axis current controller generates the q-axis voltage output value u _q and transmits it to the PARK inverse transformation module ;

与d轴实际输出电流值i_d经过第三比较器比较后的差值输入至d轴电流控制器，所述d轴电流控制器产生d轴电压输出值u_d，并传输至PARK逆变换模块；d-axis current given value

The difference between the d-axis actual output current value i _d and the third comparator is input to the d-axis current controller, and the d-axis current controller generates the d-axis voltage output value u _d and transmits it to the PARK inverse transformation module ;

The PARK inverse transformation module performs PARK inverse transformation on the received q-axis voltage output value u _q and d-axis voltage output value u _d to generate equivalent voltage control given values u _α and u _β in the α-β coordinate system , and transmit u _α and u _β to the space vector pulse width modulation module;

The space vector pulse width modulation module uses the received equivalent voltage control given values u _α and u _β in the α-β coordinate system to generate a PWM signal and transmit it to the three-phase inverter. The PWM signal controls the three-phase The phase inverter generates three-phase voltage signals, and the three-phase inverter uses the three-phase voltage signals to control the operation of the permanent magnet synchronous motor.

Compared with the prior art, the present invention has the following advantages:

1. In the permanent magnet synchronous motor speed control method based on the fractional-order integral sliding mode proposed in the present invention, the order u of the fractional-order integral sliding mode speed controller is adjustable, and the optimum order u can be obtained by selecting an appropriate order u control performance;

2. The permanent magnet synchronous motor speed control method based on fractional-order integral sliding mode proposed in the present invention can overcome the integral saturation effect caused by large initial error or actuator saturation in the traditional integer-order integral sliding mode control method and transient performance degradation;

3. The permanent magnet synchronous motor speed control method based on fractional integral sliding mode proposed in the present invention has better dynamic performance and anti-disturbance ability, and more accurate speed when the system has load disturbance and parameter perturbation Follow precision.

Description of drawings

Fig. 1 is a block diagram of the present invention;

Fig. 2 is the functional block diagram of the fractional order integral sliding mode speed controller in the present invention;

Fig. 3 is the slope response comparative figure of fractional order integral sliding mode control and traditional integral sliding mode control, PI control in the present invention;

Fig. 4 is a partial enlarged view of part a in Fig. 3;

Fig. 5 is the sinusoidal response comparative figure of fractional order integral sliding mode control and traditional integral sliding mode control in the present invention;

Fig. 6 is a partially enlarged view of part b in Fig. 5;

Fig. 7 is a sinusoidal tracking error diagram of the fractional-order integral sliding mode control in the present invention and the traditional integral sliding mode control.

Detailed ways

Below in conjunction with accompanying drawing, the present invention is described in further detail:

As shown in Figure 1, the present invention has designed a kind of permanent magnet synchronous motor speed control method based on fractional order integral sliding mode, comprises following concrete steps:

Step (1): Collect the three-phase currents i _a , i _b , and i _c of the permanent magnet synchronous motor, and perform Clark transformation to convert them into equivalent currents i _a and i _β in the α-β coordinate system;

Step (2): Obtain the actual running speed ω and rotor position θ of the permanent magnet synchronous motor through the position sensor;

Step (3): According to the motor rotor position θ in step (2), perform PARK transformation on the motor rotor position θ and the equivalent current i _a and i _β in the α-β coordinate system in step (1) to generate dq coordinates The actual output current values i _d and i _q under the system;

Step (4): Use the difference e(t) between the actual running speed ω of the motor in step (2) and the given speed ω _ref to construct the fractional-order integral sliding mode surface S;

Step (5): According to the fractional-order integral sliding mode surface S in step (4), design a fractional-order integral sliding-mode speed controller to adjust the speed, so as to obtain the given value of the q-axis current in the dq coordinate system

与q轴实际输出电流值i_q比较后的差值进行q轴电流调节处理，产生q轴电压输出值u_q，对d轴电流给定值

与d轴实际输出电流值i_d比较后的差值进行d轴电流调节处理，产生d轴电压输出值u_d，将上述d-q坐标系下的电压输出值u_d和u_q进行PARK逆变换处理，从而得到α-β坐标系下的等效电压控制给定值u_α和u_β，对所述等效电压控制给定值u_α和u_β进行空间矢量脉宽调制，产生PWM信号，利用所述PWM信号控制三相逆变器产生三相电压信号，并利用所述三相电压信号控制永磁同步电机运行。Step (6): Set the value of the q-axis current

The difference after comparison with the actual output current value i _q of the q-axis is adjusted for the q-axis current to generate the q-axis voltage output value u _q , and the given value of the d-axis current

The difference between the d-axis actual output current value i _d is compared with the d-axis current adjustment process to generate the d-axis voltage output value u _d , and the above-mentioned voltage output values u _d and u _q in the dq coordinate system are subjected to PARK inverse transformation processing , so as to obtain the equivalent voltage control given values u _α and u _β in the α-β coordinate system, perform space vector pulse width modulation on the equivalent voltage control given values u _α and u _β to generate PWM signals, and use The PWM signal controls the three-phase inverter to generate a three-phase voltage signal, and uses the three-phase voltage signal to control the operation of the permanent magnet synchronous motor.

As an optimization method of the present invention: the Clark transformation in the step (1) is expressed as the following matrix:

$\left[\begin{array}{c}{\mathrm{<span\; class="notranslate">\; i</span>}}_{\mathrm{<span\; class="notranslate">\; \&alpha;</span>}}\\ {\mathrm{<span\; class="notranslate">\; i</span>}}_{\mathrm{<span\; class="notranslate">\; \&beta;</span>}}\end{array}\right]<span\; class="notranslate">\; =</span>\frac{\mathrm{<span\; class="notranslate">\; 2</span>}}{\mathrm{<span\; class="notranslate">\; 3</span>}}\left[\begin{array}{ccc}\mathrm{<span\; class="notranslate">\; 1</span>}& <span\; class="notranslate">\; -</span>\frac{\mathrm{<span\; class="notranslate">\; 1</span>}}{\mathrm{<span\; class="notranslate">\; 2</span>}}& <span\; class="notranslate">\; -</span>\frac{\mathrm{<span\; class="notranslate">\; 1</span>}}{\mathrm{<span\; class="notranslate">\; 2</span>}}\\ \mathrm{<span\; class="notranslate">\; 0</span>}& \frac{\sqrt{\mathrm{<span\; class="notranslate">\; 3</span>}}}{\mathrm{<span\; class="notranslate">\; 2</span>}}& <span\; class="notranslate">\; -</span>\frac{\sqrt{\mathrm{<span\; class="notranslate">\; 3</span>}}}{\mathrm{<span\; class="notranslate">\; 2</span>}}\end{array}\right]\left[\begin{array}{c}{\mathrm{<span\; class="notranslate">\; i</span>}}_{\mathrm{<span\; class="notranslate">\; a</span>}}\\ {\mathrm{<span\; class="notranslate">\; i</span>}}_{\mathrm{<span\; class="notranslate">\; b</span>}}\\ {\mathrm{<span\; class="notranslate">\; i</span>}}_{\mathrm{<span\; class="notranslate">\; c</span>}}\end{array}\right]<span\; class="notranslate">\; .</span>$

As an optimization method of the present invention: the PARK transformation in the step (3) is expressed as the following matrix:

$\left[\begin{array}{c}{\mathrm{<span\; class="notranslate">\; i</span>}}_{\mathrm{<span\; class="notranslate">\; d</span>}}\\ {\mathrm{<span\; class="notranslate">\; i</span>}}_{\mathrm{<span\; class="notranslate">\; q</span>}}\end{array}\right]<span\; class="notranslate">\; =</span>\left[\begin{array}{cc}\mathrm{<span\; class="notranslate">\; cos</span>}\mathrm{<span\; class="notranslate">\; \&theta;</span>}& \mathrm{<span\; class="notranslate">\; sin</span>}\mathrm{<span\; class="notranslate">\; \&theta;</span>}\\ <span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; sin</span>}\mathrm{<span\; class="notranslate">\; \&theta;</span>}& \mathrm{<span\; class="notranslate">\; cos</span>}\mathrm{<span\; class="notranslate">\; \&theta;</span>}\end{array}\right]\left[\begin{array}{c}{\mathrm{<span\; class="notranslate">\; i</span>}}_{\mathrm{<span\; class="notranslate">\; \&alpha;</span>}}\\ {\mathrm{<span\; class="notranslate">\; i</span>}}_{\mathrm{<span\; class="notranslate">\; \&beta;</span>}}\end{array}\right]<span\; class="notranslate">\; .</span>$

As an optimization method of the present invention: the fractional-order integral sliding mode surface S constructed in the step (4) is expressed by the following formula:

$\mathrm{<span\; class="notranslate">\; S</span>}<span\; class="notranslate">\; =</span>\mathrm{<span\; class="notranslate">\; e</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; t</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; +</span>{\mathrm{<span\; class="notranslate">\; <figure-callout\; id="1"\; label="C"\; filenames="130427101550.png,130427101610.png"\; state="\{\{state\}\}">C</figure-callout></span>}}_{\mathrm{<span\; class="notranslate">\; 1</span>}}<span\; class="notranslate">\; \&Center\; Dot;</span>\mathrm{<span\; class="notranslate">\; D.</span>}_{\mathrm{<span\; class="notranslate">\; t</span>}}^{<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; u</span>}}{}_{\mathrm{<span\; class="notranslate">\; 0</span>}}\mathrm{<span\; class="notranslate">\; e</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; t</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; \&phi;</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; t</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; =</span>\mathrm{<span\; class="notranslate">\; e</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; t</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; +</span>{\mathrm{<span\; class="notranslate">\; C</span>}}_{\mathrm{<span\; class="notranslate">\; 1</span>}}{\mathrm{<span\; class="notranslate">\; C</span>}}^{<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; u</span>}}\mathrm{<span\; class="notranslate">\; e</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; t</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; \&phi;</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; t</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; ,</span>$

定义为差值e(t)的分数阶积分值，u定义为分数阶积分滑模面的阶次，且0<u<1；Among them, e(t) is defined as the difference between the speed reference value ω _ref and the actual motor speed ω, t is defined as a time variable, and the sliding mode coefficient C ₁ >0,

Defined as the fractional integral value of the difference e(t), u is defined as the order of the fractional integral sliding mode surface, and 0<u<1;

和

分别为差值e(t)在t=0时的初始值和分数阶积分值，综上所述，对任意的初始状态e(0)，在t=0时刻有S=0，即系统的初始状态就处在滑模面上，消除了到达过程。φ(t) is defined as the decay function, φ(t)=φ(0)e ^-t/n , n is the decay rate constant, and n>0, and

and

are the initial value and fractional integral value of the difference e(t) at t=0, respectively. In summary, for any initial state e(0), there is S=0 at time t=0, that is, the system The initial state is on the sliding surface, eliminating the arrival process.

As an optimization method of the present invention: the model of the fractional-order integral sliding mode speed controller designed in the step (5) is:

${\mathrm{<span\; class="notranslate">\; i</span>}}_{\mathrm{<span\; class="notranslate">\; q</span>}}^{<span\; class="notranslate">\; *</span>}<span\; class="notranslate">\; =</span>\frac{\mathrm{<span\; class="notranslate">\; 1</span>}}{\mathrm{<span\; class="notranslate">\; h</span>}}<span\; class="notranslate">\; (</span>{\stackrel{<span\; class="notranslate">\; \&Center\; Dot;</span>}{\mathrm{<span\; class="notranslate">\; \&omega;</span>}}}_{\mathrm{<span\; class="notranslate">\; ref</span>}}<span\; class="notranslate">\; +</span>{\mathrm{<span\; class="notranslate">\; g\&omega;</span>}}_{\mathrm{<span\; class="notranslate">\; ref</span>}}<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; the\; ge</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; t</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; +</span>{\mathrm{<span\; class="notranslate">\; <figure-callout\; id="1"\; label="C"\; filenames="130427101550.png,130427101610.png"\; state="\{\{state\}\}">C</figure-callout></span>}}_{\mathrm{<span\; class="notranslate">\; 1</span>}}{\mathrm{<span\; class="notranslate">\; D.</span>}}^{\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; u</span>}}\mathrm{<span\; class="notranslate">\; e</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; t</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; +</span>\stackrel{<span\; class="notranslate">\; \&Center\; Dot;</span>}{\mathrm{<span\; class="notranslate">\; \&phi;</span>}}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; t</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; \&epsiv;sgn</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; S</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; )</span><span\; class="notranslate">\; ,</span>$

P定义为永磁同步电机的极对数，J定义为永磁同步电机的转动惯量，ψ_f定义为永磁体与定子交链的磁链，B定义为阻尼系数；

定义为速度给定值的导数，sgn(·)为符号函数，ε定义为切换增益，且ε>0。Among them, the first characteristic coefficient of the motor The second characteristic coefficient of the motor

P is defined as the number of pole pairs of the permanent magnet synchronous motor, J is defined as the moment of inertia of the permanent magnet synchronous motor, ψ _f is defined as the flux linkage between the permanent magnet and the stator, and B is defined as the damping coefficient;

Defined as the derivative of the speed given value, sgn(·) is a sign function, ε is defined as switching gain, and ε>0.

As an optimization method of the present invention: the PARK inverse transformation in the step (6) is expressed as the following matrix:

$\left[\begin{array}{c}{\mathrm{<span\; class="notranslate">\; i</span>}}_{\mathrm{<span\; class="notranslate">\; \&alpha;</span>}}\\ {\mathrm{<span\; class="notranslate">\; i</span>}}_{\mathrm{<span\; class="notranslate">\; \&beta;</span>}}\end{array}\right]<span\; class="notranslate">\; =</span>\left[\begin{array}{cc}\mathrm{<span\; class="notranslate">\; cos</span>}\mathrm{<span\; class="notranslate">\; \&theta;</span>}& <span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; sin</span>}\mathrm{<span\; class="notranslate">\; \&theta;</span>}\\ \mathrm{<span\; class="notranslate">\; sin</span>}\mathrm{<span\; class="notranslate">\; \&theta;</span>}& \mathrm{<span\; class="notranslate">\; cos</span>}\mathrm{<span\; class="notranslate">\; \&theta;</span>}\end{array}\right]\left[\begin{array}{c}{\mathrm{<span\; class="notranslate">\; i</span>}}_{\mathrm{<span\; class="notranslate">\; d</span>}}\\ {\mathrm{<span\; class="notranslate">\; i</span>}}_{\mathrm{<span\; class="notranslate">\; q</span>}}\end{array}\right]<span\; class="notranslate">\; .</span>$

In a specific embodiment, we design the steps of the fractional-order integral sliding mode speed controller as shown in Figure 2:

Step a: The mechanical motion equation of the permanent magnet synchronous motor in the d-q coordinate system can be written as

$\stackrel{<span\; class="notranslate">\; \&Center\; Dot;</span>}{\mathrm{<span\; class="notranslate">\; \&omega;</span>}}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; g\&omega;</span>}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; \&Gamma;</span>}<span\; class="notranslate">\; =</span>{\mathrm{<span\; class="notranslate">\; hi</span>}}_{\mathrm{<span\; class="notranslate">\; q</span>}}<span\; class="notranslate">\; ,</span>$

T_L定义为永磁同步电机的负载转矩,考虑到电机运行过程中的参数不确定量，永磁同步电机的机械运动方程可进一步写为Among them, the third characteristic coefficient of the motor

T _L is defined as the load torque of the permanent magnet synchronous motor. Considering the parameter uncertainties during the operation of the motor, the mechanical motion equation of the permanent magnet synchronous motor can be further written as

$\stackrel{<span\; class="notranslate">\; \&CenterDot;</span>}{\mathrm{<span\; class="notranslate">\; \&omega;</span>}}<span\; class="notranslate">\; +</span><span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; g</span>}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; \&Delta;\; g</span>}<span\; class="notranslate">\; )</span>\mathrm{<span\; class="notranslate">\; \&omega;</span>}<span\; class="notranslate">\; +</span><span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; \&Gamma;</span>}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; \&Delta;\&Gamma;</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; =</span><span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; h</span>}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; \&Delta;h</span>}<span\; class="notranslate">\; )</span>{\mathrm{<span\; class="notranslate">\; i</span>}}_{\mathrm{<span\; class="notranslate">\; q</span>}}<span\; class="notranslate">\; ,</span>$

Among them, Δh is defined as the variation of the first characteristic coefficient h of the motor, Δg is defined as the variation of the second characteristic coefficient g of the motor, and ΔΓ is defined as the variation of the third characteristic coefficient Γ of the motor;

Calculate the derivative of the difference e(t) between the given speed value ω _ref of the permanent magnet synchronous motor and the actual speed ω of the motor

Among them, the process function Disturbance function δ(t)=Δgω+ΔΓ+Γ-Δhi _q ;

Step b: Calculate the derivative of the fractional integral sliding mode surface S to get:

$\stackrel{<span\; class="notranslate">\; \&Center\; Dot;</span>}{\mathrm{<span\; class="notranslate">\; S</span>}}<span\; class="notranslate">\; =</span>\stackrel{<span\; class="notranslate">\; \&Center\; Dot;</span>}{\mathrm{<span\; class="notranslate">\; e</span>}}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; t</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; +</span>{\mathrm{<span\; class="notranslate">\; <figure-callout\; id="1"\; label="C"\; filenames="130427101550.png,130427101610.png"\; state="\{\{state\}\}">C</figure-callout></span>}}_{\mathrm{<span\; class="notranslate">\; 1</span>}}{\mathrm{<span\; class="notranslate">\; D.</span>}}^{\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; u</span>}}\mathrm{<span\; class="notranslate">\; e</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; t</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; +</span>\stackrel{<span\; class="notranslate">\; \&Center\; Dot;</span>}{\mathrm{<span\; class="notranslate">\; \&phi;</span>}}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; t</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; ;</span>$

则有：Step c: When the system moves on the sliding surface, there is And use the obtained in step a

Then there are:

thus get,

Step d: Using the constant velocity approach rate, design the fractional-order integral sliding mode control law as follows:

Step e: The output of the q-axis current control quantity is obtained by the following formula:

${\mathrm{<span\; class="notranslate">\; i</span>}}_{\mathrm{<span\; class="notranslate">\; q</span>}}^{<span\; class="notranslate">\; *</span>}<span\; class="notranslate">\; =</span>\frac{\mathrm{<span\; class="notranslate">\; 1</span>}}{\mathrm{<span\; class="notranslate">\; h</span>}}<span\; class="notranslate">\; (</span>{\stackrel{<span\; class="notranslate">\; \&Center\; Dot;</span>}{\mathrm{<span\; class="notranslate">\; \&omega;</span>}}}_{\mathrm{<span\; class="notranslate">\; ref</span>}}<span\; class="notranslate">\; +</span>{\mathrm{<span\; class="notranslate">\; g\&omega;</span>}}_{\mathrm{<span\; class="notranslate">\; ref</span>}}<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; the\; ge</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; t</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; +</span>{\mathrm{<span\; class="notranslate">\; <figure-callout\; id="1"\; label="C"\; filenames="130427101550.png,130427101610.png"\; state="\{\{state\}\}">C</figure-callout></span>}}_{\mathrm{<span\; class="notranslate">\; 1</span>}}{\mathrm{<span\; class="notranslate">\; D.</span>}}^{\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; u</span>}}\mathrm{<span\; class="notranslate">\; e</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; t</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; +</span>\stackrel{<span\; class="notranslate">\; \&CenterDot;</span>}{\mathrm{<span\; class="notranslate">\; \&phi;</span>}}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; t</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; \&epsiv;sgn</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; S</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; )</span><span\; class="notranslate">\; ;</span>$

Step f: To prove the stability of the designed fractional-order integral sliding mode speed controller, the Lyapunov function is defined as follows:

$\mathrm{<span\; class="notranslate">\; V</span>}<span\; class="notranslate">\; =</span>\frac{\mathrm{<span\; class="notranslate">\; 1</span>}}{\mathrm{<span\; class="notranslate">\; 2</span>}}{\mathrm{<span\; class="notranslate">\; S</span>}}^{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; ,</span>$

Calculate the derivative of the above Lyapunov function, and use the output of the q-axis current control value obtained in step e, it can be obtained:

则根据Lyapunov稳定性理论可知，此时所设计的分数阶积分滑模转速控制器是稳定的，系统能从任意初始状态在有限时间内达到切换面。We can see that when the switching gain ε satisfies ε>|δ(t)|, there is

According to the Lyapunov stability theory, the fractional-order integral sliding mode speed controller designed at this time is stable, and the system can reach the switching surface within a limited time from any initial state.

The present invention also designs a permanent magnet synchronous motor speed control device based on fractional-order integral sliding mode, including a current acquisition module, a position sensor, a Clark transformation module, a first comparator, a second comparator, a third comparator, a PARK Transformation module, q-axis current controller, d-axis current controller, PARK inverse transformation module, space vector pulse width modulation module, three-phase inverter and permanent magnet synchronous motor, also includes fractional order integral sliding mode speed controller, where :

The current collection module is used to collect the three-phase stator currents _ia , _ib and _ic of the motor, and transmit them to the Clark transformation module;

The Clark transformation module is used to transform the motor three-phase stator currents i _a , i _b and i _c into equivalent currents i _a and i _β in the α-β coordinate system, and transmit i _a and i _β to the PARK transformation module;

The position sensor obtains the motor actual running speed ω and the rotor position θ of the permanent magnet synchronous motor, and transmits the motor rotor position θ to the PARK transformation module and the PARK inverse transformation module respectively, and transmits the motor running speed ω to the first a comparator;

The PARK transformation module performs PARK transformation on the received rotor position θ and the equivalent currents i _a and i _β in the α-β coordinate system to generate actual output current values i _d and i _q in the dq coordinate system, and The actual output current value i _d of the d-axis is transmitted to the third comparator, and the actual output current value i _q of the q-axis is transmitted to the second comparator;

，并传输至第二比较器，所述q轴电流给定值

与q轴实际输出电流值i_q经过第二比较器比较后的差值输入至q轴电流控制器，所述q轴电流控制器产生q轴电压输出值u_q，并传输至PARK逆变换模块；The first comparator transmits the difference e(t) between the given speed value ω _ref and the actual operating speed ω of the motor to the fractional-order integral sliding mode speed controller, and the fractional-order integral sliding mode speed controller performs Speed adjustment processing, output q-axis current given value in dq coordinate system

, and transmitted to the second comparator, the q-axis current given value

The difference between the actual output current value i _q of the q-axis and the second comparator is input to the q-axis current controller, and the q-axis current controller generates the q-axis voltage output value u _q and transmits it to the PARK inverse transformation module ;

与d轴实际输出电流值i_d经过第三比较器比较后的差值输入至d轴电流控制器，所述d轴电流控制器产生d轴电压输出值u_d，并传输至PARK逆变换模块；d-axis current given value

The difference between the d-axis actual output current value i _d and the third comparator is input to the d-axis current controller, and the d-axis current controller generates the d-axis voltage output value u _d and transmits it to the PARK inverse transformation module ;

The PARK inverse transformation module performs PARK inverse transformation on the received q-axis voltage output value u _q and d-axis voltage output value u _d to generate equivalent voltage control given values u _α and u _β in the α-β coordinate system , and transmit u _α and u _β to the space vector pulse width modulation module;

The space vector pulse width modulation module uses the received equivalent voltage control given values u _α and u _β in the α-β coordinate system to generate a PWM signal and transmit it to the three-phase inverter. The PWM signal controls the three-phase The phase inverter generates three-phase voltage signals, and the three-phase inverter uses the three-phase voltage signals to control the operation of the permanent magnet synchronous motor.

As shown in Figure 3, it is the slope response comparison diagram of fractional order integral sliding mode control and traditional integral sliding mode control, PI control in the present invention;

Fig. 4 is a partial enlarged view of part a in Fig. 3;

As shown in Figure 5, it is the sinusoidal response comparison diagram of fractional-order integral sliding mode control and traditional integral sliding mode control in the present invention;

Fig. 6 is a partially enlarged view of part b in Fig. 5;

As shown in Figure 7, it is the sinusoidal tracking error diagram of fractional-order integral sliding mode control and traditional integral sliding mode control in the present invention;

It can be seen from Fig. 3-Fig. 7 that when there is load disturbance in the system, the proposed method has better dynamic performance, stronger anti-disturbance ability, and more accurate speed than traditional integral sliding mode control and conventional PI control. Follow precision.

The above embodiments are only to illustrate the technical ideas of the present invention, and can not limit the protection scope of the present invention with this. All technical ideas proposed in accordance with the present invention, any changes made on the basis of technical solutions, all fall within the protection scope of the present invention. Inside.

Claims (7)

1. the permagnetic synchronous motor method for control speed based on the fractional order integration sliding formwork is characterized in that, comprises following concrete steps:

Step (1): the three-phase current i that gathers permagnetic synchronous motor _a, i _b, i _c, and carry out the Clark conversion, be converted into the equivalent current i under the alpha-beta coordinate system _aAnd i _β

Step (2): the motor actual motion speed omega and the rotor position that obtain permagnetic synchronous motor by position transducer;

Step (3): according to the motor rotor position θ in the step (2), with the equivalent current i under the alpha-beta coordinate system in motor rotor position θ and the step (1) _aAnd i _βCarry out the PARK conversion, produce the actual output current value i under the d-q coordinate system _dAnd i _q

Step (4): utilize motor actual motion speed omega and speed set-point ω in the step (2) _RefDifference e (t) relatively, structure fractional order integration sliding-mode surface S;

Step (5): carry out rotational speed regulation according to the design of the fractional order integration sliding-mode surface S in the step (4) fractional order integration sliding formwork rotational speed governor, thereby obtain q shaft current set-point under the d-q coordinate system

Step (6): to q shaft current set-point

With q axle actual output current value i _qDifference is relatively carried out the q shaft current and is regulated processing, produces q shaft voltage output valve u _q, to d shaft current set-point

With d axle actual output current value i _dDifference is relatively carried out the d shaft current and is regulated processing, produces d shaft voltage output valve u _d, with the voltage output value u under the above-mentioned d-q coordinate system _dAnd u _qCarry out the PARK inversion process, thereby obtain the equivalent voltage control set-point u under the alpha-beta coordinate system _αAnd u _β, to described equivalent voltage control set-point u _αAnd u _βCarry out space vector pulse width modulation, produce pwm signal, utilize described pwm signal control three-phase inverter to produce three-phase voltage signal, and utilize the operation of described three-phase voltage signal control permagnetic synchronous motor.

2. the permagnetic synchronous motor method for control speed based on the fractional order integration sliding formwork according to claim 1 is characterized in that, the Clark conversion in the described step (1) is expressed as matrix:

$\left[\begin{array}{c}{i}_{\mathrm{\&alpha;}}\\ i_{\mathrm{\&beta;}}\end{array}\right]=\frac{2}{3}\left[\begin{array}{ccc}1& -\frac{1}{2}& -\frac{1}{2}\\ 0& \frac{\sqrt{3}}{2}& -\frac{\sqrt{3}}{2}\end{array}\right]\left[\begin{array}{c}{i}_a\\ i_b\\ i_c\end{array}\right].$

3. the permagnetic synchronous motor method for control speed based on the fractional order integration sliding formwork according to claim 1 is characterized in that, the PARK conversion in the described step (3) is expressed as matrix:

$\left[\begin{array}{c}{i}_d\\ i_q\end{array}\right]=\left[\begin{array}{cc}\cos \mathrm{\&theta;}& \sin \mathrm{\&theta;}\\ -\sin \mathrm{\&theta;}& \cos \mathrm{\&theta;}\end{array}\right]\left[\begin{array}{c}{i}_{\mathrm{\&alpha;}}\\ i_{\mathrm{\&beta;}}\end{array}\right].$

4. the permagnetic synchronous motor method for control speed based on the fractional order integration sliding formwork according to claim 1 is characterized in that, the fractional order integration sliding-mode surface S of structure represents with following formula in the described step (4):

$S=e\left(t\right)+C_1\&CenterDot;{}_{0}D_t^{-u}e\left(t\right)+\mathrm{\&phi;}\left(t\right)=e\left(t\right)+C_1{C}^{-u}e\left(t\right)+\mathrm{\&phi;}\left(t\right),$

Wherein, e (t) is defined as speed set-point ω _RefWith the difference of motor actual speed ω after relatively, t is defined as time variable, sliding formwork coefficient C ₁0,

Be defined as the fractional order integration value of difference e (t), u is defined as the order of fractional order integration sliding-mode surface, and 0＜u＜1;

φ (t) is defined as attenuation function, φ (t)=φ (0) e ^-t/n, n is rate of decay constant, and n〉and 0, and

With

Be respectively initial value and the fractional order integration value of difference e (t) when t=0.

5. the permagnetic synchronous motor method for control speed based on the fractional order integration sliding formwork according to claim 1 is characterized in that, the model of the fractional order integration sliding formwork rotational speed governor of design is in the described step (5):

$i_q^{*}=\frac{1}{h}({\stackrel{\&CenterDot;}{\mathrm{\&omega;}}}_{\mathrm{ref}}+{\mathrm{g\&omega;}}_{\mathrm{ref}}-\mathrm{ge}\left(t\right)+C_1{D}^{1-u}e\left(t\right)+\stackrel{\&CenterDot;}{\mathrm{\&phi;}}\left(t\right)+\mathrm{\&epsiv;sgn}\left(S\right)),$

Wherein, motor first characteristic coefficient Motor second characteristic coefficient P is defined as the number of pole-pairs of permagnetic synchronous motor, and J is defined as the moment of inertia of permagnetic synchronous motor, ψ _fBe defined as the magnetic linkage of permanent magnet and stator interlinkage, B is defined as damping coefficient;

Be defined as the derivative of speed set-point, sgn () is sign function, and ε is defined as handoff gain, and ε〉0.

6. the permagnetic synchronous motor method for control speed based on the fractional order integration sliding formwork according to claim 1 is characterized in that, the PARK inverse transformation is expressed as matrix in the described step (6):

$\left[\begin{array}{c}{i}_{\mathrm{\&alpha;}}\\ i_{\mathrm{\&beta;}}\end{array}\right]=\left[\begin{array}{cc}\cos \mathrm{\&theta;}& -\sin \mathrm{\&theta;}\\ \sin \mathrm{\&theta;}& \cos \mathrm{\&theta;}\end{array}\right]\left[\begin{array}{c}{i}_d\\ i_q\end{array}\right].$

7. permagnetic synchronous motor speed control unit based on the fractional order integration sliding formwork, comprise current acquisition module, position transducer, Clark conversion module, first comparator, second comparator, the 3rd comparator, PARK conversion module, q shaft current controller, d shaft current controller, PARK inverse transform module, space vector pulse width modulation module, three-phase inverter and permagnetic synchronous motor, it is characterized in that, also comprise fractional order integration sliding formwork rotational speed governor, wherein:

Described current acquisition module is used for gathering motor threephase stator current i _a, i _bAnd i _c, and transfer to the Clark conversion module;

Described Clark conversion module is used for motor threephase stator current i _a, i _bAnd i _cBe transformed to the equivalent current i under the alpha-beta coordinate system _aAnd i _β, and with i _aAnd i _βTransfer to the PARK conversion module;

Described position transducer obtains motor actual motion speed omega and the rotor position of permagnetic synchronous motor, and described motor rotor position θ transferred to PARK conversion module and PARK inverse transform module respectively, described motor speed of service ω is transferred to first comparator;

Described PARK conversion module is with the rotor position that receives and the equivalent current i under the alpha-beta coordinate system _a, i _βCarry out the PARK conversion, produce the actual output current value i under the d-q coordinate system _dAnd i _q, and with d axle actual output current value i _dTransfer to the 3rd comparator, with q axle actual output current value i _qTransfer to second comparator;

Described first comparator is with speed set-point ω _RefTransfer to fractional order integration sliding formwork rotational speed governor with the difference e (t) of motor actual motion speed omega after relatively, described fractional order integration sliding formwork rotational speed governor carries out rotational speed regulation to be handled, q shaft current set-point under the output d-q coordinate system

And transfer to second comparator, described q shaft current set-point With q axle actual output current value i _qInput to q shaft current controller through the difference of second comparator after relatively, described q shaft current controller produces q shaft voltage output valve u _q, and transfer to the PARK inverse transform module;

D shaft current set-point

With d axle actual output current value i _dInput to d shaft current controller through the difference of the 3rd comparator after relatively, described d shaft current controller produces d shaft voltage output valve u _d, and transfer to the PARK inverse transform module;

Described PARK inverse transform module is with the q shaft voltage output valve u that receives _qWith d shaft voltage output valve u _dCarry out the PARK inverse transformation, produce the equivalent voltage control set-point u under the alpha-beta coordinate system _αAnd u _β, and with u _αAnd u _βTransfer to the space vector pulse width modulation module;

Equivalent voltage control set-point u under the alpha-beta coordinate system that the utilization of described space vector pulse width modulation module receives _αAnd u _βProduce pwm signal, and transfer to three-phase inverter, described pwm signal control three-phase inverter produces three-phase voltage signal, and three-phase inverter utilizes the operation of described three-phase voltage signal control permagnetic synchronous motor.

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