JP2019041562A - Power conversion apparatus - Google Patents

Abstract

To provide a power conversion apparatus capable of meeting a power supply harmonic regulation without need for adding a reactor of large capacity in a configuration of an electrolytic capacitor-less inverter.SOLUTION: The power conversion apparatus creates an α-axis current command value iαand a β-axis current command value iβon a stationary coordinate system on the basis of an operation amount of controlling an output torque τ, creates an α-axis voltage command value vαand a β-axis voltage command value vβso that a deviation between them and an α-axis current iα, a β-axis current iβ becomes smaller, creates a DC link current command value ion the basis of an operation amount of controlling a speed and corrects the α-axis voltage command value vαand the β-axis voltage command value vβso that deviation between it and the DC link current command value ibecomes smaller.SELECTED DRAWING: Figure 1

Description

  The present invention relates to a power converter that supplies electric power to a motor.

  A so-called electrolytic capacitorless inverter, which is a power converter with a reduced smoothing capacitor, is composed of a single-phase diode rectifier, a small-capacity film capacitor, and a three-phase voltage source inverter. In such a power converter, the smoothing capacitor is miniaturized by supplying the motor with the single-phase power pulsation as the torque pulsation without absorbing the single-phase power pulsation at the moment of inertia.

As an example of such a power converter, a small-capacity film capacitor is provided in the DC link section, and inverter power or motor power is feedback-controlled, and motor d-axis current _id and q-axis current _iq are PI-controlled. In some cases, the output power of the inverter circuit is obtained while improving the input / output waveform (see, for example, Patent Document 1).

Japanese Patent No. 5813934

  In a power conversion device such as Patent Document 1, the target of power feedback is power at the fundamental wave frequency of the motor current and the motor terminal voltage, because harmonic suppression is achieved by causing the motor current to follow the fundamental wave. Although it is possible to suppress average harmonics, it has been difficult to achieve instantaneous suppression of power supply harmonic distortion.

  Further, since the responsiveness is limited by the band of the motor current control system, there is a problem that power harmonics are generated when the frequency of the spatial harmonics is higher than the control band. That is, according to the dq axis PI control as in Patent Document 1, since the input is a DC amount, the target value responsiveness (especially steady deviation) is improved and the transient characteristics are improved, but the control is performed after coordinate conversion. Therefore, the disturbance response cannot be designed, and the motor's spatial harmonic components appear as the motor current. In the electrolytic capacitorless inverter, the motor current becomes a direct current and returns to the power supply current. Therefore, in the voltage control system, the disturbance of the motor space, which is a disturbance, appears as the power supply harmonic, resulting in current distortion.

  Therefore, in Patent Document 1, in order to reduce power supply harmonics, it is necessary to provide a large-capacity reactor that is not originally necessary for power factor improvement, leading to an increase in cost. There is also a method of providing an LC filter so that instantaneous fluctuations in the input current deviate from the power supply harmonic regulation. However, since the impedance of the power supply varies depending on the installation situation, the LC filter does not necessarily satisfy the power supply harmonic regulation. There was also a problem that was not limited.

  The present invention has been made in order to solve the conventional technical problem, and in the configuration of a so-called electrolytic capacitor-less inverter, it is possible to satisfy the power supply harmonic regulation without providing a large-capacity reactor. An object is to provide a power converter.

The power conversion device of the present invention supplies power to a motor, has a converter circuit for full-wave rectification of input AC, and a capacitor connected in parallel to the output of the converter circuit, and outputs a DC voltage. A DC link unit, an inverter circuit that switches and converts the output of the DC link unit to AC and supplies the motor to the motor, and a control unit that controls switching, the control unit controls the speed of the motor A speed control unit for determining the amount, a torque control unit for determining an operation amount for controlling the output torque τ _inv of the inverter circuit, and a voltage control unit for determining an operation amount for controlling the α-axis current iα and β-axis current iβ of the motor. provided, the speed control unit, an operation amount for controlling the speed of the motor, synchronously by pulsing twice the power source frequency omega _s of the input AC, torque control section is operated to control the speed of the motor From generates an output torque command value tau _inv ^* of the inverter circuit, based on the output torque command value tau _inv ^*, it obtains the operation amount for controlling the output torque tau _inv of the inverter circuit, the voltage control unit, the inverter circuit based on the operation amount for controlling the output torque tau _inv, generates a motor α-axis current value i.alpha ^* and β-axis current value i.beta ^* on the stationary coordinate system, these α-axis current value i.alpha ^* and β-axis current The motor α-axis voltage command value vα ^* and β-axis voltage command value vβ ^* are generated so that the deviation between the command value iβ ^* and the α-axis current iα and β-axis current iβ is small, and the motor speed is controlled. based on the operation amount to generate a DC link current command value i _dc flowing through the inverter circuit ^*, DC link current i _dc is the DC link current command value i _dc ^* and equal way α axis voltage value v? ^* and to correct the β-axis voltage command value v? ^* And butterflies.

In the power converter of the invention of claim 2, the operation amount for controlling the motor speed required by the speed controller in the above invention is obtained by dividing the input power command value of the inverter circuit by the rotational angular frequency ω _rm of the mechanical angle, The input torque command value τ _in ^* converted into a dimension, and the voltage control unit generates a DC link current command value i _dc ^* based on the input torque command value τ _in ^* .

According to a third aspect of the present invention, the operation amount for controlling the output torque τ _inv of the inverter circuit required by the torque control unit in each of the above inventions is the q-axis current command value i _q ^* of the motor, and the voltage control unit Is characterized in that an α-axis current command value iα ^* and a β-axis current command value iβ ^* are generated based on the q-axis current command value i _q ^* .

According to a fourth aspect of the present invention, there is provided the power converter according to any one of the above-described aspects, wherein the voltage control unit is configured to perform the α-axis current command value iα ^* and β-axis current command value iβ ^* , the α-axis current iα, and the β-axis current iβ by PI control. Α-axis voltage command value vα ^* and β-axis voltage command value vβ ^* are generated, and the α-axis voltage command value is set so that the DC link current i _dc becomes equal to the DC link current command value i _dc ^*. vα ^* and β-axis voltage command value vβ ^* are corrected.

According to a fifth aspect of the present invention, there is provided the power conversion device according to the above invention, wherein the voltage control unit performs feedback control with an F / B control unit including a control gain having a sine wave component of the motor current frequency ω _re of the motor. The shaft current iα and the β-axis current iβ are made to follow the motor current frequency ω _re .

According to a sixth aspect of the present invention, there is provided the power converter according to any one of the above-mentioned inventions, wherein the voltage control unit uses the correction coefficient A _IVC calculated from the ratio of the DC link current command value i _dc ^* and the DC link current i _dc. The α-axis voltage command value vα ^* and β-axis voltage command value vβ ^* are corrected so that the current i _dc and the DC link current command value i _dc ^* coincide with each other, and the corrected α-axis voltage command value vα ^* and β-axis are corrected. When the voltage command value vβ ^* exceeds the output possible voltage range, the DC link current i _dc matches the DC link current command value i _dc ^* and the output voltage becomes maximum regardless of the correction coefficient A _IVC. The shaft voltage command value vα ^* and the β-axis voltage command value vβ ^* are calculated and corrected by replacing them.

According to a seventh aspect of the present invention, there is provided the power converter according to the above invention, wherein the voltage control unit is defined when the amount of change in the corrected α-axis voltage command value vα ^* and β-axis voltage command value vβ ^* exceeds a specified value R _lim. An α-axis voltage command value vα ^* and a β-axis voltage command value vβ ^* that do not exceed the value R _lim and match the DC link current i _dc and the DC link current command value i _dc ^* are calculated.

According to an eighth aspect of the present invention, there is provided the power conversion device according to any one of the first to fifth aspects, wherein the voltage control unit predicts the value of the DC link current i _dc at the time of the next sampling and is obtained based on the predicted value. The α-axis voltage command value vα ^* and β-axis voltage command value vβ ^{* in} which the link current i _dc and the DC link current command value i _dc ^* match are calculated and replaced to calculate the α-axis voltage command value vα ^* and β-axis voltage. The command value vβ ^* is corrected.

Power conversion apparatus of the invention of claim 9, the voltage control unit in the invention, the DC link from the average value of the values of the DC link current i _dc at the value and the next DC link current i _dc at the next sampling the sampling The current i _dc is obtained.

According to a tenth aspect of the present invention, there is provided the power conversion device according to the eighth or ninth aspect, wherein the voltage control unit performs the α-axis currents iα and β of the motor at the next sampling, or at the next sampling and the next sampling. The value of the shaft current iβ is predicted, and based on the predicted value, the value of the DC link current i _dc at the next sampling, or the value of the DC link current i _dc at the next sampling and the DC link current at the next sampling The value of i _dc is calculated.

According to an eleventh aspect of the present invention, in each of the power converters according to the eleventh aspect, the voltage control unit has a small deviation between the α-axis current command value iα ^* and β-axis current command value iβ ^* and the α-axis current iα and β-axis current iβ. The α-axis voltage command value vα ^* and β-axis voltage command value vβ ^* calculated as described above are corrected so that the corrected α-axis voltage command value vα ^* and β-axis voltage command value vβ ^* have the same phase. It is characterized by that.

According to a twelfth aspect of the present invention, in each of the above-described power converters, the torque control unit is configured to rotate the α-axis current iα, β-axis current iβ, α-axis voltage command value vα ^* , β-axis voltage command value vβ ^*, and mechanical angle rotation. The output torque τ _inv of the inverter circuit is calculated from the angular frequency ω _rm, and feedback control is performed so that the output torque τ _inv matches the output torque command value τ _inv ^* .

According to the present invention, in a power converter for supplying electric power to a motor, a converter circuit that performs full-wave rectification of input AC, and a DC link that outputs a DC voltage, having a capacitor connected in parallel to the output of the converter circuit. And an inverter circuit that switches and converts the output of the DC link unit to AC and supplies the motor to the motor, and a control unit that controls the switching, and the control unit controls an operation amount that controls the speed of the motor. A speed control unit to be obtained, a torque control unit to obtain an operation amount for controlling the output torque τ _inv of the inverter circuit, and a voltage control unit to obtain an operation amount for controlling the α-axis current iα and β-axis current iβ of the motor, speed control unit, an operation amount for controlling the speed of the motor, synchronously by pulsing twice the power source frequency omega _s of the input AC, torque control unit controls the speed of the motor operating Generates an output torque command value tau _inv ^* of the inverter circuit from, based on the output torque command value tau _inv ^*, since to obtain the manipulated variable for controlling the output torque tau _inv of the inverter circuit, the control unit varies There is no need to consider the motor current frequency ω _re having a component, and the controller can be designed easily.

Further, the voltage control unit generates the α-axis current command value iα ^* and β-axis current command value iβ ^* of the motor on the stationary coordinate system based on the operation amount for controlling the output torque τ _inv of the inverter circuit, and the α The motor α-axis voltage command value vα ^* and β-axis voltage command value vβ ^* are set so that the deviation between the shaft current command value iα ^* and β-axis current command value iβ ^* and the α-axis current iα and β-axis current iβ is small. Since it is generated, disturbance response design can be performed.

Further, a DC link current command value i _dc ^* flowing in the inverter circuit is generated based on an operation amount for controlling the motor speed, and α is set so that the DC link current i _dc becomes equal to the DC link current command value i _dc ^*. Since the shaft voltage command value vα ^* and β-axis voltage command value vβ ^* are corrected, instantaneous power supply harmonic distortion can be suppressed, and the inconvenience that the convergence is deteriorated during transients such as acceleration / deceleration and torque change is compensated. can do.

  As a result, the present invention facilitates the design of the control unit for controlling the speed and torque and the disturbance response design of the motor current, and improves the convergence from the transient response, so that the motor flowing out to the power source side as a DC current is improved. Spatial harmonic components are reduced, and power supply harmonic regulations can be satisfied without providing a large-capacity reactor as in the prior art.

In this case, the operation amount for controlling the speed of the motor required by the speed control unit is converted into the torque dimension by dividing the input power command value of the inverter circuit by the rotational angular frequency ω _rm of the mechanical angle as in the invention of claim 2. The input torque command value τ _in ^* , and the voltage control unit generates the DC link current command value i _dc ^* based on the input torque command value τ _in ^* .

Further, the operation amount for controlling the output torque τ _inv of the inverter circuit obtained by the torque control unit is the q-axis current command value i _q ^* of the motor as in the third aspect of the invention, and the voltage control unit receives the q-axis current command value. The α-axis current command value iα ^* and the β-axis current command value iβ ^* are generated based on i _q ^* .

Then, the voltage control unit as in the invention of claim 4 can reduce the deviation between the α-axis current command value iα ^* and β-axis current command value iβ ^* and the α-axis current iα and β-axis current iβ by PI control. An axis voltage command value vα ^* and a β axis voltage command value vβ ^* are generated, and an α axis voltage command value vα ^* and a β axis voltage are set so that the DC link current i _dc is equal to the DC link current command value i _dc ^*. If the command value vβ ^* is corrected, it becomes possible to design a disturbance response with a relatively small amount of calculation, which is effective in a stable motor that is easy to converge from a transient response.

In addition to that, the voltage control unit performs feedback control with the F / B control unit including a control gain having a sine wave component of the motor current frequency ω _re of the motor as in the invention of claim 5, so that the α axis If the current iα and β-axis current iβ are made to follow the motor current frequency ω _re , the disturbance response and the target value response can be further improved, and the control is stable when the acceleration / deceleration rate is fast. This is effective in difficult motors.

Further, as in the sixth aspect of the invention, the voltage control unit uses the correction coefficient A _IVC calculated from the ratio between the DC link current command value i _dc ^* and the DC link current i _dc and the DC link current i _dc and the DC link. The α-axis voltage command value vα ^* and the β-axis voltage command value vβ ^* are corrected so that the current command value i _dc ^* matches, and the corrected α-axis voltage command value vα ^* and β-axis voltage command value vβ ^* are If more than the output voltage range, the correction coefficient a regardless of the _IVC, the DC link current i _dc and the DC link current command value i _dc ^* match, and the output voltage is maximized α-axis voltage value v? ^* If the correction is performed by calculating and replacing the β-axis voltage command value vβ ^* , a stable operation can be realized.

Furthermore, when the voltage control unit as in the invention of claim 7, alpha-axis voltage value v? ^* And the β-axis voltage value v? ^* The amount of change in the corrected exceeds a specified value R _lim, exceeds a specified value R _lim In addition, if the α-axis voltage command value vα ^* and β-axis voltage command value vβ ^{* in} which the DC link current i _dc and the DC link current command value i _dc ^* coincide are calculated, the phase current changes sharply. As a result, it is possible to effectively eliminate the disadvantage that the resonance current is generated in the input current.

The voltage control unit predicts the value of the DC link current i _dc at the time of the next sampling, and the DC link current i _dc and the DC link current command value i _dc obtained based on the predicted value. ^If the α-axis voltage command value vα ^* and β-axis voltage command value vβ ^* with the same ^* are calculated and replaced to correct the α-axis voltage command value vα ^* and β-axis voltage command value vβ ^* , The control error of the DC link current i _dc generated due to the difference between the sampling value and the average value between the samplings can be greatly reduced, and the parasitic component existing in the input unit and the DC link unit The resonance current of the resonance circuit by the capacitor can be reduced.

In this case, as the voltage control unit of the invention of claim 9, the DC link current i _dc from the average value of the values of the DC link current i _dc at the value and the next DC link current i _dc at the next sampling the sampling By determining, the control error can be effectively reduced.

Further, as in the invention of the tenth aspect, the values of the α-axis current iα and β-axis current iβ of the motor at the next sampling, or at the next sampling and at the next sampling are predicted, and based on the predicted values. voltage control unit Te value of the DC link current i _dc at the next sampling, or by calculating the value of the DC link current i _dc at the value and the next DC link current i _dc at the next sampling the sampling, The value of each DC link current i _dc can be accurately calculated.

Further, as in the invention of claim 11, the voltage control unit is calculated so that the deviation between the α-axis current command value iα ^* and β-axis current command value iβ ^* and the α-axis current iα and β-axis current iβ is small. Correction without any problem by correcting the α-axis voltage command value vα ^* and β-axis voltage command value vβ ^* and the corrected α-axis voltage command value vα ^* and β-axis voltage command value vβ ^{* to} be in phase. Will be able to do.

Further, as in the twelfth aspect of the invention, the torque control unit converts the α-axis current iα, the β-axis current iβ, the α-axis voltage command value vα ^* , the β-axis voltage command value vβ ^*, and the rotational angular frequency ω _{rm of the} mechanical angle from the inverter. If the output torque τ _{inv of the} circuit is calculated and feedback control is performed so that the output torque τ _inv matches the output torque command value τ _inv ^* , the inverter output power becomes a fundamental component of the input power. Accordingly, the distortion components of the input current and the motor current can be reduced.

It is a block diagram of the power converter device of the Example to which this invention is applied (Example 1). It is a figure which shows DC link current _idc (average value, estimated value) and DC link current command value _idc ^* when instantaneous voltage control is not performed in the power converter of FIG. It is a figure which shows DC link current _idc (average value, estimated value) and DC link current command value _idc ^* for demonstrating the effect | action of the instantaneous voltage control part of the voltage control part of the power converter device of FIG. It is a block diagram of the power converter device of the other Example to which this invention is applied (Example 2). It is a block diagram of the power converter device of another another Example to which this invention is applied (Example 3). It is a block diagram of the power converter device of another another Example to which this invention is applied (Example 4). A DC link current i _dc (average value, estimated value) and a DC link current command value i _dc ^* for explaining the operation of the instantaneous voltage control unit of the voltage control unit of the power conversion device of FIG. FIG. FIG. 2 is a diagram illustrating an input current i _in and an input current command value i _in ^* , a DC link current i _dc, and a DC link current command value i _dc ^* when the voltage control unit of the power conversion device in FIG. 1 does not perform predictive control. . FIG. 2 is a diagram illustrating an input current i _in and an input current command value i _in ^* , a DC link current i _dc and a DC link current command value i _dc ^* when the voltage control unit of the power conversion device in FIG. 1 performs predictive control. .

  Hereinafter, embodiments of the present invention will be described in detail with reference to the drawings.

(1) Power converter 1
FIG. 1 is a block diagram showing a configuration of a power conversion device 1 according to an embodiment of the present invention. The power conversion device 1 of this embodiment includes a converter circuit 2, a DC link unit 3, an inverter circuit 4, and a control unit 6, and converts AC power supplied from a single-phase AC power source 7 to a predetermined frequency. The electric power is converted into electric power and supplied to the motor 8. The motor 8 of the embodiment is an IPMSM (Interior Permanent Magnet Synchronous Motor) that drives a compressor that constitutes a refrigerant circuit of the refrigeration apparatus, and is driven by a voltage command generated by the control unit 6.

(2) Converter circuit 2
The converter circuit 2 is connected to an AC power source 7 and rectifies AC (input AC) from the AC power source 7 into DC. In this embodiment, the converter circuit 2 is composed of a diode bridge circuit in which a plurality (four) of diodes D1 to D4 are connected in a bridge shape. By these diodes D1 to D4, the AC voltage of the AC power supply 7 is full-wave rectified and converted to a DC voltage.

(3) DC link part 3
The DC link unit 3 includes a capacitor 9. The capacitor 9 is connected in parallel to the output of the converter circuit 2, and a DC voltage (DC link voltage) V _dc generated at both ends of the capacitor 9 is connected to an input node of the inverter circuit 4. The capacitor 9 has a capacitance that can smooth only a ripple voltage (voltage fluctuation) generated corresponding to a switching frequency when a switching element (described later) of the inverter circuit 4 performs a switching operation.

That is, the capacitor 9 is a small-capacitance capacitor that does not have a capacitance that smoothes the voltage rectified by the converter circuit 2 (voltage that varies according to the power supply voltage). In the embodiment, it is assumed that the capacity of the electrolytic capacitor necessary for smoothing the output of a general converter circuit is approximately 1/100. Therefore, the DC link voltage V _dc output from the DC link unit 3 pulsates. For example, a film capacitor can be used as the capacitor 9.

(4) Inverter circuit 4
The inverter circuit 4 has an input node connected in parallel to the capacitor 9 of the DC link unit 3, and is configured to switch the output of the DC link unit 3 to convert it into a three-phase AC and supply it to the motor (IPMSM) 8. ing. The inverter circuit 4 according to the embodiment is configured by bridge-connecting a plurality of switching elements. The inverter circuit 4 includes six switching elements S1 to S6 in order to output a three-phase alternating current to the motor 8.

Specifically, the inverter circuit 4 includes three switching legs in which two switching elements are connected in series with each other, and the middle points of the upper arm switching elements S1 to S3 and the lower arm switching elements S4 to S6 in each switching leg. Are connected to the coils of the respective phases (u-phase, v-phase, w-phase) of the motor 8. In addition, free-wheeling diodes D5 to D10 are connected in reverse parallel to the switching elements S1 to S6, respectively. Then, the inverter circuit 4 switches the DC link voltage V _dc input from the DC link unit 3 by the ON-OFF operation of these switching elements S1 to S6, converts it into a three-phase AC voltage, and supplies it to the motor 8 To do.

(5) Control unit 6
Next, the control unit 6 performs switching (ON-OFF operation) in the inverter circuit 4 so that the output torque τ _inv of the inverter circuit 4 pulsates in synchronization with twice the frequency ω _s (power frequency) of the input AC. To control. The control unit 6 according to the embodiment includes a speed control unit 11, a torque control unit 12, a voltage control unit 13, and a switching control unit 14.

(5-1) Speed control unit 11
The speed control unit 11 obtains an operation amount for controlling the speed of the motor 8. Specifically, the speed control unit 11 includes a PI calculation unit 16, a subtracter 17, a square waveform generation unit 18, and a multiplier 19. The speed control unit 11 generates an input torque command value τ _in ^* of the inverter circuit 4 as an operation amount for controlling the speed of the motor 8 according to the output of the PI calculation unit 16 and the output of the square waveform generation unit 18. . This input torque command value τ _in ^* pulsates in synchronization with twice the input AC frequency ω _s .

Specifically, in the embodiment, the subtractor 17 of the speed control unit 11 has a rotation angle frequency ω _rm of the mechanical angle of the motor 8 (estimated value in the embodiment, which may be a measured true value; hereinafter the same) and a rotation angle. _Find the deviation from the frequency command value ω _rm ^* . The PI calculation unit 16 performs a proportional / integral calculation (PI calculation) on the deviation obtained by the subtractor 17 and outputs the result. Furthermore, the square wave generator 18, the input voltage v _in, and input via the PLL circuit 21 generates and outputs a waveform sin [theta _in ² obtained by squaring the input AC. The multiplier 19 multiplies the output of the PI calculation unit 16 and the output of the square waveform generation unit 18 and outputs the result as an input torque command value τ _in ^* . As a result, the input torque command value τ _in ^* pulsates in synchronization with twice the frequency ω _s of the input AC.

(5-2) Torque control unit 12
Next, the torque control unit 12 includes subtractors 22 and 40 and a q-axis current command value generation unit 23. In the torque control unit 12, the q-axis current command value (q-axis) of the motor 8 is used as an operation amount for controlling the output torque τ _inv of the inverter circuit 4 from the input torque command value τ _in ^* output by the speed control unit 11. Current command value) i _q ^* is generated.

Specifically, in the embodiment, the subtractor 22 of the torque control unit 12 divides the input torque command value τ _in ^* output from the speed control unit 11 and the power in the capacitor 9 by the rotational angular frequency ω _rm of the mechanical angle. The deviation from the capacitor torque τ _c converted into the torque dimension (estimated value in the embodiment, may be a true value actually measured; hereinafter the same) is obtained. And it outputs as command value (output torque command value) τ _inv ^* of the output torque τ _inv of the inverter circuit 4.

Here, the input power p _in and the power p _c of a capacitor 9, and between the output power p _inv of the inverter circuit 4, a relationship of the following formula (I).
p _in = p _c + p _inv (I)
This is true even when each value is divided by the rotational angular frequency ω _rm of the mechanical angle and converted to the torque dimension, and therefore has the relationship of the following formula (II). Note that τ _in is the input torque.
τ _in = τ _c + τ _inv (II)
Then, the input torque command value tau _in ^* because of those produced without considering the current flowing through the capacitor 9 of the DC link section 3, the output torque command value tau _inv ^*, the input torque command value tau _in ^* the capacitor torque It is preferable to use a value compensated by τ _c . Therefore, in this embodiment, as described above, the subtractor 22 compensates the output torque command value τ _inv ^* to a value obtained by subtracting the capacitor torque τ _c .

Further, the subtracter 40 subtracts τ _off from the output torque command value τ _inv ^* , and outputs it as a command value (motor output torque command value) τ _mtr ^* of the motor 8. The τ _off is a value obtained by dividing the time loss of the power loss from the inverter circuit 4 to the motor 8 and the magnetic energy stored in the winding by the rotation angle frequency ω _rm of the mechanical angle and converting it to the torque dimension. The motor output torque command value τ _mtr ^* is obtained from the relationship of the following formula (III).
τ _inv ^* = τ _mtr ^* + τ _off (III)
As a result, the motor output torque τ _mtr can be controlled more accurately.

Q-axis current command value generating portion 23 of the torque control section 12, the influence of i _d with respect to the subtracter motor output torque command value 40 is output) tau _mtr ^* assumes the negligible pole pairs P of the motor 8 And the armature linkage magnetic flux φα by the permanent magnet is divided to generate a q-axis current command value i _q ^* . If the influence of i _d cannot be ignored by the motor, the q-axis current command value i _q ^* may be derived using an equation that takes i _d into account.

(5-3) Voltage control unit 13
Next, the voltage control unit 13 of this embodiment is configured to include an αβ axis PI control unit 13A, an instantaneous voltage control unit 13B, and an αβ-uvw conversion unit 38.

(5-3-1) αβ axis PI control unit 13A
The αβ-axis PI control unit 13A of the voltage control unit 13 of the embodiment includes a dq-αβ conversion unit 24, subtractors 26 and 27, and PI calculation units 31 and 32. The dq-αβ conversion unit 24 of the αβ axis PI control unit 13A converts the d axis current command value i _d ^* and the q axis current command value i _q ^* output from the torque control unit 12 into the α axis on the stationary coordinate system. The current command value iα ^* and the β-axis current command value iβ ^* are converted and output. The subtractor 26 is an α-axis current iα obtained from the α-axis current command value iα ^* output from the dq-αβ converter 24 and the phase current of the motor 8 (estimated value in the embodiment. Actually measured value may be used. The same shall apply hereinafter). Further, the PI calculation unit 31 performs a proportional / integral calculation (PI calculation) on the deviation obtained by the subtractor 26, thereby reducing the deviation between the α-axis current command value iα ^* and the α-axis current iα. Thus, the α-axis voltage command value vα ^* is generated and output.

On the other hand, the subtractor 27 is a β-axis current iβ obtained from the β-axis current command value iβ ^* output from the dq-αβ converter 24 and the phase current of the motor 8 (estimated value in the embodiment. Actually measured value may be used. The same shall apply hereinafter). Further, the PI calculation unit 32 performs a proportional / integral calculation (PI calculation) on the deviation obtained by the subtractor 27, thereby reducing the deviation between the β-axis current command value iβ ^* and the β-axis current iβ. Thus, the β-axis voltage command value vβ ^* is generated and output.

In the control of the dq coordinate system, the disturbance response cannot be designed, and the spatial harmonic component of the motor 8 appears as the motor current. Therefore, when a large-capacity electrolytic capacitor or reactor is not used, the motor current becomes a direct current (DC link current i _dc ) and returns to the power supply current (input current i _in ). May appear as power harmonics.

On the other hand, as in the embodiment, the αβ axis PI control unit 13A performs a proportional / integral operation on the deviation between the α axis current command value iα ^* and the α axis current iα, thereby obtaining the α axis current command value iα ^* . By generating an α-axis voltage command value vα ^* so that the deviation from the α-axis current iα is small, and performing a proportional / integral operation on the deviation between the β-axis current command value iβ ^* and the β-axis current iβ generates a beta axis current value i.beta ^* and beta-axis current i.beta beta-axis voltage command value so that the deviation becomes smaller with v? ^*, by the outputs respectively, able to operate disturbance response design, The disturbance current due to the spatial harmonics is suppressed, and the DC link current i _dc at the steady state is adjusted. When a large-capacity electrolytic capacitor or reactor is not used, the DC link current i _dc flows out to the input current i _in , but by suppressing this by the αβ axis PI control unit 13A, the power harmonic regulation can be satisfied. It becomes like this.

Here, when the α-axis voltage command value vα ^* and the β-axis voltage command value vβ ^* are generated by the αβ-axis PI control unit 13A as described above, the target value responsiveness deteriorates during a transient such as acceleration / deceleration or torque change. The convergence from the transient response may deteriorate. This is shown in FIG. In FIG. 2, the solid line indicates the DC link current command value i _dc ^* , and the broken line indicates the DC link current i _dc . As shown in this figure, the DC link current i _dc may not coincide with the DC link current command value i _dc ^* during the transition. In that case, when a large-capacity electrolytic capacitor or reactor is not used, the input current i _in flows rapidly, and appears as a high-frequency resonance.

(5-3-2) Instantaneous voltage control unit 13B
Therefore, in the present invention, the voltage controller 13 is provided with the following instantaneous voltage controller 13B, and the DC link current i _dc is constantly monitored at the carrier frequency, and the DC link current command value i _dc ^* and the DC link current i _dc are If there is a deviation between the two, a multiplier 33, 34, which will be described later, compensates by instantaneously changing the voltage. Specifically, the instantaneous voltage control unit 13 </ b> B includes multipliers 33, 34, 36, 37, and 39 and a subtracter 28. The multiplier 39 of the instantaneous voltage control unit 13B multiplies the input torque command value τ _in ^* output from the speed control unit 11 by the rotation angle frequency ω _rm of the mechanical angle, thereby converting it to the input power command value p _in ^*. ,Output. The multiplier 36, the input power command value p _in ^* the multiplier 39 is output, multiplying the absolute value of 1 / v _in, that is, the absolute value of the input voltage v _in the input power command value p _in ^* By dividing, the absolute value of the input current command value i _in ^* is output.

It should be noted that the input voltage v _in this case may be the input voltage command value v _in ^*. The subtractor 28 determines the DC link capacitor current i _c (average value, which is an average value in the embodiment, from the absolute value of the input current command value i _in ^* . ) Is subtracted to obtain and output a DC link current command value i _dc ^* (average value) as a deviation between them. The calculation of the DC link current command value i _dc ^* (average value) in the subtractor 28 is based on the relationship of the following formula (IV).
Absolute value of i _in ^* = i _dc ^* (average value) + i _c (average value) (IV)

The multiplier 37 adds the DC link current command value i _dc ^* (average value) output from the subtractor 28 to 1 / i _dc (average value. In the embodiment, an estimated value. The measured true value may be used. ), That is, the DC link current command value i _dc ^* (average value) is divided by the DC link current i _dc (average value) to calculate and output the correction coefficient A _IVC .

The correction coefficient A _IVC is a ratio between the DC link current command value i _dc ^* (average value) and the DC link current i _dc (average value), and the DC link current command value i _dc ^* and the DC link current i _dc If there is no deviation (deviation), the correction coefficient A _IVC is 1. On the other hand, if the DC link current i _dc becomes larger than the DC link current command value i _dc ^{*, the} correction coefficient A _IVC becomes smaller. Conversely, if the DC link current i _dc becomes smaller than the DC link current command value i _dc ^{*, the} correction coefficient. A _IVC grows.

Here, the DC link current i _dc (average value) described above is obtained from the following equation (V).
i _dc (average value) = d _u i _u + d _v i _v + d _w i _w (V)
i _u , i _v , and i _w are three-phase currents, and d _u , d _v , and d _w are phase duty ratios determined by the three-phase voltage command values.
From this equation, it can be seen that the harmonics of the motor current and voltage command generated by the spatial harmonics generate DC link current (average value) harmonics. And in the power converter device 1 of an Example, since the small capacity | capacitance film capacitor is used as the capacitor | condenser 9, a direct current link current (average value) harmonic flows out to the power supply side, and a power supply harmonic arises.

(5-3-3) Correction of α-axis Voltage Command Value vα ^* and β-axis Voltage Command Value vβ ^{* In} multiplier 33 of instantaneous voltage control unit 13B, α output from PI calculation unit 31 of αβ-axis PI control unit 13A The shaft voltage command value vα ^* is multiplied by the correction coefficient A _IVC output from the multiplier 37 of the instantaneous voltage control unit 13B, and the multiplier 34 multiplies the β-axis voltage command value vβ ^* output from the PI calculation unit 32. The correction coefficient A _IVC output from the multiplier 37 is multiplied.

As described above, if the DC link current i _dc becomes larger than the DC link current command value i _dc ^{*, the} correction coefficient A _IVC becomes smaller, and conversely if the DC link current i _dc becomes smaller than the DC link current command value i _dc ^*. since the correction coefficient a _IVC increases, the correction coefficient a _IVC includes a DC link current command value i _dc ^* and the DC link current i _dc direction α axis voltage value to cancel the deviation between the v? ^* and the β-axis voltage command value vβ ^* _Is corrected and the DC link current i _dc (average value) is controlled to improve the power supply current waveform.

That is, in the multiplier 33 and the multiplier 34, the α-axis voltage command value vα ^* and the β-axis voltage command value vβ ^* are set so that the deviation between the DC link current command value i _dc ^* and the DC link current i _dc becomes small. It will be corrected. The broken line in FIG. 3 indicates the DC link current i _dc when the instantaneous voltage control unit 13B calculates the correction coefficient A _IVC and the multipliers 33 and 34 correct the α-axis voltage command value vα ^* and the β-axis voltage command value vβ ^*. The solid line indicates the DC link current command value i _dc ^* .

As is clear from this figure, the DC link current i _dc is equal to the DC link current command value i _dc ^* even during a transition, based on the correction value A _IVC calculated by the instantaneous voltage control unit 13B, compared to the case of FIG. It can be seen that the correction has been made. This is called instantaneous voltage control. Then, the αβ-uvw conversion unit 38 converts the α-axis voltage command value vα ^* and β-axis voltage command value vβ ^* corrected by the multipliers 33 and 34 of the instantaneous voltage control unit 13B to u, v, and w. It is converted into a phase voltage command value v _uvw ^* .

(5-4) Switching control unit 14
The voltage command value v _uvw ^* is input to the switching control unit 14, and the switching control unit 14 controls the ON / OFF operations of the switching elements S 1 to S 6 based on the value of the voltage command value v _uvw ^*. Generate a gate signal. Specifically, the switching control unit 14 performs PWM (Pulse Width Modulation) control on the inverter circuit 4 in synchronization with the carrier signal (triangular wave).

Here, the deviation of the alpha-axis current command value i.alpha ^* and alpha-axis current i.alpha, and, beta-axis current command value i.beta ^* and beta-axis current i.beta alpha-axis voltage command value so that the deviation becomes smaller with v? ^*, And When the β-axis voltage command value vβ ^* is calculated and converted into the voltage command value v _uvw ^* of each phase as it is and the switching control is performed, between the DC link current command value i _dc ^* and the DC link current i _dc during the transition to the deviation occurs, pulsed shift (deviation) is generated between the DC link voltage V _dc appearing on the terminals of the capacitor 9 and the input voltage v _in, the DC link capacitor current i _c flowing to the capacitor 9 increases When a parasitic inductance exists in the system, resonance is excited between the parasitic inductance and the capacitor 9 composed of a small-capacity film capacitor, and the input current i _in vibrates.

As in the embodiment, the voltage controller 13 corrects the α-axis voltage command value vα ^* and the β-axis voltage command value vβ ^* so that the DC link current i _dc becomes equal to the DC link current command value i _dc ^* . Deviations that occur during transition can be reduced.

(5-5) In this embodiment the operation of the power conversion apparatus 1, the speed control unit 11 of the control section 6, first, the input torque command value tau _in pulsating in synchronism with twice the power source frequency omega _s of the input AC ^{* Is} generated. Next, the torque control unit 12 generates the input torque command value tau _in ^* from the output torque command value tau _inv ^*, generates a q-axis current command value i _q ^* from the output torque command value tau _inv ^*.

Next, the voltage control unit 13 generates an α-axis voltage command value vα ^* and a β-axis voltage command value vβ ^* on the stationary coordinate system from the d-axis current command value i _d ^* and the q-axis current command value i _q ^*. To do. Then, alpha axis current value i.alpha ^* and β-axis current value i.beta ^* and alpha-axis current i.alpha and β-axis current alpha-axis voltage command value so that the deviation becomes smaller with i.beta v? ^* And β-axis voltage value v? ^* Is generated. Further, a DC link current command value i _dc ^* that flows to the inverter circuit 4 is generated from the input torque command value τ _in ^* , and a correction coefficient A that is a ratio between the DC link current command value i _dc ^* and the DC link current i _dc. By calculating the _IVC and multiplying the calculated correction coefficient A _IVC by the α-axis voltage command value vα ^* and the β-axis voltage command value vβ ^* , the DC link current i _dc becomes equal to the DC link current command value i _dc ^*. Thus, the α-axis voltage command value vα ^* and the β-axis voltage command value vβ ^* are corrected.

Then, gate signals of the switching elements S1 to S6 are generated using the corrected α-axis voltage command value vα ^* and β-axis voltage command value vβ ^* . By these gate signals, switching is performed in the inverter circuit 4, and electric power is supplied to the motor 8.

  With such control, the present invention facilitates disturbance response design, improves convergence from transient response, reduces the spatial harmonic component of the motor flowing out to the power source side as a DC link current, The power supply harmonic regulation can be satisfied without providing such a large capacity reactor. As a result, a significant cost reduction can be achieved.

  Next, FIG. 4 is a block diagram showing a configuration of a power conversion device 1 according to another embodiment to which the present invention is applied. In this figure, the same reference numerals as those in FIG. 1 indicate the same or similar functions.

In the power conversion device 1 of this embodiment, the torque control unit 12 of FIG. 1 is replaced with the torque control unit 12 of another embodiment. The torque control unit 12 in this case includes the subtractor 22 described above, a subtractor 42, and a torque controller 44. The torque control unit 12 of this embodiment also uses a command value (q-axis current command value of the motor 8 as an operation amount for controlling the output torque τ _inv of the inverter circuit 4 from the input torque command value τ _in ^* output by the speed control unit 11. q-axis current command value) i _q ^* is generated.

Even in specific this example the torque control section 12 of the subtractor 22, the input torque command value tau _in the speed control unit 11 outputs ^*, the rotation angular frequency of the mechanical angle power p _c in the capacitor 9 Divide by ω _rm to obtain the deviation from the value converted to the torque dimension from the capacitor torque τ _c . The output value τ _inv command value (output torque command value) τ _inv ^* of the inverter circuit 4 is output to the subtractor 42.

Then, the subtractor 42 calculates a deviation between the output torque command value τ _inv ^* and the output torque τ _inv . Here, the output torque τ _inv of the inverter circuit 4 is the α-axis current iα (estimated value), β-axis current iβ (estimated value), α-axis voltage command value vα ^* , β-axis voltage command value vβ ^*, and mechanical angle. From the rotation angular frequency ω _rm , it is obtained based on the relationship of the following formula (VI).
τ _inv = (vα ^* iα (estimated value) + vβ ^* iβ (estimated value)) ÷ ω _rm (VI)

Then, the torque controller 44 performs feedback control on the deviation obtained by the subtractor 42 so that the output torque τ _inv and the output torque command value τ _inv ^* coincide (the deviation is eliminated). ) Generate and output q-axis current command value i _q ^* .

Because of the foregoing as input torque command value tau _in ^* those generated without considering the current flowing through the capacitor 9 of the DC link section 3, the output torque command value tau _inv ^*, a capacitor input torque command value tau _in ^* It is preferable to use a value compensated by the torque τ _c .

Then, the torque controller 44 of the torque control unit 12 receives the inverter circuit 4 from the α-axis current iα, the β-axis current iβ, the α-axis voltage command value vα ^* , the β-axis voltage command value vβ ^*, and the rotational angular frequency ω _{rm of the} mechanical angle. of calculating the output torque tau _inv, and implementing feedback control so that the output torque tau _inv and the output torque command value tau _inv ^* matches generates a q-axis current command value i _q ^*.

Thus, the torque controller 44 performs torque dimension feedback control and outputs the q-axis current command value i _q ^* . Therefore, it is not necessary to consider the motor current frequency ω _re in the torque controller 44 as compared with the power feedback control, and the torque controller 44 can be easily designed. By carrying out this feedback control, the output power p _inv of the inverter circuit 4, it is possible to follow the more fundamental wave component of the input power p _in, to reduce the distortion component of the input current i _in and the motor current Will be able to.

  Next, FIG. 5 is a block diagram showing a configuration of a power conversion device 1 of still another embodiment to which the present invention is applied. In this figure, the same reference numerals as those in FIG. 4 indicate the same or similar functions. In the power conversion device 1 of this embodiment, the αβ axis PI control unit 13A in FIG. 4 is replaced with a sine wave tracking control unit 13C.

(6) Sinusoidal tracking control unit 13C
FIG. 5 shows a block diagram of the sine wave follow-up control unit 13C. The sine wave follow-up control unit 13C includes the above-described subtractors 26 and 27, F / B control units 46 and 47, PI control units 48 and 49, control stabilization units 51 and 52, and subtracters 53 and 54. ing.

The sine wave follow-up control unit 13C is basically proportional to and integrated with respect to the deviation between the α-axis current command value iα ^* and the α-axis current iα by the PI control unit 48, similarly to the αβ-axis PI control unit 13A described above. By performing the calculation, the α-axis voltage command value vα ^* is generated so that the deviation between the α-axis current command value iα ^* and the α-axis current iα is small, and the PI control unit 49 generates the β-axis current command value iβ ^* . A β-axis voltage command value vβ ^* is generated by performing proportional / integral calculation on the deviation from the β-axis current iβ so that the deviation between the β-axis current command value iβ ^* and the β-axis current iβ is reduced. However, the F / B control units 46 and 47 including a control gain having a sine wave component of the motor current frequency ω _re perform feedback control on the subtractors 26 and 27, so that the α-axis current iα and the β-axis current are iβ a sine wave tracking control to follow the motor current frequency ω _re (αβ-sin To run the control).

Further, the control stabilization units 51 and 52 multiply the α-axis current iα and the β-axis current iβ by a feedback gain f ₃ and subtract them from the outputs of the PI control units 48 and 49 by the subtractors 53 and 54 to stabilize the control. As a result, the α-axis voltage command value vα ^* and the β-axis voltage command value vβ ^* are made closer to a sine wave, thereby making the α-axis voltage command value vα ^* and the β-axis voltage command value vβ ^* closer to a sine wave.

The feedback gains f ₁ , f ₂ , and f ₃ in the PI control units 48 and 49 and the control stabilization units 51 and 52 in the block diagram are represented by the following formulas (VII), (VIII), and (IX).
f ₁ = L _d (ω _re ² −3ω _c ² ) (VII)
f ₂ = L _d (3ω _c ω _re ² −ω _c ³ ) (VIII)
f ₃ = 3L _d ω _c −R _a (IX)
L _d is the d-axis inductance of the armature winding of the motor 8, R _a is the armature winding resistance, ω _re is the motor current frequency of the motor 8, and ω _c is the pole of the current control system.

Since the motor current frequency ω _re of the motor 8 is included in the above formulas (VII) and (VIII), in the sinusoidal follow-up control, the gain must be made variable according to the electrical angular velocity of the motor 8. By performing such sine wave follow-up control, it is possible to improve the target value response to the command value and the disturbance response.

  Next, FIG. 6 is a block diagram showing a configuration of a power conversion device 1 of still another embodiment to which the present invention is applied. In this figure, the same reference numerals as those in FIG. 1 indicate the same or similar functions. In the power conversion device 1 of this embodiment, the instantaneous voltage control unit 13B in FIG. 1 is replaced with an instantaneous voltage control unit 13D in this case. Other configurations are the same as those in FIG.

For example, in the embodiment shown in FIG. 1, the instantaneous voltage control unit 13B multiplies the α-axis voltage command value vα ^* and β-axis voltage command value vβ ^* output from the PI calculation units 31 and 32 of the αβ-axis PI control unit 13A. The correction is performed by multiplying the correction coefficient A _IVC by the devices 33 and 34. That is, the correction coefficient A _IVC acts as a variable gain with respect to the α-axis voltage command value vα ^* and β-axis voltage command value vβ ^* output from the PI calculation units 31 and 32. Therefore, for stable operation, the corrected α-axis voltage command value vα ^* and β-axis voltage command value vβ ^* can be output by the power conversion device 1 (hereinafter referred to as an outputable voltage range). Must be restricted so as not to exceed.

Further, in the correction of the α-axis voltage command value vα ^* and the β-axis voltage command value vβ ^* performed by the instantaneous voltage control unit 13B, output voltage vectors (α-axis voltage command value vα ^* and β-axis voltage command value vβ ^* ) May change significantly in one control cycle. In that case, the phase current changes sharply, and the error of the current generated in the DC link current i _dc changes in a pulse shape. A so-called electrolytic capacitorless inverter resonates with an inductance component parasitic in the system and is in a state where the input current is likely to vibrate. Therefore, when the aforementioned pulse-like control error occurs, a resonance current is generated in the input current. Therefore, it is necessary to limit the amount of change in the voltage vector (α-axis voltage command value vα ^* and β-axis voltage command value vβ ^* ) that changes in one control cycle.

(7) Instantaneous voltage controller 13D
The instantaneous voltage control unit 13D of this embodiment is configured to solve the above problem. Specifically, the instantaneous voltage control unit 13D includes an output voltage correction unit 56 in addition to the multipliers 36 and 39 and the subtractor 28 having the same functions as those described above (FIG. 1). In this case as well, the multiplier 39 multiplies the input torque command value τ _in ^* output from the speed control unit 11 by the rotation angle frequency ω _rm of the mechanical angle to convert it into the input power command value p _in ^* . Output. The multiplier 36, the input power command value p _in ^* the multiplier 39 is output, multiplying the absolute value of 1 / v _in, that is, the absolute value of the input voltage v _in the input power command value p _in ^* By dividing, the absolute value of the input current command value i _in ^* is output.

(7-1) Output voltage correction unit 56
Further, the subtractor 28 determines the DC link capacitor current i _c (average value, estimated value in the embodiment. However, it may be a measured true value from the absolute value of the input current command value i _in ^* . , The same) is subtracted to obtain and output a DC link current command value i _dc ^* (average value) as a deviation thereof, and input it to the output voltage correction unit 56. The output voltage correction unit 56 further receives an α-axis current iα and a β-axis current iβ of the motor 8. The output voltage correction unit 56 receives the α-axis voltage command value vα ^* and the β-axis voltage command value vβ ^* output from the PI calculation units 31 and 32 of the αβ-axis PI control unit 13A. The output is input to the αβ-uvw conversion unit 38.

(7-2) Instantaneous voltage control when exceeding the output possible voltage range Here, the DC link current i _dc is represented by the following formula (X). Note that vα _n ^* is a normalized α-axis voltage command value, and vβ _n ^* is a normalized β-axis voltage command value. Here, the normalization is a process of dividing the voltage command value by the DC link voltage V _dc and multiplying by 2.
That is, vα _n ^* = 2vα ^* / V _dc and vβ _n ^* = 2vβ ^* / V _dc .

Then, when a DC link current command value which is a desired DC link current i _dc is substituted into the equation (X) as i _dc ^* and arranged for vβ _n ^* , the equation (XI) is obtained. This equation (XI) is an equation of a line in the normalized voltage vector space, and no matter which point on the line represented by the equation (XI) is arranged within a range where overmodulation does not occur, The desired DC link current command value i _dc ^* is obtained. Hereinafter, the straight line represented by the formula (XI) is referred to as an isoelectric line.

The instantaneous voltage control unit 13D also calculates the correction coefficient A _{IVC in the same} manner as the above-described instantaneous voltage control unit 13B, and outputs the α-axis voltage command values vα ^* and β output from the PI calculation units 31 and 32 of the αβ-axis PI control unit 13A. The shaft voltage command value vβ ^* is basically multiplied by the calculated correction coefficient A _IVC so as to be corrected so as to be arranged on the equicurrent line (the same applies to the instantaneous voltage control unit 13B). Thus, the corrected α axis voltage value v? ^* And β-axis voltage value v? ^* Is, .alpha..beta axis PI controller α-axis voltage command value PI calculation unit 31 is output 13A v? ^* And β-axis voltage The phase is the same as the command value vβ ^* .

However, as described above, the corrected α-axis voltage command value vα ^* and β-axis voltage command value vβ ^* must not exceed the output possible voltage range of the power converter 1. Here, the output possible voltage range of the power conversion device 1 is represented by a voltage limit circle of the formula (XII). In this case, the output voltage correction unit 56 determines that the α-axis voltage command value vα ^* and β-axis voltage command value vβ ^* corrected by the correction coefficient A _IVC do not satisfy the equation (XII). Rather than multiplying the command value vα ^* and the β-axis voltage command value vβ ^* by the correction coefficient A _IVC , the values derived from the equations (XIII) and (XIV) are used as the α-axis voltage command value vα ^* and the β-axis voltage command. Let it be the value vβ ^* . In other words, the α-axis voltage command value vα ^* and the β-axis voltage command value vβ ^* output from the PI calculation units 31 and 32 are corrected by replacing them with the values calculated by the equations (XIII) and (XIV). Is set as an α-axis voltage command value vα ^* and a β-axis voltage command value vβ ^*, and is output to the αβ-uvw converter 38.

That is, in this case, the output voltage correction unit 56 of the instantaneous voltage control unit 13D arranges the voltage vector at the intersection of the voltage limit circle (output possible voltage range) represented by the above formula (XII) and the above-described isocurrent line. To do. The intersection of the voltage limiting circle and the isocurrent line is the above formula (XIII) and formula (XIV). As apparent from these equations, two intersections appear, but the one closer to the original voltage vector (α-axis voltage command value vα ^* and β-axis voltage command value vβ ^* output from the PI calculation units 31 and 32). The values are selected and set as an α-axis voltage command value vα ^* and a β-axis voltage command value vβ ^* and output to the αβ-uvw conversion unit 38.

As described above, the output voltage correction unit 56 of the instantaneous voltage control unit 13D of the present embodiment also _{uses the} correction coefficient A _IVC calculated from the ratio between the DC link current command value i _dc ^* and the DC link current i _dc. The α-axis voltage command value vα ^* and the β-axis voltage command value vβ ^* are corrected so that the link current i _dc and the DC link current command value i _dc ^* coincide with each other. The corrected α-axis voltage command value vα ^{When the *} and β-axis voltage command value vβ ^* exceeds the outputable voltage range (voltage limit circle in the formula (XII)), the DC link current i _dc and the DC link current command value i _dc are not used regardless of the correction coefficient A _IVC. ^* is (located on the equal current line) matched, and the output voltage (the voltage limiting circle) the maximum α calculates the axis voltage value v? ^* and the β-axis voltage value v? ^*, replace it So that stable operation can be realized. It becomes so that.

(7-3) Change Limit Control Further, as described above, it is output in the correction of the α-axis voltage command value vα ^* and the β-axis voltage command value vβ ^* performed by the instantaneous voltage control unit 13B of the embodiment of FIG. Voltage vectors (α-axis voltage command value vα ^* and β-axis voltage command value vβ ^* ) may change greatly in one control cycle. In that case, the phase current changes sharply, and the error of the current generated in the DC link current i _dc changes in a pulse shape. A so-called electrolytic capacitorless inverter resonates with an inductance component parasitic in the system and is in a state where the input current is likely to vibrate. Therefore, when the aforementioned pulse-like control error occurs, a resonance current is generated in the input current. Therefore, it is necessary to limit the amount of change in the voltage vector (α-axis voltage command value vα ^* and β-axis voltage command value vβ ^* ) that changes in one control cycle. Therefore, the output voltage correction unit 56 of the instantaneous voltage control unit 13D in this case is a voltage vector (α-axis voltage command value vα ^* and β-axis voltage command value vβ ^* ) that changes in one control cycle as described below. Limit the amount of change.

The output voltage correction unit 56 of this embodiment is provided with a constant specified value R _lim, voltage vector as will not exceed the specified value R _lim voltage variation magnitude (norm) exceeds this specified value R _lim To correct. Moreover, since it is necessary to arrange on the equal current line when correcting, a voltage vector is arranged at the intersection of the circle of the specified value R _lim and the equal current line. The following formulas (XV) and (XVI) indicate the intersections between the circle of the specified value R _lim and the isoelectric lines.

In the equations (XV) and (XVI), [k] means the current control cycle, and [k-1] means one control cycle before. Again, the specified value intersection between the circle and the equal current lines R _lim is to appear two, v? _Nlim of intersection closer to the voltage vector before the correction ^* [k], select vβ _nLim ^* [k] The α-axis voltage command value vα ^* and the β-axis voltage command value vβ ^* are output to the αβ-uvw converter 38.

As described above, the output voltage correction unit 56 of the instantaneous voltage control unit 13D of the voltage control unit 13 has the corrected amounts of change in the α-axis voltage command value vα ^* and the β-axis voltage command value vβ ^* exceeding the specified value R _lim . In this case, the α-axis voltage command value vα ^* and the β-axis voltage command value vβ ^* that do not exceed the specified value R _lim and the DC link current i _dc and the DC link current command value i _dc ^* match are calculated and output. As a result, it is possible to effectively eliminate the inconvenience that the phase current changes sharply and a resonance current is generated in the input current.

In FIG. 7, the broken line indicates the correction coefficient A _IVC calculated by the instantaneous voltage control unit 13D in this case, corrects the α-axis voltage command value vα ^* and the β-axis voltage command value vβ ^* (instantaneous voltage control), and then outputs them. A DC link current i _dc when the voltage correction unit 56 performs the limit control of the change amount of the α-axis voltage command value vα ^* and the β-axis voltage command value vβ ^* described above, and the solid line indicates the DC link current command value i. _dc ^* is shown.

As is clear from this figure, the change amount limiting control by the output voltage correction unit 56 is controlled so that the DC link current i _dc matches the DC link current command value i _dc ^* further than in the case of FIG. I understand that.

  Next, still another embodiment of the control by the instantaneous voltage control unit 13D shown in FIG. 6 will be described. In addition, since the whole block diagram of the power converter device 1 in this case is the same as that of FIG. 6, it demonstrates referring FIG. 6 similarly to the case of the said Example (Example 4).

In the above embodiment (embodiment 4), the instantaneous voltage control unit 13D is set to a desired DC link current i _dc based on the sampling value of the phase current that continuously changes (value sampled in the control cycle described above). The voltage vector is manipulated in an FF (feed forward) manner. However, the phase current continuously changes after the voltage vector is determined, and a difference occurs between the average value and the sampling value during sampling. Thereby, in the instantaneous voltage control based on the sampling value of the phase current, a pulse-like control error occurs in the average value of the DC link current i _dc .

  Further, in the change amount limiting control of the above-described embodiment (embodiment 4), there is a problem that the phase current control becomes unstable because the time change amount of the voltage vector is limited.

Therefore, in this embodiment, the output voltage correction unit 56 of the instantaneous voltage control unit 13D predicts the current at the next sampling of the phase current by the discrete equation of the motor (IPMSM) 8, and the next sampling value (prediction) from the current sampling value. DC link current i _dc is controlled based on the average value up to (value).

(8) Predictive control of DC link current based on IPMSM discrete equation The instantaneous voltage control of the above-described embodiment (embodiment 4) was control based on the formula (X). Strictly describing this equation (X) with respect to the sampling time series, the following equation (XVII) is obtained.

In equation (XVII), i _dc [k] is the instantaneous value of the DC link current at time k (current sampling in the control cycle), and iα [k] and iβ [k] are motors on the αβ axis at time k. The instantaneous values of current, vα _n ^* [k] and vβ _n ^* [k] are normalized voltage vectors at time k. The instantaneous voltage control of the above-described embodiment (embodiment 4) is in a state in which only the instantaneous value at the time k of the DC link current i _dc can be controlled because the voltage vector is arranged on the equicurrent line based on this equation. .

  Here, the output voltage correction unit 56 of the instantaneous voltage control unit 13D of this embodiment uses an equation that is established for the average value of the DC link current idc using two sampling points, as shown in Equation (XVIII). Then, a predicted equivalent current curve that is also established with respect to the average value of the DC link current idc is derived.

Note that i _dc [k + 1] is expressed by formulas (XIX) and (XX), iα [k + 1] and iβ [K + 1], and vα _n ^* [k + 1] and vβ _n ^* [k + 1] are _expressed in formula (XVII). I _dc [k + 3] is calculated by substituting iα [k + 2], iβ [K + 2], vα _n ^* [k + 2], and vβ _n ^* [k + 2] obtained by the equations (XXI) and (XXII). It is calculated by substituting into the formula (XVII).

In formula (XVIII), taking into account that there is a delay of 1 sampling from when the output voltage is determined until it is actually output, in order to calculate the voltage at [k + 1] (after one control period), a total of 1 sampling The formula is advanced. Here, iα [k + 1] and iβ [k + 1] is the future value, and i.alpha [k] and i.beta [k], the discrete equations of the motor (IPMSM) 8 than v? _N [k] and v? _N [k] Can be obtained. That is, iα [k + 1] and iβ [k + 1] are determined by the above formulas (XIX) and (XX).

For iα [k + 2] and iβ [k + 2], vα _n [k + 1] and vβ _n [k + 1] calculated by the discrete equation of the motor (IPMSM) 8 based on iα [k + 1] and iβ [k + 1]. It is a function. That is, iα [k + 2] and iβ [k + 2] are determined by the above formulas (XXI) and (XXII). The coefficients A _{11 to} A ₂₂ and b _{11 to} b ₂₂ in the formulas (XIX) to (XXII) will be described later. Here, although the future value V _dc [k + 1] of the DC link voltage is required, it can be obtained by shifting the phase of the power source acquired by the PLL circuit 21 by one sampling.

Iα [k + 1] and iβ [k + 1], iα [k + 2] and iβ [k + 2] of Formula (XIX) to Formula (XXII) are substituted into Formula (XVIII). The average value idc of the DC link current _{i dc (i dc [k +} 1] + i dc [k + 2]) with respect to the / 2, and the DC link current command value i _dc ^* [k]. In current sampling time ([k] time), i _{for dc} ^* [k + 1] is not required, i _dc when [k] is substituted into equation (XVIII), and finally the DC link current at the current sampling i _dc The equation of [k] (predicted equicurrent curve) is represented by equation (XXIII).

Vβ _n ^* [k + 1] obtained from the equation (XXIII) is an equation of a curve drawn in the vicinity of the conventional isoelectric line. This curve is a curve in a normalized voltage space representing an average DC link current between samplings based on a current prediction at the time of next sampling by a discrete equation of the motor (IPMSM) 8. Further, since the obtained vβ _n ^* [k + 1] includes a positive and negative sign, the predicted isocurrent curve is two curves, but is a continuous single curve connecting them.

(8-1) Predictive control of DC link current within voltage limit The output voltage correction unit 56 derives a predicted equivalent current curve by prediction as described above, and then, similarly to the above-described embodiment (embodiment 4), A voltage vector is arranged at a point on the predicted equal current curve that is in phase with the output voltage vector from the αβ axis PI control unit 13A.

The α-axis voltage command value vα ^* and β-axis voltage command value vβ ^* output from the PI calculation units 31 and 32 of the αβ-axis PI control unit 13A are represented here as α-axis voltage command value vα ^* 1 and β-axis voltage command, respectively. Assuming the value vβ ^* 1, the in-phase line of the output voltage from the αβ-axis PI control unit 13A is expressed by the following equation (XXIX).

What is necessary is just to obtain | require the intersection of the same phase line with this output voltage from (alpha) (beta) axis | shaft PI control part 13A, and the prediction equivalent current curve of Formula (XXIII), and an intersection will become like Formula (XXX). vβ _n ^* [k + 1] is obtained from the equation (XXX), and vα _n ^* [k + 1] is obtained from the relationship of the equation (XXIX). That is, the output voltage correction unit 56 of this embodiment uses the α-axis voltage command value (in this case, vα ^* 1) and the β-axis voltage command value (in this case, vβ ^* 1) output from the αβ-axis PI control unit 13A. Expressions (XXIX) and (XXX) are calculated, and the α-axis voltage command value vα ^* 1 and β-axis voltage command value vβ ^* 1 are obtained by replacing the values calculated by these expressions (XXIX) and (XXX). The corrected values are set as an α-axis voltage command value vα ^* and a β-axis voltage command value vβ ^*, and are output to the αβ-uvw conversion unit 38.

That is, the output voltage correction unit 56 uses the α-axis voltage command value vα ^* 1 and β-axis voltage command value vβ ^* 1 output from the αβ-axis PI control unit 13A, and uses the α-axis voltage command value vα ^* 1 and the β-axis. The α-axis voltage command value vα ^* and β-axis voltage command value vβ ^* having the same phase as the voltage command value vβ ^* 1 are derived, corrected by replacement, and output to the αβ-uvw conversion unit 38.

  Here, the intersection of the in-phase line and the predicted equal current curve is shown in the formula (XXX), but the case where the intersection exceeds the voltage limit or the voltage correction amount is excessive is the same as in the above-described embodiment. It is necessary to select the intersection of the predicted isocurrent curve and the voltage limit graphic (voltage limit circle in the fourth embodiment). Next, the intersection of the voltage limiting hexagon inscribed by the voltage limiting circle and the predicted isocurrent curve will be described.

(8-2) Predictive control of DC link current when voltage is saturated Similar to the instantaneous voltage control of the above-described embodiment (embodiment 4), the αβ axis is also used in predictive control of the DC link current i _dc using the predicted _isocurrent curve. The output of the PI controller 13A is corrected to control the DC link current i _dc . For this reason, in order to ensure the stability of the current control system, or when voltage saturation occurs due to correction, a voltage vector is arranged at the intersection of the voltage limit graphic and the predicted equal current curve as in the case of the instantaneous voltage control.

  In the instantaneous voltage control of the above-described embodiment (embodiment 4), the voltage vector is arranged at the intersection of the isocurrent line and the voltage limit circle (inscribed circle of the voltage limit hexagon). Since a practical analytical solution cannot be obtained with the current curve and the circle, a voltage vector is arranged at the intersection with the voltage limiting hexagon. Here, each side of the voltage limiting hexagon is expressed by an equation of six straight lines having intercepts, and the intersection point with the predicted equivalent current curve is calculated. Each side of the voltage limiting hexagon is generalized by the following linear equation (XXXI).

  1 to 6 are substituted for n in the formula (XXXI), each becomes a straight line of 6 sides, and forms a hexagonal side. The intersection of the generalized straight line equation shown in equation (XXXII) and the predicted isocurrent curve is shown in equation (XXXIII). Each of the formulas (XXXIII) is calculated for six sides, and an intersection having a solution inside the corner apex of the hexagon. Among the plurality of calculated intersections, the intersection closest to the output from the αβ axis PI control unit 13A is selected.

(8-3) IPMSM Discrete Equation The control of the DC link current i _dc using the predicted isocurrent curve is based on the discrete equation of the motor (IPMSM) 8. The IPMSM discrete equation used here is an equation obtained by discretizing the IPMSM state equation with the 0th-order hold of the sampling time T _s . The equation of state of IPMSM is shown in Formula (XXXIV). When the equation (XXXIV) is discretized by the 0th-order hold of the sampling time T _s , the equation (XXXV) is obtained.

Here, the coefficient matrix A _d is expressed by Expression (XXXVI), and the coefficient matrix b _d is expressed by Expression (XXXVII). By solving Expression (XXXVI) and Expression (XXXVII), each element of the coefficient matrix is obtained. Can do.

  The variables used in the above formula are shown below.

The input current i _in and the input current command value i _in ^* (upper stage), the DC link current i _dc and the DC link current command value i _dc ^* (lower stage) by the instantaneous voltage control of the embodiment (embodiment 4) described above in FIG. FIG. 9 shows the input current i _in and the input current command value i _in ^* (upper stage), the DC link current i _dc and the DC link current command value i _dc ^* ( (Lower).

In FIG. 8, a pulse-like error is generated in the DC link current i _dc as indicated by P1 due to the steep fluctuation of the voltage vector. As a result, the resonance current of the resonance circuit due to the parasitic component flows to the input section as indicated by P2, and a high-frequency current oscillation occurs in the power supply waveform. Further, due to the control error of the DC link current i _dc shown in P3, the conduction width of the bridge diodes (diodes D1 to D4) constituting the converter circuit 2 of the input unit is narrowed as shown in P4. These are all due to the difference between the sampling value of the motor current and the average value between samplings.

On the other hand, in FIG. 9, no pulse-like control error occurs in the DC link current i _dc as can be seen from P5. In P6, since the control error of the DC link current i _dc does not occur, the conduction width of the diode bridge of the input section is wide. It can be seen that the input current i _in is a substantially ideal sine wave current by using the predictive control of the DC link current idc.

As described above, according to the predictive control of this embodiment, the current of one sampling destination is predicted, and a predicted isocurrent curve established for the average value between samplings is used. Therefore, the difference between the sampling value and the average value between samplings is calculated. The control error of the DC link current i _dc generated due to this can be greatly reduced. As a result, it is possible to reduce the resonance current of the resonance circuit caused by the parasitic component present in the input unit and the capacitor 9 of the DC link unit 3.

  In addition, the control object of the power converter device 1 of this invention is not limited to IPMSM shown in each Example, The use of the motor 8 is not limited to a compressor.

DESCRIPTION OF SYMBOLS 1 Power converter 2 Converter circuit 3 DC link part 4 Inverter circuit 6 Control part 7 AC power supply 8 Motor 9 Capacitor 11 Speed control part 12 Torque control part 13 Voltage control part 13A alpha beta axis PI control part 13B, 13D Instantaneous voltage control part 13C Sinusoidal tracking control unit 14 Switching control unit 18 Square waveform generation unit 24 dq-αβ conversion unit 26, 27 Subtractor 31, 32 PI calculation unit 33, 34 Multiplier 46, 47 F / B control unit 56 Output voltage correction unit

Claims (12)

A power converter for supplying power to a motor,
A converter circuit for full-wave rectification of the input AC;
A DC link unit having a capacitor connected in parallel to the output of the converter circuit and outputting a DC voltage;
An inverter circuit that switches the output of the DC link unit to convert it into AC and supplies the motor to the motor;
A control unit for controlling the switching,
The control unit is a speed control unit for obtaining an operation amount for controlling the speed of the motor;
A torque control unit for _obtaining an operation amount for controlling the output torque τ _inv of the inverter circuit;
A voltage control unit for obtaining an operation amount for controlling the α-axis current iα and β-axis current iβ of the motor;
The speed control unit pulsates an operation amount for controlling the speed of the motor in synchronization with twice the power frequency ω _s of the input AC,
The torque control unit generates an output torque command value τ _inv ^* of the inverter circuit from an operation amount that controls the speed of the motor, and based on the output torque command value τ _inv ^* , the output torque τ of the inverter circuit _{Find the} amount of operation to control _inv ,
The voltage control unit generates an α-axis current command value iα ^* and a β-axis current command value iβ ^* of the motor on a stationary coordinate system based on an operation amount for controlling the output torque τ _inv of the inverter circuit,
The α-axis voltage command value vα ^* and β-axis voltage command value of the motor so that the deviation between the α-axis current command value iα ^* and β-axis current command value iβ ^* and the α-axis current iα and β-axis current iβ is small. Generate vβ ^*
In addition, a DC link current command value i _dc ^* flowing in the inverter circuit is generated based on an operation amount for controlling the speed of the motor, and the DC link current i _dc becomes equal to the DC link current command value i _dc ^*. Thus, the α-axis voltage command value vα ^* and the β-axis voltage command value vβ ^* are corrected as described above. The operation amount for controlling the speed of the motor obtained by the speed control unit is obtained by dividing the input power command value of the inverter circuit by the rotation angle frequency ω _rm of the mechanical angle, and converting it to a torque dimension, τ _in ^* The power converter according to claim 1, wherein the voltage control unit generates the DC link current command value i _dc ^* based on the input torque command value τ _in ^* . The operation amount for controlling the output torque τ _inv of the inverter circuit determined by the torque control unit is the q-axis current command value i _q ^* of the motor, and the voltage control unit is configured to output the q-axis current command value i _q ^*. The power conversion device according to claim 1, wherein the α-axis current command value iα ^* and the β-axis current command value iβ ^* are generated based on The voltage control unit is configured to reduce the deviation between the α-axis current command value iα ^* and β-axis current command value iβ ^* and the α-axis current iα and β-axis current iβ by PI control ^. and it generates the β-axis voltage command value v? ^*, further the DC link current i _dc is the DC link current command value i _dc ^* and equal manner the α-axis voltage value v? ^* and the β-axis voltage value v? The power conversion device according to claim 1, wherein ^* is corrected. The voltage control unit performs feedback control with an F / B control unit including a control gain having a sine wave component of a motor current frequency ω _re of the motor, thereby obtaining the α-axis current iα and the β-axis current iβ. The power converter according to claim 4, wherein the power converter is made to follow the current frequency ω _re . The voltage control unit, the correction coefficient A _IVC calculated from the ratio of the DC link current command value i _dc ^* and the DC link current i _dc, the DC link current i _dc and the DC link current command value i _dc ^* The α-axis voltage command value vα ^* and the β-axis voltage command value vβ ^* so as to match,
When the corrected α-axis voltage command value vα ^* and β-axis voltage command value vβ ^* exceed the outputable voltage range, the DC link current i _dc and the DC link current command are not related to the correction coefficient A _IVC. The α-axis voltage command value vα ^* and the β-axis voltage command value vβ ^{* at} which the value i _dc ^* coincides and the output voltage is maximum are calculated and corrected by replacing them. Item 6. The power conversion device according to any one of Items 5 above. When the amount of change in the corrected α-axis voltage command value vα ^* and β-axis voltage command value vβ ^* exceeds a specified value R _lim , the voltage control unit does not exceed the specified value R _lim , and 7. The power conversion according to claim 6, wherein the α-axis voltage command value vα ^* and the β-axis voltage command value vβ ^{* at} which the DC link current i _dc and the DC link current command value i _dc ^* coincide are calculated. apparatus. The voltage control unit predicts the value of the DC link current i _dc at the time of the next sampling, and the α axis where the DC link current i _dc obtained based on the predicted value matches the DC link current command value i _dc ^* The voltage command value vα ^* and the β-axis voltage command value vβ ^* are calculated and replaced to correct the α-axis voltage command value vα ^* and the β-axis voltage command value vβ ^*. 5. The power conversion device according to claim 5. The voltage control unit, according to claim, characterized in that determining the DC link current i _dc from the average value of the values of the DC link current i _dc at the value and the next DC link current i _dc at the next sampling the sampling 8. The power conversion device according to 8. The voltage control unit predicts the values of the α-axis current iα and β-axis current iβ of the motor at the next sampling, or at the next sampling and the next sampling, and at the next sampling based on the predicted values the value of the DC link current i _dc, or, the next sampling value of the DC link current i _dc at the and calculates the value of the DC link current i _dc at the time of next sampling claim 8 or claim Item 10. The power conversion device according to Item 9. The voltage control unit calculates the α-axis voltage command value vα calculated so that a deviation between the α-axis current command value iα ^* and β-axis current command value iβ ^* and the α-axis current iα and β-axis current iβ is small. ^The correction is performed so that ^* and the β-axis voltage command value vβ ^* and the corrected α-axis voltage command value vα ^* and the β-axis voltage command value vβ ^* are in phase. The power converter device in any one of Claims 10. The torque control unit is configured to calculate the inverter circuit from the α-axis current iα, the β-axis current iβ, the α-axis voltage command value vα ^* , the β-axis voltage command value vβ ^*, and the rotational angular frequency ω _{rm of the} mechanical angle. 12. The output torque τ _inv is calculated, and feedback control is performed so that the output torque τ _inv and the output torque command value τ _inv ^* coincide with each other. The power converter described.

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