JP2017154548A - Motor controller and electric power steering device mounted therewith - Google Patents

Abstract

PROBLEM TO BE SOLVED: To provide a motor controller that compensates for a counter electromotive voltage and characteristics of a counter electromotive voltage compensation path through a feedback path without any variation in sensitivity to noise nor any delay over a wide range so as to securely let a real current follow up a current command value, and to provide an electric power steering device which improves a steering feeling.SOLUTION: A motor controller which performs driving control over a motor through current feedback control based upon a current command value and also performs compensation for a motor counter electromotive voltage with a counter electromotive voltage compensation signal based upon a motor angle or motor angular velocity is characterized in that a path for compensating for a motor counter electromotive voltage is provided with an LPF for noise elimination, and a feedback path of motor control is provided with an angle feedback filter which cancels a counter electromotive voltage compensation difference as a difference between the motor counter electromotive voltage and counter electromotive voltage compensation signal.SELECTED DRAWING: Figure 8

Description

  The present invention includes a motor control device that drives a motor with current feedback based on a current command value and controls the current command value by following the actual motor current, and a current command calculated based on at least steering torque The present invention relates to an electric power steering device that applies an assist force by a motor to a steering system of a vehicle according to a value. In particular, a motor control device in which a back electromotive force voltage is canceled when the motor is driven without changing sensitivity to noise, and a compensation error canceling angle filter that performs noise compensation and LPF compensation is inserted in a feedback path, and the same The present invention relates to an electric power steering device equipped with

  An electric power steering device (EPS) equipped with a motor control device applies steering assist force (assist force) to the steering mechanism of the vehicle by the rotational force of the motor, and the driving force of the motor controlled by the inverter is A steering assist force is applied to the steering shaft or the rack shaft by a transmission mechanism such as a gear. Such a conventional electric power steering apparatus performs feedback control of the motor current in order to accurately generate the torque of the steering assist force. In feedback control, the motor applied voltage is adjusted so that the difference between the steering assist command value (current command value) and the motor current detection value is small. The adjustment of the motor applied voltage is generally performed by PWM (pulse width). This is done by adjusting the duty of modulation) control.

  The general configuration of the electric power steering apparatus will be described with reference to FIG. 6b is further connected to the steering wheels 8L and 8R via hub units 7a and 7b. Further, the column shaft 2 is provided with a torque sensor 10 for detecting the steering torque Th of the handle 1 and a steering angle sensor 14 for detecting the steering angle θ, and a motor 20 for assisting the steering force of the handle 1 is a reduction gear. 3 is connected to the column shaft 2 through 3. The control unit (ECU) 30 that controls the electric power steering apparatus is supplied with electric power from the battery 13 and also receives an ignition key signal via the ignition key 11. The control unit 30 calculates a current command value of an assist (steering assist) command based on the steering torque Th detected by the torque sensor 10 and the vehicle speed Vs detected by the vehicle speed sensor 12, and compensates the current command value. The current supplied to the EPS motor 20 is controlled by the voltage control command value Vref subjected to.

  Note that the steering angle θ is detected from the steering angle sensor 14, and the steering angle can be acquired from a rotation sensor such as a resolver connected to the motor 20.

  The control unit 30 is connected to a CAN (Controller Area Network) 40 that transmits and receives various types of vehicle information, and the vehicle speed Vs can also be received from the CAN 40. The control unit 30 can be connected to a non-CAN 41 that exchanges communications, analog / digital signals, radio waves, and the like other than the CAN 40.

  The control unit 30 is mainly composed of a CPU (including an MPU, MCU, etc.). FIG. 2 shows general functions executed by a program inside the CPU.

  Referring to FIG. 2, the control unit 30 will be described. The steering torque Th detected by the torque sensor 10 and the vehicle speed Vs detected by the vehicle speed sensor 12 (or from the CAN 40) are a current command for calculating a current command value Iref1. The value is input to the value calculation unit 31. The current command value calculation unit 31 calculates a current command value Iref1, which is a control target value of the current supplied to the motor 20, using an assist map or the like based on the input steering torque Th and vehicle speed Vs. The current command value Iref1 is input to the current limiting unit 33 via the adding unit 32A, and the current command value Irefm with the maximum current limited is input to the subtracting unit 32B, and the deviation ΔI (= Irefm−Im) is calculated, and the deviation ΔI is input to the PI control unit 34 for improving the characteristics of the steering operation. The voltage control command value Vref whose characteristics are improved by the PI control unit 34 is input to the PWM control unit 35, and the motor 20 is further PWM driven via the inverter 36. The current value Im of the motor 20 is detected by the motor current detector 37 and fed back to the subtraction unit 32B. The inverter 36 is composed of an FET bridge circuit as a semiconductor switching element.

  A rotation sensor 21 such as a resolver is connected to the motor 20, a motor rotation angle θ is output from the rotation sensor 21, and a motor speed ω is calculated by a motor speed calculation unit 22.

  Further, the compensation signal CM from the compensation signal generation unit 38 is added to the addition unit 32A, and the compensation of the steering system system is performed by adding the compensation signal CM to improve the convergence property, the inertia property, and the like. ing. The compensation signal generator 38 adds the self-aligning torque (SAT) 38-1 and the inertia 38-2 by the adder 38-4, and further adds the convergence 38-3 to the addition result by the adder 38-5. The addition result of the adder 38-5 is used as the compensation signal CM.

  In such a motor control device of an electric power steering device, since the motor 20 generates a counter electromotive voltage when the motor is driven, compensation for suppressing or attenuating the motor counter electromotive voltage is required. The reason will be described below.

A control system in which the motor 20 is driven from the current command value Iref is represented by a transfer function as shown in FIG. The current command value Iref is input to the subtraction unit 104 via the control filter (G _FF ) 101, and a deviation e _{1 from the} actual motor current Imn is calculated. The deviation e ₁ is input to the subtraction unit 105 through the control filter (G _FB ), the back electromotive voltage EMF of the motor 20 is subtracted by the subtraction unit 105, and the difference e ₃ is obtained as the electric system characteristic unit 110 (1 / (L · s + R)), and further, the torque constant Kt [Nm / A] is inputted to the mechanical system characteristic section 120 (1 / (J · s + D)). The counter electromotive voltage EMF is obtained by multiplying the motor angular velocity (motor rotational speed) ωm, which is an output of the mechanical characteristic unit 120, by a counter electromotive voltage constant Ke [V / (rad / s)]. The motor current Im from the electric system characteristic unit 110 is detected and fed back, but actually, the current detection noise Ni is mixed and fed back as the actual motor current Imn.

L in the electric system characteristic unit 110 is motor correlation inductance [H], R is motor correlation resistance [Ω], J in the mechanical system characteristic unit 120 is motor moment of inertia [Kg · m ² ], and D is motor viscosity coefficient [ Nm / (rad / s)].

In order to set the system from the current command value Iref to the motor current Im as a primary filter (1 / (T ₄ · s + 1)) whose frequency band is easy to control as shown in FIG. 4, T _{1 to} T ₄ are time constants. The transfer function of the control filter (G _FF ) 101 is set by the following formula 1, and the transfer function of the control filter (G _FB ) 102 is set by the following formula 2.

即ち、電流指令値Ｉｒｅｆからモータ電流Ｉｍまでの系を“１／（Ｔ_４・ｓ＋１）”（数３）とし、時定数Ｔ_４の設定で電流指令値Ｉｒｅｆからモータ電流Ｉｍまでの特性である電流制御帯域を決定（１次フィルタ）するために、制御フィルタ（Ｇ_ＦＦ）１０１及び制御フィルタ（Ｇ_ＦＢ）１０２の特性を決定する。

That is, the system from the current command value Iref to the motor current Im is “1 / (T ₄ · s + 1)” (Equation 3), and the characteristic from the current command value Iref to the motor current Im is set by setting the time constant T _4. In order to determine the current control band (primary filter), the characteristics of the control filter (G _FF ) 101 and the control filter (G _FB ) 102 are determined.

そして、制御フィルタ（Ｇ_ＦＢ）１０２を含む電流制御のフィードバック閉ループの系では、逆起電圧ＥＭＦを０として、数４及び図５に示すように、外乱の電流検出ノイズ（Ｎｉ）除去用の２次フィルタ（時定数Ｔ_２，Ｔ_３）と１次位相進みフィルタ（Ｔ_１）となる構成を採る。１次位相進みフィルタ（Ｔ_１）の項がないと、制御フィルタ（Ｇ_ＦＦ）１０１の分子の微分項が２次になってしまい、非常に不安定になるからである。要するに、制御フィルタ（Ｇ_ＦＦ）１０１は、電流指令値Ｉｒｅｆからモータ電流Ｉｍまでの特性を決めるフィルタであり、制御フィルタ（Ｇ_ＦＢ）１０２は、電流検出ノイズＮｉを除去するフィルタを含めた、電流指令値Ｉｒｅｆからモータ電流Ｉｍまでの特性を決めるフィルタと言える。

In the current control feedback closed loop system including the control filter (G _FB ) 102, the counter electromotive voltage EMF is set to 0, and 2 for removing the current detection noise (Ni) of disturbance as shown in Equation 4 and FIG. 5. A configuration is adopted in which a _second order filter (time constants T ₂ and T ₃ ) and a first order phase advance filter (T ₁ ) are used. This is because if there is no term of the primary phase advance filter (T ₁ ), the differential term of the numerator of the control filter (G _FF ) 101 becomes second order and becomes very unstable. In short, the control filter (G _FF ) 101 is a filter that determines the characteristics from the current command value Iref to the motor current Im, and the control filter (G _FB ) 102 is a current including a filter that removes the current detection noise Ni. It can be said that the filter determines the characteristics from the command value Iref to the motor current Im.

上記数１及び数４をＧ_ＦＢについて解くと、制御フィルタ（Ｇ_ＦＢ）１０２は上記数２となる。

When Equation 1 and Equation 4 are solved for G _FB , the control filter (G _FB ) 102 becomes Equation 2 above.

Thus, the setting of the control filter (G _FF ) 101 and the control filter (G _FB ) 102 does not take into account the mixing of the motor back electromotive voltage EMF. However, the generation of the motor back electromotive voltage EMF affects the current command value, which is a serious problem in accurately controlling the motor output. As such a countermeasure, compensation of the motor back electromotive force voltage is conceivable. For example, in Japanese Patent Laid-Open No. 2013-21870 (Patent Document 1), the back electromotive force voltage of the motor is estimated based on the voltage command value and the current detection value, The estimated back electromotive force value is added to the current command value. In Japanese Patent Laid-Open No. 2012-236472 (Patent Document 2), a counter electromotive voltage is estimated based on the rotational angular velocity, and the counter electromotive voltage compensation control value is calculated by multiplying the estimated counter electromotive voltage by a compensation coefficient. A voltage command value is obtained by adding the electromotive force compensation control value to the basic voltage.

JP 2013-2119880 A JP 2012-236472 A

  In all of the back electromotive force compensations described above, compensation is made on the assumption that position and rotation speed detection noise is not mixed. However, since detection noise is actually mixed, the current command value is accurately detected. There is a problem that the motor output cannot be made to follow.

  The present invention has been made under the circumstances as described above. The object of the present invention is to change the characteristics of the counter electromotive voltage and the counter electromotive voltage compensation path with a feedback path without changing the sensitivity to noise over a wide range. An object of the present invention is to provide a motor control device that compensates and ensures that an actual current follows a current command value, and an electric power steering device with improved steering feeling.

  The present invention relates to a motor control device that controls driving of a motor by current feedback control based on a current command value and compensates for a motor counter electromotive voltage by a counter electromotive voltage compensation signal based on a motor angle or a motor angular velocity. The object is to provide an LPF for noise removal in a path for compensating the motor back electromotive voltage, and in the motor control feedback path, a back electromotive force that is a difference between the motor back electromotive voltage and the back electromotive voltage compensation signal. This is achieved by providing an angle feedback filter that cancels the voltage compensation error.

  The above object of the present invention is that the control of the motor is a dq axis vector control, or the current command value is a dq axis current command value and the control unit is a two-phase feedback vector control system. Alternatively, the current command value is a dq-axis current command value, and the control unit is a three-phase feedback vector control system. By mounting, an electric power steering device with good steering feeling can be provided.

  According to the motor control device of the present invention, in the motor control device having a function of compensating for the motor back electromotive voltage when the motor is driven, a back electromotive force that is a difference between the motor back electromotive voltage and the back electromotive force compensation signal is provided in the feedback path. Since the angle feedback filter that cancels the voltage compensation error is inserted, the follow-up characteristic of the motor can be improved without delay without changing the sensitivity to noise.

It is a lineblock diagram showing an outline of an electric power steering device. It is a block diagram which shows the structural example of the control unit (ECU) of an electric power steering apparatus. It is a block diagram which shows the path | route of generation | occurrence | production of the back electromotive force voltage in a motor control apparatus (transfer function). It is a frequency response figure for demonstrating the characteristic of a control filter. It is a frequency response figure for demonstrating the characteristic of a control filter. It is a block diagram which shows the structural example of the vector control system (3 phase FB) which can apply this invention. It is a block diagram which shows the structural example of the vector control system (2 phase FB) which can apply this invention. It is a block diagram which shows the structural example of the back electromotive force compensation in a motor control apparatus (transfer function). FIG. 6 is a frequency characteristic diagram showing the gain / phase frequency characteristics and the effect of the present invention when LPF is used for back electromotive force compensation. It is a block block diagram which shows an example of embodiment of this invention.

The motor control device of the present invention compensates for the generation of the back electromotive voltage when the motor is driven without changing the sensitivity to noise, and uses an angle filter for compensating error compensation that performs noise compensation and LPF compensation as a feedback path. It is inserted. As a result, the actual motor current can be made to accurately follow the current command value without delay without changing the sensitivity to noise.The present invention can also be applied to a vector control system for driving and controlling a brushless motor. The control system will be described.

The vector control system of FIG. 6, and the current command value calculating section 220 is provided to correct by calculating the d-axis current command value i _d and the q-axis current command value i _q, steering the current command value calculating section 220 The torque Th, the vehicle speed Vs, the motor angle (rotation angle) θ _e , and the motor angular velocity ω calculated by the angular velocity calculation unit 226 are input from the rotation sensor 100 A connected to the motor 100. D-axis current command value i _d and the q-axis current command value i _q calculated by the current command value calculating unit 220 is input to the 2-phase / 3-phase conversion unit 221, three-phase current in synchronization with the motor angle theta _e The command values are converted into Iuref, Ivref, and Iwref. Three-phase current command values Iuref, Ivref, Iwref are input to a subtractor 222 (222u, 222v, 222w), and deviations ΔIu, ΔIv, ΔIw from motor currents Imu, Imv, Imw detected by the current detection circuit 225A are obtained. Calculated. The calculated deviations ΔIu, ΔIv, ΔIw are input to the PI control unit 223, and the current-controlled three-phase voltage control command values Vuref, Vvref, Vwref are input to the PWM control unit 224, and are calculated by the PWM control unit 224. The motor 100 is driven via the inverter 225 based on each phase duty.

  In FIG. 6, the current detection circuit 225 </ b> A is provided in the inverter 225, but it can also be detected by a supply line to the motor 100.

Also, in the vector control system of FIG. 7, the current detection circuit 225A with the detected three-phase motor current Imu, Imv, 3-phase converted to 2-phase in synchronism with the motor angle theta _e the Imw / 2 phase converter 227 Is provided. D-axis current command value computed is corrected by the current command value calculating section 220 _{i d} and the q-axis current command value _{i q} is input to the subtraction unit 222 (222d, 222q), 3-phase / 2-phase conversion by the subtraction unit 222 Deviations Δi _d and Δi _q from the two-phase currents Imd and Imq from the unit 227 are calculated. The deviations Δi _d and Δi _q are input to the two-phase / three-phase converter 221, and the converted three-phase current command values Iuref, Ivref, Iwref are input to the PI controller 223, and thereafter the same as in FIG. Actions are performed.

  The control system in FIG. 6 is a three-phase feedback vector control system that feeds back three-phase motor currents Imu, Imv, and Imw, and the control system in FIG. 7 has three-phase motor currents Imu, Imv, and Imw of 2. This is a two-phase feedback vector control system which is converted into phase currents Imd and Imq and fed back. The present invention can be applied to both the three-phase feedback vector control system and the two-phase feedback vector control system.

  FIG. 8 shows a configuration example in the case where the back electromotive force EMF is compensated based on the motor angle θm detected by a rotation sensor (for example, a resolver), corresponding to FIG. Although the motor angle θm is differentiated by the differentiating unit 130, the rotation sensor (for example, the resolver) actually includes noise Nr, and the motor angle θmr to which the noise Nr is added (mixed) by the adding unit 133 is differentiated. Input to the unit 130. The motor angular velocity ωn differentiated by the differentiating unit 130 is input to a low-pass filter (LPF) 131 for noise removal whose transfer function is expressed by Equation 5, and the motor angular velocity ωn ′ from which the noise Nr has been removed by the LPF 131 is the inverse. The back electromotive force compensation constant Ke ′ is multiplied by the back electromotive force compensation constant unit 132 and input to the adder 106 as a back electromotive voltage compensation signal EMFc.

Incidentally, omega _L of LPF131 have a relation of ω _L = 2πf _c relative to the cutoff frequency _{f c} (e.g. 40 Hz).

この逆起電圧補償はモータ角度θｍ（モータ角速度ωｍ）に依存しており、ノイズＮｒの混入がなければＬＰＦ１３１は不要である。しかし、実際には回転センサから混入するノイズＮｒの影響があり、その影響を除去若しくは低減するＬＰＦ１３１を具備している。フィルタ処理を行わないと、電動パワーステアリング装置では操舵感が悪化してしまうからである。

This back electromotive force compensation depends on the motor angle θm (motor angular velocity ωm), and the LPF 131 is not required unless noise Nr is mixed. However, there is actually an influence of noise Nr mixed from the rotation sensor, and the LPF 131 for removing or reducing the influence is provided. This is because if the filter process is not performed, the steering feeling is deteriorated in the electric power steering apparatus.

  If the frequency of the motor rotation speed is increased by the filtering process of the LPF 131, the phase is delayed with respect to the motor counter electromotive voltage EMF, and the motor counter electromotive voltage EMF may not be completely canceled. This cancellation error is mixed into the current control system as a disturbance, and the actual motor current Im, which is the role of feedback control, cannot follow the current command value Iref. If the back electromotive force compensation is complete, the relationship of the following formula 6 should be satisfied.

(Equation 6)
EMFc-EMF = 0
However, the above equation 6 does not hold due to the mixing of noise Nr. In the first place, the main role of the current feedback control is to flow an actual current without delay according to the current command value. In the control by the filter process, the followability can be improved as the control band is higher, but the sensitivity to noise is increased, and the steering feeling is deteriorated in the electric power steering apparatus.

  FIG. 9 shows the frequency characteristics when the back electromotive force compensation is performed using the LPF. FIG. 9A shows the frequency characteristics of the gain. Before the countermeasure according to the present invention, the gain is obtained in the frequency range of 10 to 100 Hz. Is lowered in a concave shape. Further, FIG. 9B shows the frequency characteristics of the phase, and before the countermeasure according to the present invention, there is a phase delay and advance, not the flat characteristics, and the back electromotive force EMF cannot be completely compensated. The compensation cancellation error enters the control loop as a disturbance, and this causes the detected current to not follow the current command value.

As a countermeasure, in the present invention, an angle feedback (FB) filter (G _ANG ) 200 having a current tracking characteristic of Im / Iref = 1 / (T ₄ · s + 1) is inserted in a feedback path for motor control. . FIG. 10 shows the configuration corresponding to FIG.

When the current command value Iref to the motor current Im output from the electrical system characteristic unit 110 are expressed by a transfer function, the following Expression 7 is obtained. The transfer function of the control filter (G _FF ) 101 is Equation 1 described above, and the transfer function of the control filter (G _FB ) 102 is Equation 2 described above.

そして、理想的には数７が上記数３の特性のように、下記数８で表わされれば良い。

Ideally, Equation 7 may be expressed by Equation 8 below as shown in Equation 3 above.

そのためには、数７の分母の（）内が０になれば良い。即ち、下記数９が成立すれば良い。

For that purpose, it is sufficient that the value in parentheses in the denominator of Equation 7 is zero. That is, it is sufficient that the following formula 9 is established.

換言すれば、数７の分母の（）内を０とする数９が成立するような、角度フィードバックフィルタ（Ｇ_ＡＮＧ）２００をフィードバック経路に介挿すれば良い。

In other words, an angle feedback filter (G _ANG ) 200 may be inserted in the feedback path so that _Formula 9 in which the value in () of the denominator of Formula 7 is 0 is established.

Therefore, when the angle feedback filter G _ANG is calculated from _Equation 9, _Equation 10 is obtained.

数１０で決定された角度フィードバックフィルタ（Ｇ_ＡＮＧ）を電流指令値Ｉｒｅｆに適用すると、電流値Ｉｍは下記数１１で表わされ、数１１にＧ_ＦＦの伝達関数（数１）及びＧ_ＦＢの伝達関数（数２）を代入すると数１２となり、数１２の時定数Ｔ_４の調整により所望の特性を得ることができる。

When the angle feedback filter determined by the number 10 _{(G ANG)} to apply to the current command value Iref, the current value Im is represented by the following Expression 11, the number 11 in the transmission of _{G FF} function (Equation 1) and _{G FB} the number 12 becomes substituting the transfer function (equation 2), it is possible to obtain desired characteristics by adjusting the constant T ₄ when the number 12.

図９は、操舵時に、検出電流が電流指令値に追従しない問題におけるメカニズムをボード線図で比較しており、対策前では逆起電圧補償ロジックにはＬＰＦ（４０Ｈｚ）があり、逆起電圧を完全に補償できない。この相殺誤差が外乱として制御ループに混入し、これが要因となって検出電流が電流指令値に追従できない。しかし、上述したモータ逆起電圧と逆起電圧補償信号との差である逆起電圧補償誤差を相殺する角度フィードバックフィルタ２００（Ｇ_ＡＮＧ）を設けることにより、対策前の特性が改善されていることが分かる。即ち、ゲイン特性も位相特性も平坦な特性となっている。

FIG. 9 compares the mechanism of the problem that the detected current does not follow the current command value during steering with a Bode diagram. Before the countermeasure, the back electromotive voltage compensation logic has LPF (40 Hz), and the back electromotive voltage is Cannot fully compensate. This cancellation error enters the control loop as a disturbance, and this causes the detected current to not follow the current command value. However, by providing the angle feedback filter 200 (G _ANG ) that cancels the back electromotive force compensation error, which is the difference between the motor back electromotive voltage and the back electromotive force compensation signal, the characteristics before countermeasures are improved. I understand. That is, the gain characteristic and the phase characteristic are flat.

1 Handle 2 Column shaft (steering shaft, handle shaft)
10 Torque sensor 12 Vehicle speed sensor 20, 100 Motor 30 Control unit (ECU)
31, 220 Current command value calculation unit 34, 223 PI control unit 35, 224 PWM control unit 36, 225 Inverter 101, 102 Control filter 110 Electrical system characteristic unit 120 Mechanical system characteristic unit 131 Low pass filter (LPF)
200 Angle Feedback (FB) Filter 221 2 Phase / 3 Phase Converter 227 3 Phase / 2 Phase Converter

Claims (5)

In the motor control device that performs drive control of the motor by current feedback control based on the current command value and compensates for the motor back electromotive force by the back electromotive force compensation signal based on the motor angle or the motor angular speed,
An LPF for noise removal is provided in a path for compensating the motor back electromotive voltage, and a back electromotive voltage compensation error which is a difference between the motor back electromotive voltage and the back electromotive voltage compensation signal is provided in the feedback path of the motor control. An angle feedback filter that cancels out the noise is provided. The motor control device according to claim 1, wherein the control of the motor is dq axis vector control. The motor control device according to claim 2, wherein the current command value is a dq-axis current command value, and the control unit is a two-phase feedback vector control system. The motor control device according to claim 2, wherein the current command value is a dq-axis current command value, and the control unit is a three-phase feedback vector control system. An electric power steering device equipped with the motor control device according to claim 1.

Images