JP2016134970A - Controller for induction motor - Google Patents

Abstract

PROBLEM TO BE SOLVED: To solve the problem that, when current is caused to flow by PWM in a controller that performs vector control of an induction motor, current controllability may be deteriorated as a rotational frequency ω becomes higher, if ω is significantly corrected by secondary resistance identification correction value calculation means, current controllability may be deteriorated and especially, the current controllability may become worst at the time of deceleration from high-speed rotation.SOLUTION: The controller has two kinds of gains for identifying secondary resistance and as a gain at the time of deceleration, a smaller setting value is used, thereby preventing the deterioration of the current controllability.SELECTED DRAWING: Figure 1

Description

  The present invention relates to an induction motor control device that is used for driving a spindle of a machine tool and arbitrarily controls an output torque of an induction motor.

  For applications such as spindle driving of machine tools, induction motors driven by slip frequency vector control are often used. In this slip frequency vector control, in order to arbitrarily control the output torque, it is necessary to give an accurate slip frequency to the motor in accordance with the secondary resistance R2, excitation inductance M, and torque current i1q of the motor. In general, the secondary resistance R2 fluctuates approximately twice as much as the temperature of the motor changes. As a result, the slip frequency applied to the motor becomes inaccurate, and the output torque cannot be controlled accurately. In order to solve this problem, the present applicants already disclosed in Japanese Patent Laid-Open No. 10-52100, Japanese Patent No. 3363732, etc., even when the value of the secondary resistance R2 of the induction motor and other parameters fluctuate. We have proposed a control device for an induction motor that can control torque accurately.

  FIG. 5 shows an example of a system configuration of a control device for an induction motor according to the prior art. A torque command T * and a magnetic flux density command φ * are input to the control device as input commands from the outside. The converter 1 is an exciting current command generator that generates a necessary exciting current command value id * according to the magnetic flux density command φ *, and the relationship between the magnetic flux density and the exciting current means an exciting inductance M as described later. ing. In this converter 1, the exciting current command value id * is output by multiplying the reciprocal of the exciting inductance M.

  The divider 2 divides the input torque command T * by the input magnetic flux density command φ *, and the output torque of the induction motor is proportional to the product of the magnetic flux density and the torque current value. The output of the device 2 is output as a torque current command value iq *.

The operation of this induction motor control device will be described based on the equivalent circuit of the induction motor shown in FIG. The primary current I1, the excitation current Io, and the primary voltage E1 of the motor are expressed as follows using the currents id, iq, iod, ioq, and voltages e1d, e1q on the dq axis coordinates that rotate in synchronization with the rotation frequency ω of the magnetic flux. expressed.
I1 = id · sin ωt + iq · cos ωt (1)
Io = iod · sin ωt + ioq · cos ωt (2)
E1 = e1d · sin ωt + e1q · cos ωt (3)

At this time, the voltage equation for the primary circuit of the induction motor is expressed as follows.
e1d = (R1 + pLσ) id−ωLσ · iq + pM · iod−ωM · ioq (4)
e1q = ωLσ · id + (R1 + pLσ) iq + ωM · iod + pM · ioq (5)

  Here, p is a differential operator d / dt, R1 is a primary winding resistance, Lσ is a leakage inductance, and M is an excitation inductance.

Next, the voltage equation for the secondary circuit is similarly expressed as follows.
−R2 · id + (R2 + pM) iod−ωs · M · ioq = 0 (6)
-R2 · iq + ωs · M · iod + (R2 + pM) ioq = 0 (7)

Here, R2 is the secondary winding resistance, and ωs is the slip frequency. This ωs is expressed as follows using the rotational angular frequency ωm of the motor.
ωs = ω−ωm (8)

Assuming that the magnetic flux direction coincides with the d-axis, the excitation current Io inside the motor is expressed as follows.
Io = φ / M = iod, ioq = 0 (9)

When the relationship between Io and id is obtained from the equations (9) and (6), the following equation is obtained.
Io / id = 1 / (1 + pM / R2) (10)

  That is, the exciting current Io responds to id with a first-order lag, and its time constant is M / R2. This time constant is several hundreds of milliseconds in a general induction motor, and it can be approximated that the change of Io is sufficiently slow.

On the other hand, the following equation is obtained from the equations (9) and (7).
ωs = R2 · iq / (M · Io) (11)

This is a so-called vector control condition. When ωs satisfying this equation is given to the motor, the magnetic flux direction coincides with the d-axis. At this time, since iq is orthogonal to the magnetic flux, the generated torque T of the motor is as follows.
T = φ · iq = M · Io · iq (12)
Therefore, the torque can be arbitrarily controlled by controlling iq.

When it is approximated that the change of Io is sufficiently slow as described above, the equations (4) and (5) can be rewritten as follows.
e1d = (R1 + pLσ) id−ωLσ · iq (13)
e1q = ωLσ · id + (R1 + pLσ) iq + ωM · Io (14)

From these equations, when id and iq are to be controlled arbitrarily, the voltage applied to the motor may be controlled as follows.
e1d * = Gd · Δid−ωLσ · iq * (15)
e1q * = ωLσ · id * + Gq · Δiq + ωM · Io * (16)

Here, the subscript * means a command value, and Δid and Δiq are current errors expressed by the following equations.
Δid = id * -id, Δiq = iq * -iq (17)

  Gd and Gq are sufficiently large gains, and are realized using a pi operational amplifier or the like. The equations (15) and (16) are calculated by the dq-axis voltage command calculation unit 4 in FIG. 5, and the internal block diagram is shown in FIG. Further, ωs is output by the divider 7 and the converter 8 so as to satisfy the expression (11).

When the output torque T expressed by the equation (12) is to be accurately controlled, the vector control condition of the equation (11) needs to be satisfied and the magnetic flux position needs to coincide with the d-axis. However, as described above, since the secondary resistance R2 in the equation (11) varies, as a result, the output torque accuracy deteriorates. In Japanese Patent Laid-Open No. 10-52100, the secondary resistance R2 is identified based on the following equation.
Gq · Δiq = ΔR2 (ω / ωs) iq (18)

Here, ΔR2 is an error between the value R2c assumed on the controller side and the actual value R2 inside the motor. This equation (18) is derived as follows. First, equation (11) is transformed to obtain equation (19).
ωM · Io = (ω / ωs) R2 · iq (19)
By substituting the equation (19) into the equation (14), the q-axis voltage actually generated in the motor can be expressed as follows.
e1q = ωLσ · id + (R1 + pLσ) iq + (ω / ωs) R2 · iq (20)

On the other hand, the voltage e1q * output from the controller can be expressed as follows from the equations (19) and (16).
e1q * = Gq · Δiq + ωLσ · id * + R1 · iq * + (ω / ωs) R2c · iq * (21)
When the difference between the expressions (20) and (21) is obtained with id * = id, iq * = iq, and e1q = e1q * by the action of the current control system, the following expression is obtained.
e1q−e1q * = pLσ · iq−Gq · Δiq + (ω / ωs) · (R2−R2c) · iq = 0 (22)
Since the first term is relatively smaller than the other terms, ignoring it yields equation (18).

In other words, since the output Gq · Δiq of the q-axis current error amplifier represents the setting error ΔR2 of the secondary resistance R2 from the equation (18), the set value of the secondary resistance can be corrected using this Gq · Δiq. Is possible. In FIG. 5, Gq · Δiq is described as Δeq, and the amplifier 25 multiplies Δeq by the identification gain Gr to compensate the secondary resistance R2 set as a coefficient of the converter 8 according to the output. doing. The identification gain Gr is configured by using a general pi operational amplifier or the like, and outputs a correction value of the secondary resistance by a calculation as shown in the following equation, where the proportional component gain is Grp and the integral component gain is Gri.
ΔR2 = Grp · Gq · Δiq + Gri · ∫ (Gq · Δiq) dt (23)

  As described above, in the invention of JP-A-10-52100, the true value in the actual motor is identified for the parameter used for control: the secondary resistance R2, and the control parameter is automatically compensated appropriately. The desired output torque can be obtained with high accuracy without being affected by the fluctuation of the secondary resistance R2 due to the temperature change of the motor.

Here, as described in Japanese Patent No. 3363732, when the angular frequency ω becomes equal to or lower than the angular frequency ωb at the base speed of the motor, the sensitivity of secondary resistance identification decreases, and the output torque in a region where the torque command is small or in a low rotation region. Accuracy deteriorates. In order to improve this, the secondary resistance identifier 31 in FIG. 6 receives the torque current command and the torque current error amplifier output Δeq, that is, (Gq · Δiq) in the dq axis voltage command calculation unit 4, or the torque Current command and torque current error amplifier output (Gq · Δiq) and angular frequency command are input, or excitation current command, torque current command, torque current error amplifier output (Gq · Δiq) and angular frequency are input. A secondary resistance compensation value is calculated and added to a secondary resistance value assumed in advance in the controller to output a secondary resistance identification value.

  The internal configuration of the secondary resistance identifier 31 is shown in FIG. Δeq in the figure is an error amplifier output of torque current, and Δeq = Gq · Δiq. A is the torque current command iq * or iq * / id * calculated from the torque current command iq * and the excitation current command i1 *. Here, A = iq * / id * is used for explanation. r2c is a secondary resistance initial value preset in the controller. Aref in FIG. 7 is, for example, a value iq * / id * = 2 in which the value of K does not rapidly decrease in FIG. Since the relationship between the angular frequency ω and ωφ * is as shown in FIG. 10, for example, ωref = ωb (ωb is the angular frequency at the base speed of the motor). The comparator 32 outputs High when ω> ωref, that is, when ω> ωb. The comparator 33 outputs High when A> Aref, that is, iq * / id *> 2. When both outputs of the comparators 32 and 33 are High, the AND unit 34 outputs High. When the output of the AND unit 34 is High, the switch 38 in the identification value switching unit 35 is closed, and the secondary resistance compensation value calculated by the calculator 25 using the current error amplifier output Δeq is output. When the output of the AND unit 34 is Low, the switch 38 in the identification value switch 35 is opened, so that the value when the switch is closed is held and output. In the holder 36 in FIG. 7, when the above condition has not been met, that is, when the output of the AND unit 34 remains Low, the output remains High, and when the above condition is met even once, that is, the AND unit 34. Once the output becomes high, the output becomes low. The identification value switching unit 35 is configured such that when the output of the holder 36 is High, the switch 39 in the identification value switching unit 35 is closed, and the secondary resistance compensation calculated by the calculator 25 using the current error amplifier output Δeq. Output the value. The adder 37 adds the output of the identification value switch 35 and the preset secondary resistance initial setting value r2c, and outputs it as a secondary resistance identification value.

  As a result, when the motor rotates only in a region where the secondary resistance identification sensitivity is low, the secondary resistance is identified and the induction motor is controlled even if the sensitivity is low. Once identification is performed in a region where the secondary resistance identification sensitivity is high, the induction motor is controlled using an identification value with higher identification accuracy that is identified in a region where the secondary resistance identification sensitivity is high in a region where the identification sensitivity is low. can do.

  The operation of the other components in the induction motor control device according to the prior art will be briefly described below. The dq-axis voltage command calculation unit 4 performs the calculations of equations (15) and (16), and its internal configuration is shown in FIG. The subtractor 11 in the figure subtracts the excitation current detection value id from the excitation current command value id * to output an excitation current error Δid, and the amplifier 12 amplifies Δid and outputs Gd · Δid. The converter 13 and the multiplier 14 calculate the second term of the equation (15) from the torque current command value iq *, the rotation frequency ω, and the leakage inductance Lσ, and add this to Gd · Δid to obtain the d-axis voltage command e1d *. Is output.

  Similarly, the subtractor 15 subtracts the torque current detection value iq from the torque current command value iq * to obtain a torque current error Δiq, which is amplified by the amplifier 16 and the second term Gq · Δiq in the equation (16) is obtained. Has been obtained. The converter 18 multiplies the excitation current command id * by the leakage inductance Lσ, and further multiplies the rotation frequency ω by the multiplier 19 to obtain the first term ωLσ · id * of the equation (16). The third term ωMc · Io * in the equation (16) is output by the multiplier 19 instead of em * = Mc · Io * in the diagram of FIG. 8, and the first term, the second term, The q-axis voltage command e1q * is output after addition. Note that em * is obtained by multiplying the magnetic flux density command φ * by the induced voltage coefficient Kem by the converter 17 of FIG.

  The e1d * and e1q * output from the dq-axis voltage command calculation unit 4 are converted into three-phase AC voltage commands eu *, ev *, and ew * by the two-phase / three-phase converter 3 in FIG. Is done. The inverter 26 uses a DC power supply 27 as an energy source, and applies a voltage corresponding to the three-phase AC voltage commands eu *, ev *, and ew * to the motor 28, whereby a three-phase AC current iu, iv, and iw flows.

  The three-phase alternating currents iu, iv, iw are detected by the current detectors 6a, 6b, 6c, and converted into the excitation current detection value id and the torque current detection value iq by the three-phase two-phase converter 9. Signals sinωt and cosωt used for coordinate conversion by the two-phase three-phase converter 3 and the three-phase two-phase converter 9 are output by the two-phase sine wave generator 10 based on the rotational frequency ω. This ω is obtained by adding the slip frequency ωs to the rotational speed ωm obtained by differentiating the rotational position of the motor 28 detected by the position detector 29 with the differentiator 30.

Japanese Patent Laid-Open No. 10-52100 Japanese Patent No. 3363732

  As described above, in Patent Documents 1 and 2 and the like, a desired output torque can be obtained with high accuracy by automatically compensating for a change in the secondary resistance R2 due to a temperature change of the motor. In order to compensate the slip frequency ωs by compensating the secondary resistance R2, the compensation amount is added to the rotational frequency ω of the magnetic flux.

  On the other hand, many induction motors used for driving a spindle of a machine tool are often used at high speed and have a high rotational frequency ω. In the current control by PWM of the inverter, as ω becomes larger, the period 1 / ω of the sine wave current becomes smaller and the ratio to the PWM carrier frequency becomes smaller, and the number of times of switching within one current period is reduced. Gets worse. Furthermore, due to this influence, the voltage generated inside the motor becomes high, and when decelerating from high-speed rotation, the current is suddenly changed from a state where ω is large, so that the current controllability is the worst.

  When the secondary resistance R2 is compensated, ω increases, and the current controllability during motor deceleration further deteriorates. In some cases, the current that flows from the motor to the inverter cannot be controlled during deceleration, and a current that greatly exceeds the rated current of the device flows, causing the protection circuit to work or machine down, or in the worst case, the device will fail There is a case.

  In order to solve the above-described problems, an induction motor control device according to the present invention is an induction motor control device driven by a three-phase alternating current converted from a direct current, and includes a torque command and a magnetic flux density command. In order to control the primary current of the motor, the phase command is converted into a three-phase voltage command calculated based on the excitation current common-mode voltage command and the torque current common-mode voltage command, and the actual three-phase primary of the motor In the induction motor control device that performs feedback control by converting the current into a two-phase detection value of the torque current detection value iq and the excitation current detection value id, the excitation current command value id * is calculated based on the magnetic flux density command. Excitation current command generation means, excitation voltage command calculation means for calculating an excitation current in-phase voltage command in phase with the excitation current based on the excitation current command id * and the excitation current detection value id Torque current command generating means for calculating a torque current command iq * based on the torque command and the magnetic flux density command, and a torque current in-phase with the torque current based on the torque current command iq * and the torque current detection value iq A torque voltage command calculation means for calculating a voltage command, and a secondary resistance correction value of the motor is calculated by multiplying the torque current common-mode voltage command by a secondary resistance identification gain, and a ratio A ( = Torque current / excitation current) is equal to or less than a predetermined value, a correction value for the secondary resistance calculated when the ratio A between the torque current and the excitation current exceeds a predetermined value. The secondary resistance correction value calculating means for holding and outputting, the torque current command iq * is gradually calculated by the magnetic flux density command, and further corrected by the secondary resistance correction value. A slip frequency calculating means for calculating a slip frequency by multiplying the secondary resistance value R2 of the motor, and a frequency command calculating means for calculating a frequency command by adding the detected value of the slip frequency and the rotational speed of the motor; The secondary resistance identification correction value calculation means selects either Gr1 or Gr2 smaller than Gr1 as the secondary resistance identification gain according to the rotation direction of the motor and the torque command. And output.

  In the induction motor control apparatus according to another embodiment of the present invention, the secondary resistance identification correction value calculating means may determine the secondary resistance identification when the rotation direction of the motor matches the polarity of the torque command. The gain is set to Gr1, and if they do not match, the secondary resistance identification gain is set to Gr2. In another form, the secondary resistance identification correction value calculating means, when the rotation direction of the motor matches the polarity of the torque command, or when the rotation speed of the motor is less than a specified reference value, If the secondary resistance identification gain is Gr1, and the rotation direction of the motor does not match the polarity of the torque command and the rotation speed of the motor exceeds a specified reference value, the secondary resistance identification gain is Gr2. .

  According to the present invention, since the secondary resistance identification gain is switched in accordance with the rotation direction of the motor and the torque command, the current controllability is stabilized and the internal voltage of the motor is prevented from becoming extremely large. It is possible to prevent a situation in which a current that greatly exceeds the rated current of the device flows and the protection circuit operates to cause a machine down or the device breaks down.

It is a figure which shows one Embodiment of the control apparatus of the induction motor by this invention. It is a block diagram which shows the internal structure of the secondary resistance identification correction value calculation means in the control apparatus of the induction motor by this invention. It is a figure which shows the relationship of the secondary identification gain switching in Embodiment 1. FIG. It is a figure which shows the relationship of the secondary identification gain switching in Embodiment 2. It is a block diagram of the induction motor control apparatus by a prior art. It is a block diagram of the induction motor control apparatus which uses the exact identification value identified when identification sensitivity is high even when the secondary resistance identification sensitivity by a prior art is low. FIG. 7 is a block diagram of a secondary resistance identifier according to FIG. 6. It is a block diagram of the voltage command calculation part of the induction motor control apparatus by a prior art. It is an equivalent circuit diagram of an induction motor. It is a figure which shows the relationship between the product of an angular frequency, an angular frequency, and a magnetic flux density command.

  FIG. 1 is a block diagram of an embodiment of an induction motor control apparatus according to the present invention. The same components as those of the conventional induction motor control device shown in FIG. 6 are denoted by the same reference numerals, and the description thereof will be omitted because it is duplicated.

  An example of the internal configuration of the secondary resistance compensator 5 is shown in the block diagram of FIG. The secondary resistance compensation unit 5 of the present embodiment includes a secondary resistance identification gain selection unit 40. The secondary resistance identification gain selection unit 40 selects an identification gain corresponding to the input frequency ω and torque command T *. Specifically, in the present embodiment, when the input frequency ω and the polarity of the torque command T * coincide, the identification gain Gr1 is used to correct the change in the secondary resistance due to the temperature change. On the other hand, when the polarity of the frequency ω and the torque command T * do not match (that is, during deceleration), correction is performed using the identification gain Gr2. Gr2 is set smaller than Gr1.

  FIG. 3 shows the relationship between the frequency ω, the torque command T *, and the identification gains Gr1 and Gr2 according to FIG. That is, when the frequency ω and the polarity of the torque command T * match, the secondary resistance R2 is corrected with the identification gain Gr1 having a large set value. On the other hand, when the polarity of the frequency ω and the torque command T * do not match, that is, when the motor decelerates, the secondary resistance R2 is corrected with the identification gain Gr2 having a small set value. As shown in the lower part of FIG. 2, the secondary resistance identification gain selection unit 40 includes comparators 41 and 42 that compare the frequency ω or the torque command T * with the value 0, and outputs of the two comparators 41 and 42 are , XOR operation is performed by the XOR unit 45. The output from the XOR unit 45 is input to the switch 43 as it is, and is inverted by the NOT unit 48 to be input to the switch 44. When the switch 43 is ON, Gr1 is output. When the switch 44 is ON, Gr2 is output.

  FIG. 4 shows the relationship between the frequency ω, the torque command T *, and the identification gains Gr1 and Gr2 according to the second embodiment of the secondary resistance. The fact that the current controllability is deteriorated by identifying the secondary resistance R2 is most conspicuous during deceleration from high-speed rotation. Therefore, in the second embodiment, the secondary resistance R2 is identified using the identification gain Gr2 having a small set value only when decelerating from high-speed rotation, that is, only when decelerating at a speed higher than the frequency ω1. In other cases, identification is performed using Gr1. Various configurations of the secondary resistance identification gain selection unit that outputs the identification gains Gr1 and Gr2 are conceivable. For example, in the configuration illustrated in the lower part of FIG. 2, a frequency is further provided between the switch 44 and the amplifier 47. You may make it incorporate the circuit which switches the identification gain output according to (omega) to Gr1 or Gr2.

  According to the induction motor control device of the present invention, the identification gain is set to Gr1 when the rotation direction of the motor matches the polarity of the torque command, and the identification gain is set to Gr2 when they do not match. By setting Gr2 to a small value, it is possible to obtain a desired output torque with high accuracy by identifying the secondary resistance by Gr1 except when the motor is easily decelerated. When the motor is easily decelerated, it is possible to prevent the current controllability from deteriorating by identifying the secondary resistance using the identification gain Gr2 having a small set value. As a result, the current controllability from high-speed rotation is stabilized and it is possible to prevent the voltage inside the motor from becoming extremely large. A current that greatly exceeds the rated current of the device flows, and the protection circuit operates to bring down the machine. It is possible to prevent a situation in which an accident occurs or a device breaks down.

1, 8, 13, 17, 18, 20 converter, 2, 7 divider, 3 2-phase 3-phase converter, 4 dq-axis voltage command calculation unit, 5 secondary resistance compensation unit, 6 current detector, 9 3 Phase 2-phase converter, 10 2-phase sine wave generator, 11, 15 subtractor, 12, 16, 25, 46, 47 amplifier, 14, 19 multiplier, 26 inverter, 27 DC power supply, 28 motor, 29 position detection , 30 Differentiator, 31 Secondary resistance identifier, 32, 33, 41, 42 Comparator, 34 AND device, 35 Identification value switcher, 36 Holder, 37 Adder, 38, 39, 43, 44 Switch, 40 Secondary resistance identification gain selection unit, 45 XOR unit, 48 NOT unit.

Claims (3)

A control device for an induction motor driven by a three-phase alternating current converted from a direct current,
The two-phase command of the torque command and the magnetic flux density command is converted into a three-phase voltage command calculated based on the excitation current common-mode voltage command and the torque current common-mode voltage command in order to control the primary current of the motor, and the motor In the induction motor control apparatus that converts the actual three-phase primary current of the current into a two-phase detection value of the torque current detection value iq and the excitation current detection value id, and performs feedback control.
Excitation current command generating means for calculating an excitation current command value id * based on the magnetic flux density command;
Excitation voltage command calculation means for calculating an excitation current common-mode voltage command in phase with the excitation current based on the excitation current command id * and the excitation current detection value id;
Torque current command generating means for calculating a torque current command iq * based on the torque command and the magnetic flux density command;
Torque voltage command calculating means for calculating a torque current common-mode voltage command in phase with the torque current based on the torque current command iq * and the torque current detection value iq;
A secondary resistance correction value of the motor is calculated by multiplying the torque current common-mode voltage command by a secondary resistance identification gain, and a ratio A (= torque current / excitation current) between the torque current and the excitation current is predetermined. A secondary resistance correction value calculating means for holding and outputting a correction value of the secondary resistance calculated when the ratio A between the torque current and the excitation current exceeds a predetermined value when the torque current is less than the predetermined value; ,
Slip frequency calculating means for calculating a slip frequency by multiplying the torque current command iq * by the magnetic flux density command and multiplying by the secondary resistance value R2 of the motor corrected by the secondary resistance correction value. When,
A frequency command calculating means for calculating a frequency command by adding the detected value of the slip frequency and the rotational speed of the motor;
Have
The secondary resistance identification correction value calculating means selects and outputs either Gr1 or Gr2 smaller than Gr1 as the secondary resistance identification gain according to the rotation direction of the motor and the torque command.
An induction motor control device characterized by the above. The induction motor control device according to claim 1,
The secondary resistance identification correction value calculation means sets the secondary resistance identification gain to Gr1 when the rotation direction of the motor matches the polarity of the torque command, and sets the secondary resistance identification gain to Gr2 when they do not match. And
An induction motor control device characterized by the above. The induction motor control device according to claim 1,
The secondary resistance identification correction value calculating means calculates the secondary resistance identification gain when the rotation direction of the motor matches the polarity of the torque command, or when the rotation speed of the motor is equal to or less than a predetermined reference value. Gr1, and when the rotation direction of the motor does not match the polarity of the torque command and the rotation speed of the motor exceeds a specified reference value, the secondary resistance identification gain is Gr2. Motor control device.

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