JP2013061161A - Rotation angle detecting device - Google Patents

Abstract

PROBLEM TO BE SOLVED: To provide a rotation angle detecting device dispensing with a dedicated circuit such as a multiplying DAC, a SIN-ROM table, and COS-ROM table.SOLUTION: A rotation angle detecting device comprises: an A/D converter 3A for performing the A/D conversion on a rotation detection signal sinθ*f(t); an A/D converter 3B for performing the A/D conversion on a rotation detection signal cosθ*f(t); Fourier analysis means 4A for extracting an amplitude intensity sinθ of the fundamental wave component from the rotation detection signal sinθ*f(t) converted by the A/D converter 3A; Fourier analysis means 4B for extracting an amplitude intensity cosθ of the fundamental wave component from the rotation detection signal cosθ*f(t) converted by the A/D converter 3B; a composite wave generator/phase controller 5 for generating a composite wave sin(θ-φ) from the amplitude intensities sinθ and cosθ, and an internally estimated angle φ, and for controlling the internally estimated angle φ to be identical with a rotation angle θ on the basis of a polarity of the composite wave sin(θ-φ); and an excitation signal generator 7 for generating a sine wave or a binary excitation signal for the rotation angle detecting device.

Description

  The present invention relates to a rotation angle detection device that converts an analog angle detection signal output from a rotation detector into a digital angle signal.

  As a rotation angle detection device, a resolver with 1-phase excitation and 2-phase output is used as a rotation detector, and two output signals cosθ and sinθ having a 90-degree phase difference obtained from the resolver are processed by a dedicated integrated circuit. An R / D converter that detects a rotational angle θ of a resolver is known (for example, see Patent Document 1).

  As shown in FIG. 6, this R / D converter is based on angle information φ output from an up / down counter 32 whose up / down is controlled by an up / down signal output from a compensator 31, and COS − The peak values cosφ / sinφ of the sine wave / cosine wave are sequentially read from the ROM table 33 and the SIN-ROM table 34. These are multiplied by sinθ · f (t) and cosθ · f (t) output from the two-phase secondary windings of the resolver 1 in the 10-bit multiplying DACs 35 and 36, respectively, and sinθ · f ( t) · cosφ and cosθ · f (t) · sinφ are obtained, the difference sin (θ−φ) · f (t) is obtained by the subtractor 37, the polarity is detected by the comparator 38, and the synchronous detector 39 is obtained. Then, the excitation component f (t) is removed to obtain the control component sin (θ−φ), which is input to the compensator 31, and the up / down signal is generated according to θ> φ and θ <φ. The tracking loop is configured as follows. Then, the rotation angle θ of the rotor of the resolver 1 is obtained by controlling so that θ−φ becomes zero. Reference numeral 40 denotes an excitation signal generator for generating an excitation signal f (t). This excitation signal f (t) is applied to the primary winding of the resolver 1 and is used for synchronous detection.

Japanese Patent No. 3442316

  However, when a dedicated integrated circuit such as Patent Document 1 is used, the integrated circuit such as the 10-bit multiplying DACs 35 and 36, the synchronous detector 39, the COS-ROM table 33, and the SIN-ROM table 34 has a very large area scale. There are many large circuits and the processing becomes complicated, which increases the cost. In particular, when using the COS-ROM table 33 and the SIN-ROM table 34 to increase the angular accuracy, the higher the angular resolution, the more precisely each peak value must be stored in the ROM. Problem arises. Interpolation processing is a method for improving accuracy without enlarging the table, but in this case, the circuit configuration and processing are further complicated.

  An object of the present invention is to provide a rotation angle detection device that eliminates the need for dedicated circuits such as a multiplying DAC, a SIN-ROM table, and a COS-ROM table.

In order to achieve the above object, the rotation angle detection device according to the first aspect of the present invention digitally converts the two-phase rotation detection signals sinθ · f (t) and cosθ · f (t) output from the rotation detector. In the rotation angle detecting device for obtaining the rotation angle θ, the rotation detection signal sinθ · f (t) is A / D converted by a first A / D conversion means, and the rotation detection signal cosθ · f (t) is converted to A Second A / D conversion means for D / D conversion, and first Fourier for extracting amplitude intensity sin θ of the fundamental wave component from the rotation detection signal sin θ · f (t) converted by the first A / D conversion means Analysis means, second Fourier analysis means for extracting the amplitude intensity cosθ of the fundamental wave component from the rotation detection signal cosθ · f (t) converted by the second A / D conversion means, and the amplitude intensity sinθ, cosθ And a synthesized wave sin (θ−φ) from the internal estimated angle φ and the internal estimated angle φ based on the polarity of the synthesized wave sin (θ−φ) And a synthetic wave generation / phase control means for controlling the rotation angle θ to coincide with the rotation angle θ, and an excitation signal generation means for generating a sine wave or binary excitation signal for the rotation detector. .
According to a second aspect of the present invention, in the rotation angle detection device according to the first aspect, the first and second Fourier analysis means are input to a first integrator and an output side of the first integrator. A first ω coefficient unit connected to the output side, a second integrator connected to the output side of the first ω coefficient unit, and an output side connected to the output side of the second integrator. Each of which is composed of a two-phase oscillator comprising a second ω coefficient unit connected to the input side of the first integrator, and the first Fourier analysis means is connected to the first integrator to detect the rotation. The signal sinθ · f (t) is updated and set, the amplitude intensity sinθ is extracted from the output side of the first integrator, and the second Fourier analysis means detects the rotation in the first integrator. The signal cosθ · f (t) is updated and set, and the amplitude intensity cosθ is extracted from the output side of the first integrator. It is characterized by that.
According to a third aspect of the present invention, in the rotation angle detection device according to the first aspect, the synthesized wave generation / phase control means includes a first integrator and an input side connected to an output side of the first integrator. The first ω coefficient unit, the second integrator whose input side is connected to the output side of the first ω coefficient unit, the input side is connected to the output side of the second integrator, and the output side is the aforementioned The second integrator is composed of a second ω coefficient unit connected to the input side of the first integrator, the amplitude intensity sin θ is updated and set in the first integrator, and the second integrator The amplitude intensity cos θ is updated and set in an integrator, and the combined wave sin (θ−φ) is generated from the output side of the first integrator, and based on the polarity of the combined wave sin (θ−φ) The polarities of the first ω coefficient unit and the second ω coefficient unit are controlled so that the internal estimated angle φ matches the rotation angle θ. It is characterized by.
According to a fourth aspect of the present invention, in the rotation angle detecting device according to the first aspect, the first and second Fourier analysis means, the synthesized wave generation / phase control means, and an excitation signal generation means for generating a sine wave Includes a first integrator, a first ω coefficient unit whose input side is connected to the output side of the first integrator, and a second one whose input side is connected to the output side of the first ω coefficient unit. And a second ω-coefficient generator having an input side connected to the output side of the second integrator and an output side connected to the input side of the first integrator. The first and second Fourier analysis means, the synthesized wave generation / phase control means, and the excitation signal generation means operate in a time-sharing manner, and the values of the first integrator and the second integrator Is replaced according to the time-sharing operation.

  According to the rotation angle detection device of the present invention, dedicated circuits such as a multiplying DAC, a synchronous detector, a compensator, a SIN-ROM table, and a COS-ROM table, which are conventionally required, are not required. In particular, since the multiplying DAC, the SIN-ROM table, and the COS-ROM table have a large ratio to the chip area in the semiconductor device, the chip area can be significantly reduced by reducing the cost and the cost can be reduced. Further, even when the present invention is realized by a DSP, a microcomputer, etc., there is an advantage that it can be realized by a cheaper DSP, a microcomputer, etc. Furthermore, since the two-phase oscillator itself can be configured with a general logic circuit, a Fourier analyzer, synthetic wave generation / phase control means, and excitation signal generation means that apply the two-phase oscillator can also be realized with a general circuit configuration for signal processing. With respect to the entire unit, the chip area can be reduced in the semiconductor device, and the processing is lightened in the DSP, microcomputer, etc., and there is an advantage that it can be realized at a lower cost.

It is a block diagram which shows the structure of the rotation angle detection apparatus of one Example of this invention. It is explanatory drawing which shows the structure of a digital two phase oscillator. It is explanatory drawing which shows Fourier analyzer 4A, 4B comprised by applying a two-phase oscillator. It is a block diagram which shows the synthetic wave production | generation / phase controller 5 comprised applying the two-phase oscillator. It is explanatory drawing of the structure which performs the processing of Fourier analyzer 4A, 4B, the synthetic wave production | generation / phase controller 5, and the excitation signal generator 7 by a common 2 phase oscillator by a time division. It is a block diagram which shows the structure of the conventional R / D converter.

  Embodiments of the rotation angle detection device of the present invention will be described below. FIG. 1 is a block diagram showing the configuration of an embodiment of the present invention. Reference numeral 1 denotes a resolver body which is a rotation detector, and 2 denotes a rotation angle detector. In the rotation angle detector 2, 3A is a sine wave side A / D converter, 3B is a cosine wave side A / D converter, 4A is a sine wave side Fourier analyzer, and 4B is a cosine wave side Fourier analysis. It is a vessel. The A / D converters 3A and 3B have the same circuit configuration, and the Fourier analyzers 4A and 4B have the same circuit configuration. Reference numeral 5 denotes a composite wave generator / phase controller, 6 denotes a controller, and 7 denotes an excitation signal generator. The Fourier analyzers 4A and 4B, the synthesized wave generator / phase controller 5 and the excitation signal generator 7 are all configured by a circuit based on a two-phase oscillator.

  FIG. 2 shows the configuration of a digital two-phase oscillator, where 8 is an adder, 9A and 9B are integrators, and 10A and 10B are ω coefficient units. The integrators 9A and 9B include a register 91, a 1-δ coefficient unit 92 for preventing output divergence of the two-phase oscillator, and an adder 93. When the two-phase oscillator is digitally expressed, the coefficient values of the ω coefficient units 10A and 10B and the 1-δ coefficient unit 92 can be realized by combining a bit shift circuit and an adder.

式（１）を逆ラプラス変換すると、Z(t)＝Ａcosωtとなる。
同様に、左下側の出力Z'(s)は以下のように求められる。

式（２）を逆ラプラス変換すると、Z'(t)＝−Ａsinωtとなる。つまり、２相発振器においては同一の入力信号（振幅Ａのインパルス）に対して、Ａcosωt、−Ａsinωtの２相の出力を同時に得ることができる。 The two-phase oscillator itself is a known technique, but its principle will be briefly described. When integrators 9A and 9B have a transfer function 1 / s and an impulse of amplitude A is input to adder 8 in FIG. 2 as an initial value of oscillation, output Z (s) on the upper right side is obtained as follows.

When equation (1) is subjected to inverse Laplace transform, Z (t) = Acosωt.
Similarly, the lower left output Z ′ (s) is obtained as follows.

When Expression (2) is subjected to inverse Laplace transform, Z ′ (t) = − Asinωt. That is, in the two-phase oscillator, two-phase outputs of Acosωt and −Asinωt can be simultaneously obtained with respect to the same input signal (impulse of amplitude A).

The application of this principle is the Fourier analyzers 4A and 4B, and FIG. 3 shows the transfer function of the Fourier analyzer. The integrators 41A and 41B are composed of a register 411, a 1-δ coefficient unit 412 and an adder 413. Reference numerals 42A and 42B denote ω coefficient units. For the idea, refer to the literature (Fuminori Kobayashi, Michio Nakano, “Analog Fourier Analyzer Applying Two-Phase Oscillator”, IEICE Transactions, Trans.IECE '80 / 6 Vol.63-C No.6) I have to. The detailed description will be given in the above document, but in brief, when a periodic signal f (t) (= f (t + nT)) having a component of ω is input to the adder 43 of the system of FIG. Therefore, from the cos side output after T period, a value (primary component) proportional to the amplitude value of cosωt of the signal is output, and from the sin side output A value (primary component) proportional to the amplitude value of sinωt of the signal is output.

  Thus, in the Fourier analyzers 4A and 4B in FIG. 1, if each adder 43 is input with sin θ · f (t) and cos θ · f (t) output from the A / D converters 3A and 3B, respectively. Since the frequency component f (t) is eliminated, sinθ corresponding to the amplitude value of the excitation signal f (t) is obtained from the Fourier analyzer 4A on the sinθ · f (t) side, and Fourier analysis on the cosθ · f (t) side is obtained. Cos θ is obtained from the device 4B. Further, in the Fourier analyzers 4A and 4B, since the excitation component f (t) is removed, synchronous detection becomes unnecessary. In the present invention, a sinusoidal wave having a ω component is used for the excitation signal (f (t) = sinωt). However, it is clear that a similar analysis result can be obtained using a square wave, In that case, the excitation signal generator 7 can be configured with only a frequency divider without using a two-phase oscillator, and its output is only two kinds of voltage levels of “Low / High”. For example, if an open drain output terminal or the like is used, signal amplification can be performed very efficiently only by changing the power supply voltage for pull-up.

The synthesized wave generator / phase controller 5 obtains the control deviation ε = sin (θ−φ) from the amplitude values sinθ and cosθ obtained from the Fourier analyzers 4A and 4B. The synthesized wave generator / phase controller 5 can also be realized by application of a two-phase oscillator. The transfer function of the composite wave generator / phase controller 5 is shown in FIG. The composite wave generator / phase controller 5 is a superposition of two-phase oscillators, and its operation is as follows. In FIG. 4, 51A and 51B are integrators, 52A and 52B are ω coefficient units, and 53A and 53B are adders. The integrators 51A and 51B include a register 511, a 1-δ coefficient unit 512, and an adder 513.

Consider a case where an impulse of amplitude A is input to the adder 53A of FIG. In this case, an Acosωt signal is obtained from the output on the upper right side. Similarly, consider a case where an impulse having an amplitude value B is input to the adder 53B in FIG. In this case, a signal of −Bsinωt is obtained from the output on the upper right side. Therefore, considering the case where the amplitudes A and B are input to the two-phase oscillator system at the same timing, Acosωt−Bsinωt that is the sum of the outputs can be obtained from the output on the upper right side.

Here, in the block diagram of FIG. 1, when the amplitude A input to the combined wave generator / phase controller 5 is sin θ, the amplitude B is cos θ, and the angle ω t is the internal estimated angle φ, the above combined wave is obtained from the trigonometric addition theorem. The output is as follows:

  From this equation (3), it can be seen that the signal output from the combined wave generator / phase controller 5 of FIG. 1 is the control deviation ε itself. In this control deviation ε = sin (θ−φ), θ is the rotation angle of the resolver 1 and φ is the internal estimated angle, so that both are matched, that is, the composition of FIG. The wave generator / phase controller 5 operates. Thereby, θ can be detected from the rotation angle θ−φ = 0.

  Specifically, since the control deviation ε is represented by a two's complement expression, positive / negative discrimination can be made with the most significant sign bit (MSB), and the MSB of the register 511 of the upper integrator 51A in FIG. In the case of ', θ> φ, so the phase of the internal estimation angle φ is advanced while the signs of the ω coefficient units 52A and 52B in FIG. Conversely, when the MSB is ‘1’, θ <φ, so the sign of the ω coefficient units 52A and 52B in FIG. 4 is reversed to delay the phase of the internal estimated angle φ. This converges to φ = θ.

となる。一方、ε＜０、つまりＭＳＢ＝‘１’のときは、

となる。このように、合成波生成・位相制御器５の内部で制御偏差εのフィードバック制御を行うことで、常にθ＝φを得ることができる。 When this is actually expressed by an expression including the concept of the clock period ΔT, it is as follows. When the control deviation ε is ε> 0, that is, MSB = '0',

It becomes. On the other hand, when ε <0, that is, when MSB = '1',

It becomes. In this way, by performing feedback control of the control deviation ε inside the composite wave generator / phase controller 5, θ = φ can always be obtained.

  Further, the controller 6 uses the MSB obtained by the synthesized wave generator / phase controller 5 as an up / down signal to increase / decrease the count value of the up / down counter representing the internal estimated angle φ at the same timing. For this reason, the internal estimated angle φ of the synthesized wave generation / phase controller 5 expressed by the equation (3) and the internal estimated angle φ output from the controller 6 are always synchronized. That is, the controller 6 outputs a signal representing the rotation angle θ (= φ) of the resolver 1.

In FIG. 1, the excitation signal generator 7 is the two-phase oscillator itself shown in FIG. 2, and outputs the excitation component f (t) = sinωt including the component ω to the excitation side of the rotation detector 1.

  In the embodiment of the present invention, a sine wave is used for the excitation signal f (t). However, as described in the description of the Fourier analyzer, a similar result can be obtained by using a square wave. When a square wave is used, an excitation signal can be generated using a frequency divider. Therefore, the present invention can be realized even when the present invention is realized by a semiconductor device such as a microcomputer that does not have a DAC output.

  As can be seen from the above description and FIGS. 2 to 4, the Fourier analyzers 4 </ b> A and 4 </ b> B, the synthesized wave generator / phase controller 5, and the excitation signal generator 7, which are main components in the present invention, are all two-phase. It is constituted by almost the same circuit based on an oscillator.

  Since the two-phase oscillator itself is configured with a simple circuit as is apparent from FIG. 2, when the rotation angle detection device according to the present invention is configured with a dedicated semiconductor device, there is a patent other than the A / D conversion circuit. The dedicated circuit as in Document 1 is not required, and the hardware can be greatly simplified. Even in the case of a DSP, microcomputer, etc., it can be realized by a circuit generally provided by the DSP, microcomputer, etc., and does not require complicated processing, so it can be realized by a cheaper DSP, microcomputer. Is possible.

  Now, the excitation signal f (t) (= sinωt) generated by the excitation signal generator 7 is generated by the two-phase oscillator of FIG. 2, and sinθ and cosθ generated by the Fourier analyzers 4A and 4B are represented by 2 in FIG. When the composite wave sin (θ−φ) generated by the phase oscillator and generated by the composite wave generator / phase controller 5 is generated by the two-phase oscillator of FIG. 4, the 1-δ coefficient constituting these two-phase oscillators Can be shared by switching the combination, and at this time, the register that configures the integrator replaces the data set there Can also share the register.

  FIG. 5 is a diagram showing a configuration in the case where three types of circuits of the excitation signal generator 7, the Fourier analyzers 4A and 4B, and the synthesized wave generator / phase controller 5 are realized by one two-phase oscillator as described above. . The integrators 60A and 60B include a register 61, a 1-δ coefficient unit 62, an adder 63, and a selector 64. The 1-δ coefficient unit 62 includes a bit shift circuit group 621, selectors 622 to 624, and an adder 625. The ω coefficient units 70A and 70B include a bit shift circuit group 71, selectors 72 to 74, an adder 75, a multiplier 76 having a gain of -1, and a selector 77.

When the circuit of FIG. 5 is used as the excitation signal generator 7, the integrators 60A and 60B select 1-δ coefficients obtained by a predetermined bit shift circuit of the bit shift circuit group 621 using the selectors 622 to 624. Then, an initial value (a value corresponding to the initial pulse A in FIG. 2) input externally is taken into the register 61 by the selector 64 of one integrator 60A. The selector 64 of the other integrator 60B selects the output of the adder 63. In the ω coefficient units 70A and 70B, the ω coefficient obtained by a predetermined bit shift circuit of the bit shift circuit group 71 is selected by the selectors 72 to 74, and the polarity of the ω coefficient is selected by the selector 77. For example, when T = 50, the value of the ω coefficient is set by ω = 2π / T and ω = 0.125663706... ≈2 ⁻³ +2 ⁻¹¹ +2 ⁻¹³ . Thereby, f (t) = sinωt is generated from the integrator 60B. However, the number of selectors of this bit shift circuit (determining the 1-δ coefficient and the ω coefficient) is not limited to three, but is determined by the required accuracy.

  When the circuit of FIG. 5 is used as the Fourier analyzers 4A and 4B, each of the Fourier analyzers 4A and 4B is configured by the circuit of FIG. In integrators 60A and 60B, selectors 622 to 624 select 1-δ coefficients obtained by a predetermined bit shift circuit of bit shift circuit group 621. Then, the value (sinθ · f (t) or cosθ · f (t)) input from the A / D converters 3A and 3B in FIG. 1 is updated by the selector 64 of one integrator 60A. Each time, an initial value is taken into the register 61 and oscillation is performed. The selector 64 of the other integrator 60B selects the output of the adder 63. In the ω coefficient units 70A and 70B, the ω coefficient obtained by a predetermined bit shift circuit of the bit shift circuit group 71 is selected by the selectors 72 to 74, and the polarity of the ω coefficient is selected by the selector 77. Thereby, cos θ or sin θ, which is the amplitude value of the input signal, is generated from the integrator 60A.

  When the circuit of FIG. 5 is used as the synthesized wave generator / phase controller 5, the integrators 60A and 60B use selectors 622 to 624 to obtain 1-δ coefficients obtained by a predetermined bit shift circuit of the bit shift circuit group 621. select. Then, the selector 64 of one integrator 60A takes in the value (sinθ) input from the Fourier analyzer 4A of FIG. 1 into the register 61 as an initial value every time the value is updated, and the selector of the other integrator 60B. 64, the value (cos θ) input from the Fourier analyzer 4B of FIG. 1 is loaded into the register 61 as an initial value every time the value is updated (that is, simultaneously with the loading of cos θ), and oscillation is performed. Further, in the ω coefficient units 70A and 70B, the ω coefficient obtained by a predetermined bit shift circuit of the bit shift circuit group 71 is selected by the selectors 72 to 74, and according to the MSB value of the data in the register 61 on the integrator 60A side. The selector 77 selects the polarity of the ω coefficient. Thereby, sin (θ−φ) is generated from the integrator 60A.

  In the configuration of FIG. 5, the operations of the three types of four two-phase oscillators such as the excitation signal generator 7, the Fourier analyzers 4 </ b> A and 4 </ b> B, and the synthesized wave generator / phase controller 5 are performed by a time division operation. . In this time division operation, when the excitation signal generator 7 is operated, the values of the registers 61 of the integrators 60A and 60B generated by the operations of the Fourier analyzers 4A and 4B and the synthesized wave generator / phase controller 5 are used. When temporarily storing in a memory (register) (not shown) and operating as one of the Fourier analyzers 4A and 4B, the other of the Fourier analyzers 4A and 4B, the excitation signal generator 7 and the combined wave generator / phase controller 5 When the value of the register 61 of the integrators 60A and 60B generated by the operation is temporarily saved in the memory (register) and operated as the synthesized wave generator / phase controller 5, the Fourier analyzers 4A and 4B and the excitation signal generator are generated. The value of the register 61 of the integrators 60A and 60B generated by the operation of the generator 7 is temporarily saved in the memory (register). When each of the excitation signal generator 7, the Fourier analyzers 4 A and 4 B, and the combined wave generator / phase controller 5 is operated, the memory value is taken from the selector 64 and returned to the register 61 to be set. Let the action take place. By such a method, it is possible to use only one register 61 for each of the integrators 60A and 60B, and the circuit elements can be greatly reduced.

  The rotation angle detection device of the present invention is suitable for realizing signal processing of a rotation detector that outputs signals having phases different from each other by 90 degrees in addition to the resolver 1 having a single-phase excitation and two-phase output described in the above embodiments. .

1: Resolver 2: Rotation angle detector 3A, 3B: A / D converter, 4, 4A, 4B: Fourier analyzer, 5: Synthetic wave generator / phase controller, 6: Controller, 7: Excitation signal generator 8: adder, 9A, 9B: integrator, 91: register, 92: 1-δ coefficient multiplier, 93: adder, 10A, 10B: ω coefficient multiplier 41A, 41B: integrator, 411: register, 412: 1 -Δ coefficient unit, 413: adder, 42A, 42B: ω coefficient unit, 43: adder 51A, 51B: integrator, 511: register, 512: 1-δ coefficient unit, 513: adder, 52A, 52B: ω coefficient unit, 53A, 53B: adder 60A, 60B: integrator, 61: register, 62: 1-δ coefficient unit, 621: bit shift circuit group, 622-624: selector, 625: adder, 63: addition , 64: selector 70A, 7 B: omega coefficient unit, 71: bit shift circuit groups 72-74: selector, 75: adder, 76: multiplier, 77: selector

Claims (4)

In the rotation angle detection device for obtaining the rotation angle θ by digitally converting the two-phase rotation detection signals sinθ · f (t) and cosθ · f (t) output from the rotation detector,
First A / D conversion means for A / D converting the rotation detection signal sinθ · f (t);
Second A / D conversion means for A / D converting the rotation detection signal cos θ · f (t);
First Fourier analysis means for extracting the amplitude intensity sinθ of the fundamental wave component from the rotation detection signal sinθ · f (t) converted by the first A / D conversion means;
Second Fourier analysis means for extracting the amplitude intensity cosθ of the fundamental wave component from the rotation detection signal cosθ · f (t) converted by the second A / D conversion means;
A composite wave sin (θ−φ) is generated from the amplitude intensities sinθ, cosθ and the internal estimated angle φ, and the internal estimated angle φ is made to coincide with the rotation angle θ based on the polarity of the composite wave sin (θ−φ). Synthetic wave generation / phase control means for controlling
Excitation signal generating means for generating a sine wave or binary excitation signal to the rotation detector;
A rotation angle detection device comprising: The rotation angle detection device according to claim 1,
The first and second Fourier analysis means include a first integrator, a first ω coefficient unit whose input side is connected to an output side of the first integrator, and a first ω coefficient unit. A second integrator whose input side is connected to the output side, and a second ω coefficient unit whose input side is connected to the output side of the second integrator and whose output side is connected to the input side of the first integrator Each is composed of a two-phase oscillator consisting of
The first Fourier analysis means updates and sets the rotation detection signal sinθ · f (t) in the first integrator, extracts the amplitude intensity sinθ from the output side of the first integrator,
In the second Fourier analysis means, the rotation detection signal cosθ · f (t) is updated and set in the first integrator, and the amplitude intensity cosθ is extracted from the output side of the first integrator. ,
A rotation angle detection device characterized by that. The rotation angle detection device according to claim 1,
The synthetic wave generation / phase control means includes a first integrator, a first ω coefficient unit whose input side is connected to an output side of the first integrator, and an output side of the first ω coefficient unit A second integrator whose input side is connected to the second integrator, and a second ω coefficient multiplier whose input side is connected to the output side of the second integrator and whose output side is connected to the input side of the first integrator. Consisting of a two-phase oscillator
The amplitude intensity sinθ is updated and set in the first integrator, the amplitude intensity cosθ is updated and set in the second integrator, and the combined wave sin is output from the output side of the first integrator. (Θ−φ) is generated, and based on the polarity of the combined wave sin (θ−φ), the first ω coefficient unit and the second ω coefficient are set so that the internal estimated angle φ matches the rotation angle θ. The rotation angle detection device is characterized in that the polarity of the vessel is controlled. The rotation angle detection device according to claim 1,
The first and second Fourier analysis means, the synthesized wave generation / phase control means, and the excitation signal generation means for generating a sine wave are input to a first integrator and an output side of the first integrator. A first ω coefficient unit connected to the output side, a second integrator connected to the output side of the first ω coefficient unit, and an output side connected to the output side of the second integrator. A common two-phase oscillator consisting of a second ω coefficient unit connected to the input side of the first integrator,
The first and second Fourier analysis means, the combined wave generation / phase control means, and the excitation signal generation means operate in a time division manner, and the values of the first integrator and the second integrator are: A rotation angle detection device, wherein the rotation angle detection device is replaced in accordance with the time division operation.

Images