JP2012253943A - Rotary machine controller - Google Patents

Abstract

PROBLEM TO BE SOLVED: To provide a nevel rotary machine controller that controls control amounts having at least one of current flowing in the rotary machine, torque, and magnetic flux by switching on and off a switching element of a DC-to-AC converting circuit.SOLUTION: An operation state (voltage vector) in which an evaluation function J obtained by adding an offset value Δ to the inner product of the difference between predicted currents ide and iqe predicted by a predicting unit 33 and the difference between vectors idr and iqr of a command current is the minimum is determined as the operation state of an inverter IV. At the time of rectangular wave control, the offset value Δ in the evaluation function J for states other than an operation state according to a phase determined by torque feedback control is increased.

Description

  The present invention relates to a DC / AC converter circuit including a switching element that selectively connects a terminal of a rotating machine to each of a positive electrode and a negative electrode of a DC voltage source, by turning on / off the switching element of the DC / AC converter circuit. The present invention relates to a control device for a rotating machine that controls a control amount having at least one of a current flowing through the rotating machine, a torque of the rotating machine, and a magnetic flux of the rotating machine.

  As this type of control device, for example, as seen in Patent Document 1 below, the current of the three-phase motor is predicted for each case where the operation state of the inverter is set variously, and the predicted current and the command current are An apparatus has been proposed that performs so-called model predictive control for operating an inverter so as to achieve an operation state in which a deviation can be minimized. According to this, since the inverter is operated so as to optimize the behavior of the current predicted based on the operation state of the inverter, the followability to the command current at the time of transition can be improved. For this reason, model predictive control is considered to be highly useful for applications that require particularly high performance as transient tracking characteristics, such as a motor generator control device as an in-vehicle main unit.

JP 2008-228419 A

  However, when model predictive control is performed, an operation state that minimizes the deviation between the predicted current and the command current is selected each time, so that the switching frequency of the inverter switching state increases, and consequently, between the output lines of the inverter. It becomes difficult to increase the effective value of the fundamental wave component of the voltage.

  The present invention has been made in the process of solving the above-described problems, and an object thereof is a DC / AC conversion circuit including a switching element that selectively connects a terminal of a rotating machine to each of a positive electrode and a negative electrode of a DC voltage source. A new control for controlling a control amount having at least one of the current flowing through the rotating machine, the torque of the rotating machine, and the magnetic flux of the rotating machine by turning on and off the switching element of the DC / AC converter circuit. It is to provide a control device for a rotating machine.

  Hereinafter, means for solving the above-described problems and the operation and effect thereof will be described.

  According to the first aspect of the present invention, there is provided a DC / AC converter circuit including a switching element that selectively connects a terminal of a rotating machine to each of a positive electrode and a negative electrode of a DC voltage source, and the switching element of the DC / AC converter circuit is turned on / off. In a control device for a rotating machine that controls a control amount having at least one of a current flowing through the rotating machine, a torque of the rotating machine, and a magnetic flux of the rotating machine by operating, a voltage vector in a fixed coordinate system Predicting means for predicting the control amount when the operation state of the DC / AC conversion circuit to be expressed is temporarily set, and an operation state corresponding to the predicted control amount based on the control amount predicted by the prediction means Determining means for determining an operation state having a high evaluation as an operation state of the DC-AC converter circuit, and the DC switching so as to be the determined operation state. An operation means for operating a conversion circuit, and an operation state that increases the evaluation of a specific operation state and increases the evaluation when the determination unit evaluates the operation state in order to make the determination. Variable means for variably setting according to the rotation angle.

  In the above-described invention, since the evaluation of a specific operation state is enhanced by the variable means according to the rotation angle, switching of the switching state can be limited by the design of the variable means.

  According to a second aspect of the present invention, in the first aspect of the invention, when the DC-AC converter circuit is operated according to the operation state that is most highly evaluated by the variable means, each terminal of the rotating machine is connected to the rotating machine. In one electrical angle cycle, the rectangular wave control is connected to the positive electrode and the negative electrode of the DC voltage source once each.

  The invention according to claim 3 is the invention according to claim 1 or 2, wherein the determining means determines an operation state having the highest evaluation as an operation state of the DC-AC converter circuit, and the variable means. Is characterized in that when the difference between the predicted control amount corresponding to the specific operation state and the command value is equal to or less than a specified value, the evaluation of the specific operation state is the highest. .

  In the above invention, when the difference between the controlled variable and its command value exceeds the specified value, it is possible to adopt other than a specific operation state, so that the controlled variable is excessively deviated from the command value by setting the specified value. It becomes possible to suppress the situation.

  According to a fourth aspect of the present invention, in the third aspect of the present invention, the specified value is variably set by inputting at least one of a current flowing through the rotating machine, a torque of the rotating machine, and a rotational speed of the rotating machine. It is characterized by that.

  The absolute value of the controlled variable to be evaluated by the determining means depends on the current and torque. For this reason, the maximum value allowed as the difference between the controlled variable and its command value depends on the magnitude of the absolute value of current and torque. In this regard, if the specified value is variably set according to the current and torque, the specified value can be set to the maximum allowable value. The rotation speed is a parameter having a correlation with the change rate of the control amount. For this reason, if the allowable maximum value of the difference between the controlled variable and its command value is variably set according to the rotational speed, this controlled maximum value and its command value are controlled by the control under normal control conditions. It is a value that does not deviate from the above and can be made as small as possible.

  According to a fifth aspect of the present invention, in the invention according to the third or fourth aspect, the determining means increases the evaluation as the difference between the predicted control amount and its command value is small. The variable means Is obtained by correcting the relative error obtained by subtracting the absolute value of the difference corresponding to the specific operation state from the absolute value of the difference corresponding to the operation state other than the specific operation state by the specified value. It is characterized by being left to evaluation by a determination means.

  In the said invention, when the absolute value of the said difference corresponding to a specific operation state becomes beyond a regulation value, another operation state can be employ | adopted.

  The invention according to claim 6 is the invention according to any one of claims 1 to 5, wherein the phase of the output voltage vector of the DC-AC converter circuit is set as an operation amount for controlling the control amount. A phase setting means is further provided, wherein the variable means increases the evaluation of the operating state of the DC / AC converter circuit determined from the phase set by the phase setting means.

  In the above-described invention, this can be appropriately performed under the situation where the control amount is controlled according to the phase of the voltage.

  According to a seventh aspect of the present invention, in the invention of the sixth aspect, the variable means is provided for each angle region obtained by dividing "1/2" of one electrical angle period of the rotating machine by the number of phases of the rotating machine. The operation state for increasing the evaluation is variably set.

  In the rectangular wave control, the switching state of the switching element connected to each terminal of the rotating machine is switched once to “½” of one electrical angle cycle. For this reason, the interval between the timings at which the switching state is switched at any terminal connected to the DC / AC converter circuit is assumed to be the number of phases of the rotating machine, assuming a normal case where each phase is evenly distributed. The angle region is obtained by dividing "1/2" of one electrical angle cycle of the rotating machine. In the above invention, in view of this point, the variable means is configured.

  The invention according to claim 8 is the invention according to claim 6 or 7, wherein the phase setting means operates the phase as an operation amount for feedback-controlling the torque of the rotating machine to its command value. And

  According to a ninth aspect of the present invention, in the invention according to any one of the first to eighth aspects, when the variable setting is performed by the variable unit, the operation unit sets the update timing of the operation state to the rotating machine. The rotation angle period is set as follows.

  In the above invention, since the update timing is set by the angular period, an error from the control intended by the variable means can be reduced regardless of the rotation speed.

  The invention according to claim 10 is the invention according to any one of claims 6 to 8, wherein when the phase is set by the phase setting means, the operation state is updated as the operation state update timing. The operation state change timing indicated by the set phase is included.

  In the above invention, compared with the case where the update timing is set to a time period, the operation state can be switched accurately according to the phase by the phase setting means.

1 is a system configuration diagram according to a first embodiment. FIG. The figure which shows a voltage vector. The flowchart which shows the process sequence of the model prediction control concerning the said embodiment. The figure which shows the temporary setting candidate of the operation state concerning the embodiment. The figure explaining the problem of an overmodulation area | region. The flowchart which shows the procedure of the switching process of the evaluation object of the overmodulation area | region concerning the said embodiment. The flowchart which shows the procedure of the switching process of the evaluation method in the rectangular wave control concerning the embodiment. The figure which shows the driving | operation method which verified the effect of the embodiment. The time chart which shows the effect of the embodiment. The time chart which shows the effect of the embodiment. The time chart which shows the effect of the embodiment. The time chart which shows the effect of the embodiment. The flowchart which shows the procedure of the process which prescribes | regulates the update period of the model prediction control concerning 2nd Embodiment. The flowchart which shows the procedure of the process which prescribes | regulates the update period of the model prediction control concerning 3rd Embodiment. The flowchart which shows the procedure of the setting process of the command current at the time of the rectangular wave control concerning 4th Embodiment. The flowchart which shows the procedure of the switching process of the evaluation method in the rectangular wave control concerning the embodiment. The system block diagram concerning 5th Embodiment.

<First Embodiment>
Hereinafter, a first embodiment in which a control device for a rotating machine according to the present invention is applied to a control device for a rotating machine as an in-vehicle main machine will be described with reference to the drawings.

  FIG. 1 shows the overall configuration of a motor generator control system according to this embodiment. The motor generator 10 as an in-vehicle main machine is a three-phase permanent magnet synchronous motor. The motor generator 10 is a rotating machine (saliency pole machine) having saliency. Specifically, the motor generator 10 is an embedded magnet synchronous motor (IPMSM).

  Motor generator 10 is connected via inverter IV to high voltage battery 12 having a terminal voltage of, for example, 100 V or more. The inverter IV includes three sets of series connection bodies of switching elements S * p, S * n (* = u, v, w), and the connection points of these series connection bodies are U, V, Each is connected to the W phase. In the present embodiment, insulated gate bipolar transistors (IGBTs) are used as the switching elements S * # (* = u, v, w; # = p, n). In addition, a diode D * # is connected in antiparallel to each of these.

  In this embodiment, the following is provided as detection means for detecting the state of the motor generator 10 and the inverter IV. First, a rotation angle sensor 14 for detecting the rotation angle (electrical angle θ) of the motor generator 10 is provided. Further, a current sensor 16 that detects currents iu, iv, and iw flowing through the phases of the motor generator 10 is provided. Further, a voltage sensor 18 for detecting an input voltage (power supply voltage VDC) of the inverter IV is provided.

  The detection values of the various sensors are taken into the control device 20 constituting the low voltage system via an interface (not shown). The control device 20 generates and outputs an operation signal for operating the inverter IV based on the detection values of these various sensors. Here, the signal for operating the switching element S * # of the inverter IV is the operation signal g * #.

  The control device 20 operates the inverter IV to control the torque of the motor generator 10 to the required torque Tr. Specifically, inverter IV is operated so that the command current for realizing required torque Tr matches the current flowing through motor generator 10. That is, in the present embodiment, the torque of the motor generator 10 becomes the final control amount. In order to control the torque, the current flowing through the motor generator 10 is directly controlled as a control current. To do. In particular, in the present embodiment, in order to control the current flowing through the motor generator 10 to the command current, the current of the motor generator 10 is predicted when the operation state of the inverter IV is set to each of a plurality of ways, and Among these, model predictive control is performed in which the predicted current close to the command current is adopted as the actual operation state of the inverter IV.

In the following, “1. Control at a low modulation rate” will be described first, followed by “2. Control at an overmodulation region”, and finally “3. To do.
“1. Control at low modulation rate”
The phase currents (currents iu, iv, iw) detected by the current sensor 16 are converted into actual currents id, iq in the rotating coordinate system by the dq converter 22. Further, the electrical angle θ detected by the rotation angle sensor 14 is input to the speed calculation unit 23, and thereby the rotation speed (electrical angular speed ω) is calculated. On the other hand, the command current setting unit 24 receives the required torque Tr and outputs command currents idr and iqr in the dq coordinate system. The command currents idr and iqr, the actual currents id and iq, and the electrical angle θ are input to the model prediction control unit 30. The model prediction control unit 30 determines a voltage vector Vi that defines the operation state of the inverter IV based on these input parameters, and outputs the voltage vector Vi to the operation unit 26. The operation unit 26 generates the operation signal based on the input voltage vector Vi and outputs it to the inverter IV.

  Here, the voltage vectors expressing the operation state of the inverter IV are eight voltage vectors shown in FIG. For example, a voltage vector representing an operation state (indicated as “lower” in the drawing) in which the low-potential side switching elements Sun, Svn, Swn are turned on is the voltage vector V0, and the high-potential side switching elements Sup, A voltage vector representing an operation state (indicated as “upper” in the drawing) in which Svp and Swp are turned on is a voltage vector V7. These voltage vectors V0 and V7 are for short-circuiting all phases of the motor generator 10 and are called zero voltage vectors because the voltage applied to the motor generator 10 from the inverter IV becomes zero. On the other hand, the remaining six voltage vectors V1 to V6 are defined by an operation pattern in which switching elements that are turned on exist in both the upper arm and the lower arm, and are called effective voltage vectors. Yes. As shown in FIG. 2B, each of the voltage vectors V1, V3, and V5 corresponds to the positive side of the U phase, the V phase, and the W phase, respectively.

  Next, details of the processing of the model prediction control unit 30 will be described. In the operation state setting unit 31 shown in FIG. 1, the operation state of the inverter IV is set. Here, voltage vectors V0 to V7 shown in FIG. 2 are set as the operation state of inverter IV. The dq conversion unit 32 calculates the voltage vector Vdq = (vd, vq) in the dq coordinate system by performing dq conversion on the voltage vector set by the operation state setting unit 31. In order to perform such conversion, the voltage vectors V0 to V7 in the operation state setting unit 31 are, for example, “upper” as “VDC / 2” and “lower” as “−VDC / 2” in FIG. It can be expressed as above. In this case, for example, the voltage vector V0 is (−VDC / 2, −VDC / 2, −VDC / 2), and the voltage vector V1 is (VDC / 2, −VDC / 2, −VDC / 2). .

In the prediction unit 33, the current id when the operation state of the inverter IV is set to the state set by the operation state setting unit 31 based on the voltage vector (vd, vq), the actual currents id, iq, and the electrical angular velocity ω. , Iq is predicted. Here, the voltage equation expressed by the following (c1) and (c2) is discretized from the following state equations (formulas (c3) and (c4)) obtained by solving the current differential term. Predict.
vd = (R + pLd) id−ωLqiq (c1)
vq = ωLdid + (R + pLq) iq + ωφ (c2)
pid
= − (R / Ld) id + ω (Lq / Ld) iq + vd / Ld (c3)
piq
= −ω (Ld / Lq) id− (Rd / Lq) iq + vq / Lq−ωφ / Lq (c4)
Incidentally, in the above formulas (c1) and (c2), the resistance R, the differential operator p, the d-axis inductance Ld, the q-axis inductance Lq, and the armature flux linkage constant φ are used.

  The prediction of the current is performed for each of a plurality of operation states set by the operation state setting unit 31.

  On the other hand, the operation state determination unit 34 receives the predicted currents ide and iq predicted by the prediction unit 33 and the command currents idr and iqr, and determines the operation state of the inverter IV. Here, each operation state set by the operation state setting unit 31 is evaluated by the evaluation function J, and the operation state with the highest evaluation is selected. As this evaluation function J, in the present embodiment, a function whose value increases as the evaluation becomes lower is adopted. Specifically, the evaluation function J is calculated based on the inner product value of the difference between the command current vector Idqr = (idr, iqr) and the predicted current vector Idqe = (ide, iqe). This is a technique for expressing that the evaluation is lower as the value is larger in view of the fact that the deviation of each component between the command current vector Idqr and the predicted current vector Idqe can be both positive and negative values. As a result, it is possible to construct an evaluation function J in which the evaluation becomes lower as the difference between the components of the command current vector Idqr and the predicted current vector Idqe is larger. In the present embodiment, the evaluation function J is the sum of the inner product value and the offset value Δ. The offset value Δ will be described in detail later.

  FIG. 3 shows a model prediction control processing procedure according to the present embodiment. This process is repeatedly executed at a predetermined cycle (control cycle Tc).

  In this series of processing, first, in step S10, the electrical angle θ (n) and the actual currents id (n) and iq (n) are detected, and the voltage vector V (n) determined in the previous control cycle. Is output. In subsequent step S12, the current (ide (n + 1), iqe (n + 1)) in one control cycle ahead is predicted. This is a process of predicting what will happen to the current one control cycle ahead based on the voltage vector V (n) output in step S10. Here, the predicted currents ide (n + 1) and iqe (n + 1) are calculated by using the model expressed by the above equations (c3) and (c4) discretized by the forward difference method with the control cycle Tc. To do. At this time, the actual currents id (n) and iq (n) detected in step S10 are used as the initial value of the current, and the voltage vector V (n) is used as the voltage vector on the dq axis. And dq-transformed by using an angle obtained by adding “ωTc / 2” to the electrical angle θ (n) detected in step (b).

  In subsequent steps S14 and S16, a process of predicting a current two control cycles ahead is performed for each of cases where a plurality of voltage vectors are set in the next control cycle. That is, first, in step S14, the voltage vector V (n + 1) in the next control cycle is temporarily set. Here, the voltage vector V (n + 1) in the next control cycle is defined as the voltage vector V (n + 1) in the next control cycle in which the number of switching phases from the operation state represented by the voltage vector V (n) in the current control cycle is equal to or less than “1”. Set temporarily.

  For example, when the voltage vector V (n) is an effective voltage vector (Vi; i = 1 to 6), the voltage vector V (n + 1) is set to the voltage vectors Vi−1, Vi, Vi + 1 (i: mod 6). Or a zero voltage vector. However, as the zero voltage vector, if V (n) = V2k (k = 1 to 3), the voltage vector V7 is selected, and if V (n) = V2k−1, the voltage vector V0 is selected. . FIG. 4A shows four voltage vectors that can be temporarily set as the voltage vector V (n + 1) when V (n) = V1. Further, when the current voltage vector V (n) is a zero voltage vector (voltage vector V0), as shown in FIG. 4B, the voltage vector V (n + 1) is changed to odd voltage vectors V1, V3, V5. Alternatively, the voltage vector V0. Further, when the current voltage vector V (n) is a zero voltage vector (voltage vector V7), as shown in FIG. 4C, the voltage vector V (n + 1) is converted into an even voltage vector V2, V4, V6. Alternatively, the voltage vector V7 is used.

  In subsequent step S16, predicted currents ide (n + 2) and iqe (n + 2) are calculated in the same manner as in step S12. However, here, the predicted currents ide (n + 1) and iqe (n + 1) calculated in step S12 are used as the initial value of the current, and the voltage vector V (n + 1) is used as the voltage vector on the dq axis. What dq-transformed by the angle which added "3 (omega) Tc / 2" to the electrical angle (theta) (n) detected in step S10 is used.

  In the subsequent step S18, a process for determining the voltage vector V (n + 1) in the next control cycle is performed. Here, four values of the evaluation function J are calculated using the predicted currents ide (n + 2) and iqe (n + 2) corresponding to the four voltage vectors temporarily set in step S14, and among these, the evaluation function J Let the voltage vector that minimizes J be the final voltage vector V (n + 1). In the subsequent step S20, the voltage vectors V (n) and V (n + 1) are set as voltage vectors V (n-1) and V (n), respectively, and the electrical angle θ (n) is the electrical angle θ (n-1). And real currents id (n) and iq (n) are assumed to be real currents id (n-1) and iq (n-1), respectively.

In addition, when the process of step S20 is completed, this series of processes is once complete | finished.
“2. Control in the overmodulation region”
By the way, by inputting the command currents idr and iqr (fixed values) into the above formulas (c1) and (c2), the norm of the command voltage vector (vdr, vqr) is uniquely determined. This norm rotates “360 °” in the two-dimensional fixed coordinate system shown in FIG. 5A when the electrical angle θ of the motor generator 10 rotates “360 °”. Here, as shown in FIG. 5A, the output voltage that can be realized by the inverter IV is limited to a hexagonal region in which the length of one side is “√ (2/3) × VDC”. For this reason, the upper limit of the curve (circle) drawn by the end point of the command voltage vector (vdr, vqr) is the circle (hexagonal inscribed circle) with the radius “VDC / √2” shown in FIG. Become. This state means that the amplitude of the fundamental wave component of the output line voltage of the inverter IV coincides with the input voltage, and the modulation rate at this time is defined as “2 / √3”.

  In the present embodiment, an area where the modulation rate is larger than “2 / √3” is defined as an overmodulation area. It is well known that the modulation rate can be further increased in the overmodulation region. However, in the overmodulation region, that is, when the norm of the command voltage vector corresponding to the command currents idr and iqr is larger than the radius of the circle, the command currents idr and iqr may be changed depending on the electrical angle of the motor generator 10. This cannot be realized, and as a result, a harmonic component of six times the electrical angular frequency is superimposed on the control amount of the motor generator 10. That is, in the case of the command voltage vector represented by the circle shown in FIG. 5B, the voltage is output as commanded at the broken line portion where the end point of the command voltage vector falls within the hexagonal region, and the hexagonal region A voltage constrained to a hexagon is output from the one-dot chain line portion that protrudes, and this is repeated at a cycle that is six times the electrical angular frequency. For this reason, a harmonic of 6 times the electrical angular frequency is superimposed on the control amount of the motor generator 10.

  Even if the current fluctuates due to the superposition of the sixth harmonic, if the average value becomes the command currents idr and iqr, the torque of the motor generator 10 can be controlled to the required torque Tr. However, when the operation state of the inverter IV is determined using the evaluation function J, a steady divergence occurs between the actual torque of the motor generator 10 and the required torque Tr. This is due to the fact that the time interval (predicted interval) between the predicted timing of the predicted currents ide (n + 2) and iqe (n + 2) and the current time (predicted interval) is smaller than the period of the harmonics. . That is, in this case, the operation state for minimizing the difference between the predicted currents ide and iq and the command currents idr and iqr in the local time case is selected. However, there is a region where the actual current cannot be used as the command current due to restrictions from the output voltage of the inverter IV and a region where control exceeding this is possible. In an area where control is possible to exceed, since an operation state that minimizes the difference between the command current and the predicted current is selected, use of the operation state exceeding the command current is avoided, and the average value of the current is Lacks against.

  Incidentally, the steady-state deviation can be eliminated if the next operation state is determined by evaluating the difference between the predicted currents ide and iq and the command currents idr and iqr each time over a long period of time equal to or higher than the period of the harmonics. However, in this case, the calculation load becomes excessive.

  Therefore, in this embodiment, the steady deviation in the overmodulation region is suppressed by the process shown in FIG.

  That is, the actual current id output from the dq converter 22 is taken into a low-pass filter 40 such as a first-order lag filter. The low-pass filter 40 selectively transmits the fundamental wave component while attenuating the harmonic component of the actual current id, and variably sets the cutoff frequency according to the electrical angular velocity ω. The switching unit 42 switches whether to output the output signal (average actual current idL) of the low-pass filter 40 to the prediction unit 33. On the other hand, the actual current iq output from the dq converter 22 is taken into a low-pass filter 44 such as a first-order lag filter. The low-pass filter 44 selectively transmits the fundamental wave component while attenuating the harmonic component of the actual current iq, and variably sets the cutoff frequency according to the electrical angular velocity ω. The switching unit 46 switches whether to output the output signal (average actual current iqL) of the low-pass filter 44 to the prediction unit 33.

  Thus, when the average actual currents idL and iqL are input to the prediction unit 33 by the switching units 42 and 46, the prediction unit 33 calculates the predicted currents ide and iq based on the actual currents id and iq, The average predicted currents ideL and iqL are calculated based on the average actual currents idL and iqL.

  Then, a value obtained by multiplying the average predicted current ideL predicted by the prediction unit 33 based on the average actual currents idL and iqL by the weighting coefficient α (> 0) and the prediction predicted by the prediction unit 33 based on the actual currents id and iq. A value obtained by multiplying the current ide by a weighting factor “1-α (> 0)” is added by the adder 48. Thereby, the weighted average processing of the average predicted current ideL and the predicted current ide is performed. Then, the selector 52 selectively outputs either the value of the addition unit 48 or the predicted current ide to the operation state determination unit 34.

  In addition, a value obtained by multiplying the average predicted current iqeL predicted by the prediction unit 33 based on the average actual currents idL and iqL by the weighting coefficient α (> 0) and a prediction predicted by the prediction unit 33 based on the actual currents id and iq. The addition unit 50 adds the value obtained by multiplying the current iq by the weight coefficient “1−α (> 0)”. Thereby, the weighted average processing of the average predicted current iqL and the predicted current iq is performed. Then, the selector 54 selectively outputs either the value of the adding unit 50 or the predicted current iq to the operation state determining unit 34.

  Here, the outputs of the adders 48 and 50 are those in which the phase delay of the fundamental wave component is reduced while the harmonic components of the predicted currents ide and iq predicted by the prediction unit 33 are removed. This is because it includes components that are not filtered by the low-pass filters 40 and 44. Here, the effect of the phase lag compensation increases as the weight coefficient α approaches zero.

  FIG. 6 shows a procedure for switching the model prediction method by operating the switching units 42 and 46 and the selectors 52 and 54. This process is repeatedly executed in the control device 20, for example, at a predetermined cycle.

  In this series of processing, first, in step S30, an average voltage vector Va is calculated. This is performed by inputting the command current vector Idqr to the above equations (c1) and (c2) from which the differential operator p is removed. Here, the average voltage vector Va is an effective value of a fundamental wave component having an electrical angular frequency in the output voltage of the inverter IV. In other words, the inverter IV switches the switching state at a time interval shorter than one electrical angular cycle, so that the output voltage simulates a sinusoidal voltage having an electrical angular frequency component. The sinusoidal voltage simulated by the inverter IV is the average voltage vector Va. Incidentally, the norm of the average voltage vector Va is a physical quantity proportional to the modulation rate and the voltage utilization rate. Here, the modulation rate is a Fourier coefficient of the fundamental wave component for the output voltage of the inverter IV. In calculating the Fourier coefficient, the amplitude center of the fundamental wave and the median value of the fluctuation range of the output voltage of the inverter IV are matched.

  In the subsequent step S32, the modulation factor M is calculated. This is “(| Va | / VDC) √ (8/3)” using the power supply voltage VDC and the average voltage vector Va. In a succeeding step S34, it is determined whether or not the region is an overmodulation region. If an affirmative determination is made in step S34, the operation state of the inverter IV is evaluated in step S36 using the average predicted currents ideL and iqeL (based on the output signals of the adders 48 and 50). On the other hand, when a negative determination is made in step S36, the operation state of the inverter IV is evaluated using only the predicted currents ide and iqe in step S38.

When the processes in steps S36 and S38 are completed, this series of processes is temporarily terminated.
"3. Rectangular wave control"
As is well known, the control that can maximize the modulation factor of the inverter IV is rectangular wave control. However, the inventors have found that when the control is performed in the above-described modulation region, the rectangular wave control is not performed due to the evaluation of the voltage vector using the evaluation function J. Therefore, in the present embodiment, in the rectangular wave control, the process of switching the evaluation method is performed by operating the offset value Δ that defines the evaluation function J.

  FIG. 7 shows the procedure of the switching process of the evaluation method according to the present embodiment. This process is repeatedly executed, for example, at a predetermined cycle.

  In this series of processing, first, in step S40, it is determined whether or not a rectangular wave flag indicating that rectangular wave control is being performed is ON. If a negative determination is made in step S40, it is determined in step S42 whether or not the modulation factor M is greater than the specified value Mth. Here, the prescribed value Mth is set to about the upper limit value of the modulation rate that can be realized by performing the evaluation using the average predicted currents ideL and iqeL. Since this upper limit value tends to vary according to the operating point determined by the torque and the electrical angular velocity ω, it is desirable to variably set the specified value Mth according to the operating point.

  If a positive determination is made in step S42, the rectangular wave flag is turned on in step S44. In the following step S52, the phase command θv of the output voltage of the inverter IV is calculated as an operation amount for feedback-controlling the estimated torque estimated from the actual currents id and iq to the required torque Tr. Here, the phase command θv may be the sum of the phase δ with respect to the d axis and the electrical angle θ. Further, the phase δ with respect to the d axis may be calculated as the sum of the proportional element and the integral element of the difference between the estimated torque and the required torque Tr. The estimated torque may be predicted by the following equation (c5).

T = P {Φ · iq + (Ld−Lq) id · iq} (c5)
Incidentally, the number P of pole pairs is used in the above formula (c5).

  In the subsequent step S54, among the offset values Δ corresponding to the respective operation states, those corresponding to the voltage vector having the smallest angle with the vector Vθv in the phase command θv direction are set to zero, and other values are set to the specified values. Let A (> 0). Here, the specified value A is set to be larger than the maximum value of values that can be assumed as the inner product value of the difference between the command current vector Idqr = (idr, iqr) and the predicted current vector Idqe = (ide, iqe). To do. Thereby, it is possible to always select the voltage vector having the smallest angle with the vector Vθv.

  On the other hand, if an affirmative determination is made in step S40, it is determined in step S46 whether the difference between the phase of the vector of the predicted currents ide and iqe and the phase of the vector of the command currents idr and iqr is equal to or less than a predetermined value. . This determination can be made by setting a predetermined range according to the phase of the vectors of the command currents idr and iqr. If a negative determination is made in step S46, the process proceeds to step S52. If an affirmative determination is made in step S46, the rectangular wave flag is turned off in step S48. When the process of step S48 is completed or when a negative determination is made in step S42, the offset value Δ is set to zero in step S50.

  In addition, when the process of step S50, S54 is completed, this series of processes is once complete | finished.

  The effect of this embodiment is shown in FIGS. Here, FIG. 8 shows an operation region for evaluating the effect of the present embodiment. Case 1 corresponds to FIG. 9, case 2 corresponds to FIG. 10, and case 3 corresponds to FIG. .

  Specifically, as case 1, as shown in FIG. 9 (b), from a sine wave mode evaluated using only the predicted currents ide and iqe, to an overmodulation mode evaluated using the average predicted currents ideL and iqL. The case where the rectangular wave control is performed has been shown. On the other hand, FIG. 9A shows a case where the rectangular wave control is continued. Further, as the case 2, as shown in FIG. 10A, when the rectangular wave control is performed from the sine wave mode through the overmodulation mode, and as shown in FIG. 10B, from the rectangular wave control to the overmodulation mode. The case where the mode is changed to the sine wave mode is shown. In addition, as the case 3, the case where the rectangular wave control is continued for each of the case where the torque is increased (FIG. 11A) and the case where the torque is decreased (FIG. 11B) is shown.

  12A shows a case where the processing shown in FIG. 7 is performed, and FIG. 12B shows a case where the offset value Δ is always zero. As shown in the figure, the number of switchings can be reduced by manipulating the offset value Δ by the process shown in FIG. 7, and the switching cycle of the switching state of each phase is set to “1/2 of one electrical angle cycle. ".

  According to the embodiment described in detail above, the following effects can be obtained.

(1) The evaluation of the operation state of the inverter IV expressed by the voltage vector having the smallest phase difference from the vector Vθv in the phase command θv direction is the highest. Thereby, the phase of the output voltage of the inverter IV in the rectangular wave control can be manipulated as an operation amount of the torque feedback control.
<Second Embodiment>
Hereinafter, the second embodiment will be described with reference to the drawings with a focus on differences from the first embodiment.

  In the present embodiment, the rotation angle period is set by variably setting the time corresponding to the operation state update period (control period Tc) by the model predictive control according to the electrical angular velocity ω.

  FIG. 13 shows a procedure of update cycle switching processing according to the present embodiment. This process is repeatedly executed, for example, at a predetermined cycle.

  In this series of processing, it is first determined in step S60 whether or not the rectangular wave flag is on. If an affirmative determination is made in step S60, the control cycle Tc is set to “2π / N” in step S62. When the process of step S62 is completed or when a negative determination is made in step S60, this series of processes is temporarily terminated.

  According to the above processing, in the rectangular wave control, the operation state of the inverter IV is updated every “2π / N”. Therefore, compared with the case where the control cycle Tc is determined by a fixed time, the actual switching phase of the switching state and the switching state indicated by the phase command θv under the situation where the phase of the output voltage of the inverter IV is manipulated. It is possible to reduce fluctuations in error from the switching phase.

  According to the present embodiment described above, the following effects can be obtained in addition to the effect (1) of the first embodiment.

(2) The update timing of the operation state of the inverter IV is set by the rotation angle cycle of the motor generator 10. Thereby, it is possible to reduce the fluctuation of the phase error of the output voltage of the inverter IV due to the electrical angular velocity ω.
<Third Embodiment>
Hereinafter, the third embodiment will be described with reference to the drawings with a focus on differences from the first embodiment.

  In the present embodiment, interrupt processing is performed in which the timing at which the angle between the voltage vectors V1 to V6 and the vector Vθv in the direction of the phase command θv becomes the minimum is changed to the operation state update timing by model predictive control.

  FIG. 14 shows the procedure of the update cycle switching process according to this embodiment. This process is repeatedly executed, for example, at a predetermined cycle.

  In this series of processing, it is first determined in step S70 whether or not the rectangular wave flag is on. If ON in step S70, it is determined in step S72 whether or not the voltage vector that minimizes the angle formed by the vector Vθv in the phase command θv direction has changed. If an affirmative determination is made in step S72, the operation state of inverter IV is updated (outputs voltage vector V (n + 1)) in step S74. When the process of step S74 is completed or when a negative determination is made in steps S70 and S72, this series of processes is temporarily ended.

  According to the above processing, the error between the phase of the output voltage of the inverter IV and the phase command θv can be sufficiently reduced, and more accurate phase operation can be performed.

  According to the present embodiment described above, the following effects can be obtained in addition to the effect (1) of the first embodiment.

(3) The timing at which switching of the switching state is instructed by the phase command θv is included in the update timing of the operation state of the inverter IV. Thereby, since the precision of the phase as an operation amount can be improved, the controllability of torque feedback control can be improved.
<Fourth Embodiment>
Hereinafter, the fourth embodiment will be described with reference to the drawings with a focus on differences from the first embodiment.

  In the present embodiment, when the rectangular wave control is executed, if the difference between the predicted currents ide (n + 2) and iqe (n + 2) and the command currents idr and iqr exceeds the allowable range, the command currents idr and iqr and the predicted current ide , Iqe, the operation state with the smallest difference is selected. For this reason, in this embodiment, the command currents idr and iqr for rectangular wave control are set. That is, in the rectangular wave control, field-weakening control is performed according to the operation of the phase of the output voltage of the inverter IV by torque feedback control. The command currents idr and iqr at this time are different from the output of the command current setting unit 24 set to realize, for example, minimum current / maximum torque control. For this reason, in order to determine whether or not the current flowing during the rectangular wave control is normal, a command current for rectangular wave control is required. Therefore, in the present embodiment, during the rectangular wave control, the command currents idr and iqr are set by the process shown in FIG.

  FIG. 15 shows a procedure for setting a command current during rectangular wave control. This process is repeatedly executed, for example, at a predetermined cycle.

  In this series of processing, first, in step S80, it is determined whether or not the rectangular wave flag is on. When an affirmative determination is made in step S80, a magnetic flux norm command value Φfwc for field weakening control is calculated by the following equation (c6) in step S82.

ここでは、ノルム目標値Ｎｒ、ｄ軸正方向と鎖交磁束ベクトルとのなす角度θｆ（＝ａｒｃｔａｎ（Ｌｑｉｑ／（Ｌｄｉｄ＋φ）））を用いている。

Here, the norm target value Nr, the angle θf (= arctan (Lqiq / (Ldid + φ))) formed by the d-axis positive direction and the flux linkage vector are used.

  Hereinafter, derivation of this equation will be described.

When the induced voltage Vo and the current I are used, the terminal voltage Vam of the motor generator 10 is expressed by the following formula (c7) when the influence due to the fluctuation of the current is ignored.
Vam = Vo + RI (c7)
The above equation (c7) becomes the following equation (c8) using an angle θf (= arctan (Lqiq / (Ldid + φ))) formed by the d-axis positive direction and the flux linkage vector.

ただし、上記の式（ｃ８）において、モータジェネレータ１０の正回転がプラス符号に対応し、逆回転がマイナス符号に対応する。

However, in the above formula (c8), the forward rotation of the motor generator 10 corresponds to a plus sign and the reverse rotation corresponds to a minus sign.

  When the above equation (c8) is solved for the induced voltage Vo, the following equation (c9) is obtained.

磁束ノルム指令値Φｆｗｃは、「｜ω｜・Φｆｗｃ＝Ｖｏ」に基づき上記の式（ｃ９）を変形することで、上記の式（ｃ６）にて表現される。ただし、上記の式（ｃ６）においては、端子電圧Ｖａｍを、ノルム目標値Ｎｒに代えている。ここで、ノルム目標値Ｎｒは、電源電圧ＶＤＣに、「Ｍｒ・√（３／８）」を乗算したものである。ここでは、変調率指令値Ｍｒを用いている。ここで、「Ｍｒ・√（３／８）」は、電圧利用率指令値である。電圧利用率は、変調率と同様、インバータＩＶの出力電圧ベクトルの大きさを定量化した物理量である。なお、変調率指令値Ｍｒを、本実施形態では、矩形波制御の変調率である「１．２７」とする。

The magnetic flux norm command value Φfwc is expressed by the above formula (c6) by modifying the above formula (c9) based on “| ω | · Φfwc = Vo”. However, in the above formula (c6), the terminal voltage Vam is replaced with the norm target value Nr. Here, the norm target value Nr is obtained by multiplying the power supply voltage VDC by “Mr · √ (3/8)”. Here, the modulation factor command value Mr is used. Here, “Mr · √ (3/8)” is a voltage utilization rate command value. The voltage utilization rate is a physical quantity obtained by quantifying the magnitude of the output voltage vector of the inverter IV, like the modulation rate. In this embodiment, the modulation factor command value Mr is set to “1.27”, which is a modulation factor of rectangular wave control.

  In the subsequent step S84, the command currents idr and iqr are map-calculated based on the magnetic flux norm command value Φfwc and the required torque Tr. As a result, the command currents idr and iqr are currents required for the required torque Tr when the modulation rate is the same as the modulation rate of the rectangular wave control.

  When the process of step S84 is completed or when a negative determination is made in step S80, this series of processes is temporarily terminated.

  FIG. 16 shows a procedure of the evaluation method switching process according to the embodiment. This process is repeatedly executed, for example, at a predetermined time cycle (control cycle Tc). In FIG. 16, processes corresponding to the processes shown in FIG. 7 are given the same step numbers for convenience.

  In this series of processes, when an affirmative determination is made in step S42, the rectangular wave flag is turned on in step S44a, and the instantaneous value evaluation that calculates the evaluation function J only by the predicted currents ide (n + 2) and iqe (n + 2) is performed. Switch to. This is a setting for evaluating whether or not the difference between the command currents idr, iqr and the predicted currents ide, iqe is out of the allowable range by an instantaneous value.

  After the process of subsequent step S52, in step S54a, the specified value A when the offset value Δ is set to a value other than zero is variably set according to the required torque Tr and the electrical angular velocity ω. This specified value A is set to an allowable upper limit value for the difference between the predicted currents ide (n + 2) and iqe (n + 2) and the command currents idr and iqr. Here, the required torque Tr is a parameter having a correlation with the absolute value of the current. Therefore, even if the allowable upper limit is a fixed value (error ratio) obtained by dividing the above difference by the vector norm of the command current, the specified value A is variably set according to the required torque Tr. It would be desirable to The electrical angular velocity ω is a parameter having a correlation with the current change rate. For this reason, by variably setting the prescribed value A according to the electrical angular velocity ω, it is possible to appropriately set a value at which the difference does not exceed the prescribed value A during normal operation. The specified value A is equal to or greater than the maximum value at the normal time for the difference between the predicted currents ide (n + 2) and iqe (n + 2) and the command currents idr and iqr. Since this maximum value changes according to the operating point determined according to the torque and the electrical angular velocity ω, it is desirable to variably set the specified value A according to the required torque Tr and the electrical angular velocity ω also from this viewpoint.

  According to the present embodiment described above, the following effects can be obtained in addition to the effect (1) of the first embodiment.

(4) The operation state determined by the phase command θv is determined on condition that the difference between the predicted currents ide (n + 2) and iqe (n + 2) and the command currents idr and iqr is equal to or less than the specified value A. Thereby, it is possible to avoid a situation in which the current flowing through motor generator 10 greatly deviates from command currents idr and iqr.
<Fifth Embodiment>
Hereinafter, a fifth embodiment will be described with reference to the drawings, focusing on differences from the first embodiment.

  FIG. 17 shows a system configuration according to the present embodiment. Note that, in FIG. 17, processes corresponding to the processes shown in FIG.

  As shown in the figure, the torque / magnetic flux predicting unit 64 predicts the magnetic flux vector Φ and the torque T of the motor generator 10 based on the predicted currents ide and iqe. Here, the magnetic flux vector Φ = (Φd, Φq) is predicted by the following equations (c10) and (c11), and the torque is predicted by the following equation (c12).

Φd = Ld · id + φ (c10)
Φq = Lq · iq (c11)
T = P (Φd · iq−Φq · id) (c12)
On the other hand, in the magnetic flux map 24a, the command magnetic flux vector Φr is set based on the required torque Tr. Here, the command magnetic flux vector Φr is set according to a request for realizing, for example, maximum torque control that can obtain the maximum torque with the minimum current among those satisfying the required torque Tr.

  The operation state determination unit 34a determines a final operation state based on the evaluation function J. Here, the evaluation function J is quantified based on the difference between the predicted torque Te and the required torque Tr and the difference between the components of the predicted magnetic flux vector Φe and the command magnetic flux vector Φr. Specifically, it is determined based on the sum of values obtained by multiplying the squares of these differences by weighting factors a and b, respectively. Here, the weighting factors a and b are based on the fact that the magnitudes of torque and magnetic flux are different. That is, for example, when setting a unit in which the numerical value of the torque is larger, the torque deviation tends to be larger. Therefore, when the weighting factors a and b are not used, the voltage vector has low controllability of magnetic flux. However, there is a possibility that disadvantages may occur, such as the evaluation is not so low. For this reason, the weighting factors a and b are used as means for compensating for the difference in absolute value of the plurality of input parameters of the evaluation function J. Considering that the relational expression between torque and magnetic flux is “T = P × Ia × Φ × sin (θv−θi); Ia: current amplitude, θi: current phase angle”, “a = 1, b = | P × Ia × sin (θv−θi) | ”.

  Here, in the present embodiment, in the overmodulation region, the input parameter of the evaluation function J is set to the predicted torque Te predicted by the magnetic flux predicting unit 64 and the predicted magnetic fluxes Φde and Φqe, respectively. Switch to the one filtered by 70. In a region where the modulation factor is small, the input parameter of the evaluation function J is switched between the predicted torque Te predicted by the magnetic flux predicting unit 64 and the predicted magnetic fluxes Φde and Φqe by operating the selectors 72, 72, and 76.

The rectangular wave control can be performed by manipulating the offset value Δ in the evaluation function J.
<Other embodiments>
Each of the above embodiments may be modified as follows.

“Variable setting parameter of offset value Δ”
The torque is not limited to the required torque Tr and the electrical angular velocity ω. For example, an estimated torque or a current flowing through the motor generator 10 may be used as a parameter having a correlation with the torque. Further, in order to grasp the change rate of the current more accurately, it is effective to use the power supply voltage VDC in addition to the electrical angular velocity ω.

  Furthermore, if the variable means is applied at a modulation rate smaller than that of the rectangular wave control, it is effective to variably set according to the modulation rate. Here, the modulation factor is a parameter having a correlation with the magnitude of the sixth-order fluctuation amount of the control amount (torque, current, etc.). Even in this case, when the inverter IV is operated in accordance with the operation state that is the highest evaluation by the variable means, the variable means is a means for operating the phase of the output voltage of the inverter IV.

"Variable means"
What is not limited to the case where the rectangular wave control is performed is as described in the column “variable setting parameter of offset value Δ”.

"Phase setting method"
The operation amount of torque feedback control is not limited to the one that sets the phase. For example, the voltage calculated by substituting the command currents idr and iqr at the time of the rectangular wave control in the fourth embodiment into those obtained by deleting the differential operator p in the above formulas (c1) and (c2). It may be a phase. In this case, the phase is manipulated as an open loop manipulated variable of torque.

“Temporarily set operation status”
The number of switching phases in the switching state is not limited to “1” or less, and may be “2” or less. Moreover, all of voltage vectors V0-V7 may be sufficient.

About prediction means
It is not limited to predicting only the control amount generated by the next voltage vector V (n + 1). For example, the control amount by the operation of the inverter IV at the update timing after several control cycles may be predicted sequentially.

About the decision means
For example, in the first embodiment, the weighted average processing value of the difference between the predicted current ide (n + 2) and the command current idr (n + 2) and the difference between the predicted current iqe (n + 2) and the command current iqr (n + 2). May be used as a parameter for evaluation of the degree of deviation. In short, in order to quantify that the evaluation becomes lower as the degree of divergence is larger, it may be quantified by a parameter having a positive or negative correlation between the degree of divergence and the evaluation.

  For example, if the difference between a controlled variable and its command value is quantified by dividing the difference between the controlled variable and its command value by the command value, the same magnitude and the same regardless of the absolute value of the controlled variable Error (ratio of error to command value) can be quantified. Therefore, in this case, even if the offset value Δ is set to a fixed value, it is possible to obtain an effect according to the fourth embodiment.

"About controlled variables"
The control amount used for determining the operation of the inverter IV based on the command value and the predicted value is not limited to any of torque, magnetic flux, and current. For example, only torque or magnetic flux may be used. For example, torque and current may be used. Here, when the control amount is other than the current, the direct detection target by the sensor may be other than the current.

  In each of the above embodiments, the ultimate control amount of the rotating machine (the control amount that is ultimately required to be a desired amount regardless of whether or not it is a prediction target) is the torque. For example, the rotational speed may be used.

"Overmodulation control"
As means for reducing the harmonic component of the controlled variable from the difference between the controlled variable as the input parameter of the determining means and its command value, low-pass is applied to the actual currents id and iq which are initial values used for calculating the predicted currents ide and iq It is not restricted to what performs a filter process. For example, it may be a means for subtracting a high frequency component extracted by high-pass filtering the actual currents id and iq from the difference between the control amount and its command value.

  Further, the present invention is not limited to the one provided with means for reducing the harmonic component of the control amount from the difference between the control amount as the input parameter of the determining means and its command value. For example, as described in Japanese Patent Application Laid-Open No. 2011-019319, a proportional element that inputs a difference between a controlled variable and its command value and an integral element that receives a difference between the controlled variable and its command value The evaluation function may be configured by adding the absolute values thereof.

"About rotating machines"
It is not limited to a three-phase rotating machine. For example, it may be a four-phase or more rotating machine such as a five-phase rotating machine in which the windings of the five terminals are connected to each other. Even in such a case, the priority area of the voltage vector corresponding to each stator should be an area of “π / N: N is the number of phases” on the advance side and the retard side of the voltage vector, respectively. Is desirable.
Further, in the above-described embodiment, it is assumed that the stator windings are star-connected. However, the present invention is not limited to this and may be, for example, delta-connected. In this case, although the voltage applied to the terminal of the rotating machine is different from the voltage applied to the phase, the priority area of the voltage vector corresponding to each stator is set to the advance side and the retard side of the voltage vector. In each case, it is desirable to set an angle region of “π / N: N is the number of phases”.

  The rotating machine is not limited to an embedded magnet synchronous machine, and may be an arbitrary synchronous machine such as a surface magnet synchronous machine or a field winding type synchronous machine. Furthermore, it is not limited to a synchronous machine, but may be an induction rotating machine such as an induction motor.

  The rotating machine is not limited to that mounted on a hybrid vehicle, but may be mounted on an electric vehicle. Further, the rotating machine is not limited to that used as the main machine of the vehicle.

"others"
For example, in the model predictive control, the conventional rectangular wave control may be performed when the modulation rate is equal to or higher than a specified value.

  The DC voltage source is not limited to the high voltage battery 12 and may be, for example, an output terminal of a converter that boosts the voltage of the high voltage battery 12.

  DESCRIPTION OF SYMBOLS 10 ... Motor generator, 12 ... High voltage battery (one Embodiment of DC voltage source), 14 ... Control apparatus (One Embodiment of the control apparatus of a rotary machine).

Claims (10)

A DC / AC converter circuit including a switching element that selectively connects a terminal of a rotating machine to each of a positive electrode and a negative electrode of a DC voltage source, by turning on / off the switching element of the DC / AC converter circuit. In a control device for a rotating machine that controls a control amount having at least one of a flowing current, a torque of the rotating machine, and a magnetic flux of the rotating machine,
Predicting means for predicting the control amount when temporarily setting the operation state of the DC-AC converter circuit represented by a voltage vector in a fixed coordinate system;
A determination unit that evaluates an operation state corresponding to the predicted control amount based on the control amount predicted by the prediction unit, and determines a highly evaluated operation state as an operation state of the DC-AC converter circuit;
Operating means for operating the DC-AC converter circuit so as to be in the determined operating state;
When the determination means evaluates the operation state in order to make the determination, the evaluation of the specific operation state is increased, and the operation state that increases the evaluation is variably set according to the rotation angle of the rotating machine. A control device for a rotating machine comprising a variable means.   When the DC / AC converter circuit is operated according to the operation state that is most highly evaluated by the variable means, each terminal of the rotating machine is connected to the positive and negative terminals of the DC voltage source during one electrical angle cycle of the rotating machine. 2. The control device for a rotating machine according to claim 1, wherein the control is a rectangular wave control that is connected once to each. The determining means determines the operation state having the highest evaluation as the operation state of the DC-AC conversion circuit,
The variable means has the highest evaluation of the specific operation state when a difference between the predicted control amount corresponding to the specific operation state and a command value thereof is a predetermined value or less. The control device for a rotating machine according to claim 1 or 2, characterized in that:   4. The control device for a rotating machine according to claim 3, wherein the specified value is variably set by inputting at least one of a current flowing through the rotating machine, a torque of the rotating machine, and a rotation speed of the rotating machine. . The determination means increases the evaluation as the difference between the predicted control amount and the command value is small,
In the variable means, the relative error obtained by subtracting the absolute value of the difference corresponding to the specific operation state from the absolute value of the difference corresponding to the operation state other than the specific operation state is corrected to increase by the specified value. 5. The control device for a rotating machine according to claim 3, wherein an object is subjected to evaluation by the determining means. A phase setting means for setting a phase of an output voltage vector of the DC-AC converter circuit as an operation amount for controlling the control amount;
The rotation according to any one of claims 1 to 5, wherein the variable means increases the evaluation of the operating state of the DC / AC converter circuit determined by the phase set by the phase setting means. Machine control device.   The variable means variably sets an operation state for increasing the evaluation for each angle region obtained by dividing “1/2” of one electrical angle cycle of the rotating machine by the number of phases of the rotating machine. The control device for a rotating machine according to claim 6.   The control device for a rotating machine according to claim 6 or 7, wherein the phase setting means operates the phase as an operation amount for feedback-controlling the torque of the rotating machine to its command value.   The said operation means sets the update timing of the said operation state by the rotation angle period of the said rotary machine, when the variable setting by the said variable means is made, The rotation angle period of the said rotary machine is characterized by the above-mentioned. Rotating machine control device.   The operation means includes an operation state change timing instructed by the set phase as the operation state update timing when the phase is set by the phase setting means. The control apparatus of the rotary machine of any one of -8.

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