JP2012080774A - Power supply circuit for driving semiconductor switching element - Google Patents

Power supply circuit for driving semiconductor switching element Download PDF

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JP2012080774A
JP2012080774A JP2012013711A JP2012013711A JP2012080774A JP 2012080774 A JP2012080774 A JP 2012080774A JP 2012013711 A JP2012013711 A JP 2012013711A JP 2012013711 A JP2012013711 A JP 2012013711A JP 2012080774 A JP2012080774 A JP 2012080774A
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circuit
power supply
switching element
capacitor
driving
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JP5278568B2 (en
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Makoto Tanitsu
誠 谷津
Toshihisa Shimizu
敏久 清水
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Fuji Electric Co Ltd
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Abstract

PROBLEM TO BE SOLVED: To resolve problems that, in a conventional self-contained gate driving power supply circuit, a charging and discharging loss in a snubber resistance becomes large since an RCD snubber circuit is used in charging and discharging operations, and that a conversion efficiency is reduced in a high-frequency operation, and that a device becomes large in size, and further, to resolve a problem that a gate of an IGBT (Insulated Gate Bipolar Transistor) cannot be reverse-biased since only a single power source is produced as a driving power source, resulting in a large turn-off loss.SOLUTION: A power supply circuit for driving a semiconductor switching element has a bypass capacitor in which semiconductor switching elements in an upper-and-lower-arm pair are on-off operated alternately to repeat charge and discharge by variation in a potential difference between an upper arm side switching element driving circuit and a lower arm side switching element driving circuit. A charge and discharge current of the bypass capacitor is rectified by rectification circuits provided in respective power supply parts provided in the upper arm side switching element driving circuit and the lower arm side switching element driving circuit. The outputs of the rectification circuits are given as respective power sources for the switching element driving circuits.

Description

本発明は、半導体スイッチング素子をオンオフさせるための駆動回路の駆動用電源回路に関するもので、詳しくは自給式の駆動用電源回路における小形、低価格化のための回路構成技術に関する。   The present invention relates to a drive power supply circuit for a drive circuit for turning on and off a semiconductor switching element, and more particularly to a circuit configuration technique for reducing the size and cost of a self-powered drive power supply circuit.

図9に、従来の自給式駆動用電源回路を用いたIGBT駆動回路を、図10にゲート駆動回路の詳細を示す。図9の回路構成では、IGBT4の主端子(コレクタ、エミッタ)と並列にダイオード101、抵抗102およびスナバコンデンサ103からなるRCDスナバ回路100が並列に、このRCDスナバ回路100のスナバコンデンサ103と並列にコンデンサ104とコンデンサ105の直列回路が、各々接続されている。さらにコンデンサ105にはダイオード107が並列に接続され、コンデンサ104と105の直列接続点はダイオード106を介しゲート駆動回路3の電源用コンデンサ8に接続されている。図10にゲート駆動回路30の詳細を示す。電源用コンデンサ8と並列にトランジスタ301とトランジスタ302からなるコンプリメンタリ回路が接続され、さらにこのコンプリメンタリ回路の出力は抵抗303を介してIGBT4のゲートに接続される。このような構成において、トランジスタ301がオンすると、IGBT4のゲート容量が充電され、IGBTはオン状態となる。トランジスタ302がオンすると、IGBTのゲート容量は放電され、IGBTはオフ状態となる。   FIG. 9 shows an IGBT drive circuit using a conventional self-powered drive power supply circuit, and FIG. 10 shows details of the gate drive circuit. In the circuit configuration of FIG. 9, an RCD snubber circuit 100 including a diode 101, a resistor 102 and a snubber capacitor 103 is parallel to the main terminal (collector, emitter) of the IGBT 4, and parallel to the snubber capacitor 103 of the RCD snubber circuit 100. A series circuit of a capacitor 104 and a capacitor 105 is connected to each other. Further, a diode 107 is connected in parallel to the capacitor 105, and a series connection point between the capacitors 104 and 105 is connected to the power supply capacitor 8 of the gate drive circuit 3 via the diode 106. FIG. 10 shows details of the gate drive circuit 30. A complementary circuit composed of a transistor 301 and a transistor 302 is connected in parallel with the power supply capacitor 8, and the output of the complementary circuit is connected to the gate of the IGBT 4 through a resistor 303. In such a configuration, when the transistor 301 is turned on, the gate capacitance of the IGBT 4 is charged, and the IGBT is turned on. When the transistor 302 is turned on, the gate capacitance of the IGBT is discharged, and the IGBT is turned off.

図9の回路構成において、IGBT4がオン状態からオフになると、スナバコンデンサ103はダイオード101を介して充電され、これに伴い、コンデンサ104、105も充電される。この充電動作によりコンデンサ105の電圧が電源用コンデンサ8の電圧より高くなるとダイオード106が導通して、電源用コンデンサ8が充電されることになる。次にIGBT4がオンすると、コンデンサ103、104、105に蓄えられていた電荷はスナバ抵抗102、IGBT4を介して放電される。図5の回路の詳細な動作については、特許文献1に開示されている。
特開8−51770号公報
In the circuit configuration of FIG. 9, when the IGBT 4 is turned off from the on state, the snubber capacitor 103 is charged via the diode 101, and accordingly, the capacitors 104 and 105 are also charged. When the voltage of the capacitor 105 becomes higher than the voltage of the power supply capacitor 8 by this charging operation, the diode 106 becomes conductive and the power supply capacitor 8 is charged. Next, when the IGBT 4 is turned on, the electric charge stored in the capacitors 103, 104, and 105 is discharged through the snubber resistor 102 and the IGBT 4. The detailed operation of the circuit of FIG. 5 is disclosed in Patent Document 1.
JP-A-8-51770

上述のように、図9に示した従来の自給型ゲート駆動用電源回路では、RCDスナバ回路100を充放電動作で使用するため、スナバ抵抗102における充放電損失が大きくなり高周波動作では変換効率が低下すると同時に、装置が大型になる。さらに駆動用の電源として正極の単一電源しか作れないため、IGBTのゲートに逆バイアスをかけることができずターンオフ損失が大きいという課題がある。   As described above, the conventional self-powered gate drive power supply circuit shown in FIG. 9 uses the RCD snubber circuit 100 in charge / discharge operation, so that the charge / discharge loss in the snubber resistor 102 increases, and the conversion efficiency is improved in high-frequency operation. At the same time, the device becomes larger. Furthermore, since only a positive single power source can be made as a power source for driving, there is a problem that a reverse bias cannot be applied to the gate of the IGBT and a turn-off loss is large.

上述の課題を解決するために、第1の発明においては、直流電源と、前記直流電源に並列に接続される直列接続された上下アーム対の半導体スイッチング素子と、これら上下アームをそれぞれ駆動する二つのスイッチング素子駆動回路を含むスイッチング回路において、前記上下アーム対の半導体スイッチング素子が交互にオンオフ動作することにより、前記上アーム側スイッチング素子駆動回路と、前記下アーム側スイッチング素子駆動回路との間に生じる電位差変動により充放電を繰り返すバイパスコンデンサを備え、前記バイパスコンデンサの充放電電流を前記上アーム側スイッチング素子駆動回路と前記下アーム側スイッチング素子駆動回路の各々の電源部に設けた整流回路で整流し、前記整流回路の出力を各々のスイッチング素子駆動回路用電源とする。
第2の発明においては、第1の発明における前記バイパスコンデンサと直列にリアクトルを接続する。
In order to solve the above-described problems, in the first invention, a DC power source, a semiconductor switching element of a pair of upper and lower arms connected in parallel to the DC power source, and two upper and lower arms respectively driven. In the switching circuit including two switching element driving circuits, the semiconductor switching elements of the upper and lower arm pairs are alternately turned on and off, so that the upper arm side switching element driving circuit and the lower arm side switching element driving circuit are A bypass capacitor that repeatedly charges and discharges due to potential difference fluctuations generated, and the charge and discharge current of the bypass capacitor is rectified by a rectifier circuit provided in each power supply section of the upper arm side switching element driving circuit and the lower arm side switching element driving circuit The output of the rectifier circuit is connected to each switching element driver. The power supply for the dynamic circuit.
In the second invention, a reactor is connected in series with the bypass capacitor in the first invention.

第3の発明においては、第1の発明における前記各々の電源部に設けた整流回路は、ダイオード直列回路とコンデンサ直列回路の並列回路で構成する。
第4の発明においては、第1の発明における前記各々の電源部に設けた整流回路は、ダイオード直列回路とコンデンサ直列回路をリアクトルを介して並列接続して構成する。
第5の発明においては、第1〜第4の発明における前記直流電源から直流−直流変換回路を通して前記整流回路のコンデンサ直列回路を充電する。
In the third invention, the rectifier circuit provided in each of the power supply units in the first invention is constituted by a parallel circuit of a diode series circuit and a capacitor series circuit.
In the fourth invention, the rectifier circuit provided in each of the power supply units in the first invention is configured by connecting a diode series circuit and a capacitor series circuit in parallel via a reactor.
In the fifth invention, the capacitor series circuit of the rectifier circuit is charged from the DC power source in the first to fourth inventions through a DC-DC converter circuit.

本発明では、スイッチング素子の主端子間を整流して正負の駆動用電源を得ているため、ゲート駆動用の順バイアス電源だけでなく逆バイアス電源も作ることができ、動作損失を低損失化できる。結果として、装置の高効率化だけでなく、低価格化、小形化及び逆バイアス電源利用による装置の高信頼性化が可能となる。   In the present invention, the positive and negative drive power supplies are obtained by rectifying the main terminals of the switching elements, so that not only a forward bias power supply for gate drive but also a reverse bias power supply can be created, and operating loss is reduced. it can. As a result, not only high efficiency of the apparatus but also low cost, downsizing, and high reliability of the apparatus by using a reverse bias power source are possible.

本発明の第1の要点は、半導体スイッチング素子の両端電圧を直列コンデンサを介し、ダイオード直列回路とコンデンサ直列回路とを並列接続して構成したハーフブリッジ形ダイオード整流器により整流して得られた直流中間点を持つ正負の直流電圧のうち、正側直流電圧を前記半導体スイッチング素子をオン駆動するための駆動回路の順バイアス側電源として供給し、負側直流電圧を前記半導体スイッチング素子をオフ駆動するための駆動回路の逆バイアス側電源として供給することである。   The first essential point of the present invention is a DC intermediate obtained by rectifying a voltage across a semiconductor switching element through a series capacitor and a half-bridge type diode rectifier configured by connecting a diode series circuit and a capacitor series circuit in parallel. Among positive and negative DC voltages having a point, a positive DC voltage is supplied as a forward bias side power source of a drive circuit for driving the semiconductor switching element on, and a negative DC voltage is driven off the semiconductor switching element Is supplied as a reverse bias side power source of the driving circuit.

また、本発明の第2の要点は、上下アーム対の半導体スイッチング素子が交互にオンオフ動作することにより、上アーム側スイッチング素子駆動回路と、下アーム側スイッチング素子駆動回路との間に生じる電位差変動により充放電を繰り返すバイパスコンデンサを備え、前記バイパスコンデンサの充放電電流を前記上アーム側スイッチング素子駆動回路と前記下アーム側スイッチング素子駆動回路の各々の電源部に設けた整流回路で整流して得られた直流中間点を持つ正負の直流電圧のうち、正側直流電圧を前記半導体スイッチング素子をオン駆動するための駆動回路の順バイアス側電源として供給し、負側直流電圧を前記半導体スイッチング素子をオフ駆動するための駆動回路の逆バイアス側電源として供給することである。  In addition, the second essential point of the present invention is that the potential difference fluctuation generated between the upper arm side switching element driving circuit and the lower arm side switching element driving circuit by alternately turning on and off the semiconductor switching elements of the upper and lower arm pairs. A bypass capacitor that repeatedly charges and discharges, and the charge and discharge current of the bypass capacitor is obtained by rectifying by a rectifier circuit provided in each power supply section of the upper arm side switching element driving circuit and the lower arm side switching element driving circuit. Among positive and negative DC voltages having a DC intermediate point, a positive DC voltage is supplied as a forward bias side power source of a driving circuit for driving the semiconductor switching element on, and the negative DC voltage is supplied to the semiconductor switching element. The power supply is supplied as a reverse bias side power source of a drive circuit for driving off.

図1に、本発明の第1の実施例を示す回路構成図を示す。図1では、IGBT4を駆動するためのゲート駆動回路3の出力がIGBT4のゲートとエミッタに接続されている。また電源用コンデンサ8、9とダイオード6、7によりハーフブリッジ形ダイオード整流回路200が構成され、その電源用コンデンサ8、9は、各々ゲート駆動回路3の順バイアス用電源及び逆バイアス用電源となる。ハーフブリッジ形ダイオード整流回路200のダイオード側交流入力点は直列にコンデンサ5を介してIGBT4のコレクタに、電源用コンデンサ8、9の直列接続点は直接IGBT4のエミッタに、各々接続された構成である。   FIG. 1 is a circuit configuration diagram showing a first embodiment of the present invention. In FIG. 1, the output of the gate drive circuit 3 for driving the IGBT 4 is connected to the gate and emitter of the IGBT 4. The power supply capacitors 8 and 9 and the diodes 6 and 7 constitute a half-bridge type diode rectifier circuit 200. The power supply capacitors 8 and 9 serve as a forward bias power supply and a reverse bias power supply for the gate drive circuit 3, respectively. . The diode-side AC input point of the half-bridge type diode rectifier circuit 200 is connected in series to the collector of the IGBT 4 via the capacitor 5, and the series connection point of the power capacitors 8 and 9 is directly connected to the emitter of the IGBT 4. .

このような回路構成において、IGBT4がオン状態からオフすると、直流電源1→負荷2→コンデンサ5→ダイオード6→電源用コンデンサ8→直流電源1の経路で電流が流れ、コンデンサ5と同時に電源用コンデンサ8を充電する。次にIGBT4がオンすると、コンデンサ5に充電されていたエネルギーにより、コンデンサ5→IGBT4→電源コンデンサ9→ダイオード7→コンデンサ5の経路で電流が流れ、電源コンデンサ9を充電する。  In such a circuit configuration, when the IGBT 4 is turned off from the ON state, a current flows through the path of the DC power source 1 → the load 2 → the capacitor 5 → the diode 6 → the power source capacitor 8 → the DC power source 1. 8 is charged. Next, when the IGBT 4 is turned on, due to the energy charged in the capacitor 5, a current flows through the path of the capacitor 5 → IGBT 4 → the power supply capacitor 9 → the diode 7 → the capacitor 5, and the power supply capacitor 9 is charged.

このIGBT4のスイッチングによるオンオフの一連の動作により、IGBT4を駆動するための電源となる電源用コンデンサ8、9の充電が低損失で可能となる。
図4にゲート駆動回路3の詳細例を示す。電源用コンデンサ8と9の直列回路と並列にトランジスタ301とトランジスタ302からなるコンプリメンタリ回路が接続され、さらにこのコンプリメンタリ回路の出力は抵抗303を介してIGBT4のゲートに、電源用コンデンサ8と9の直列接続点はIGBTのエミッタに、各々接続される。トランジスタ301がオンすると、電源用コンデンサ8でIGBT4のゲート容量が充電され、IGBTはオン状態となる。トランジスタ302がオンすると、電源用コンデンサ9でIGBTのゲート容量は放電後逆バイアスされ、IGBTはオフ状態となる。
This series of on / off operations by switching of the IGBT 4 enables charging of the power supply capacitors 8 and 9 serving as a power source for driving the IGBT 4 with low loss.
FIG. 4 shows a detailed example of the gate drive circuit 3. A complementary circuit comprising a transistor 301 and a transistor 302 is connected in parallel with the series circuit of the power supply capacitors 8 and 9, and the output of this complementary circuit is connected to the gate of the IGBT 4 via the resistor 303 in series with the power supply capacitors 8 and 9. The connection point is connected to the emitter of the IGBT. When the transistor 301 is turned on, the gate capacitor of the IGBT 4 is charged by the power supply capacitor 8 and the IGBT is turned on. When the transistor 302 is turned on, the gate capacitance of the IGBT is reverse-biased after discharging by the power supply capacitor 9, and the IGBT is turned off.

図2に、本発明の第2の実施例を示す。図2は図1に示したIGBT駆動用電源回路を、ブリッジ接続されたIGBT式インバータ回路に適用した場合の構成図である。インバータ回路装置の起動前に、直流電源300により、ダイオード401、402を介して、2つのゲート駆動回路3a、3bの順バイアス側電源用コンデンサ8a、8bを予め充電しておくことで、インバータ回路の起動が可能になる。起動後の動作は図1と同様であるため、詳細は省略する。   FIG. 2 shows a second embodiment of the present invention. FIG. 2 is a configuration diagram when the IGBT driving power supply circuit shown in FIG. 1 is applied to a bridge-connected IGBT inverter circuit. Before starting the inverter circuit device, the forward bias side power supply capacitors 8a and 8b of the two gate drive circuits 3a and 3b are charged in advance by the DC power supply 300 via the diodes 401 and 402, thereby the inverter circuit. Can be activated. Since the operation after activation is the same as that shown in FIG.

図3に、本発明の第3の実施例を示す。本発明による実施例図1の変形例である。図1との相違点は、コンデンサ5とハーフブリッジ形ダイオード整流回路200との間にダイオード202と電流制限用抵抗201の並列回路が挿入されている点である。抵抗201により、IGBT4がオンした時のコンデンサ5の放電電流最大値を制限する。IGBT4の許容最大電流値が問題となる場合には有効な手段となる。   FIG. 3 shows a third embodiment of the present invention. Embodiment according to the present invention is a modification of FIG. The difference from FIG. 1 is that a parallel circuit of a diode 202 and a current limiting resistor 201 is inserted between the capacitor 5 and the half-bridge diode rectifier circuit 200. The resistor 201 limits the maximum discharge current value of the capacitor 5 when the IGBT 4 is turned on. This is an effective means when the allowable maximum current value of the IGBT 4 becomes a problem.

図5に、本発明の第4の実施例を示す。直列接続されたIGBT4a、4bを駆動するためのゲート駆動回路3a、3bが各々IGBT4a、4bのゲート−エミッタに接続される。また電源用コンデンサ8a、9aとダイオード6a、7aによりハーフブリッジ形ダイオード整流回路200aが構成され、その電源用コンデンサ8a、9aは、各々ゲート駆動回路3aの順バイアス用電源及び逆バイアス用電源として接続される。同様に、電源用コンデンサ8b、9bとダイオード6b、7bによりハーフブリッジ形ダイオード整流回路200bが構成され、その電源用コンデンサ8b、9bは、各々ゲート駆動回路3bの順バイアス用電源及び逆バイアス用電源として接続される。
各々のハーフブリッジ形ダイオード整流回路200a、200bのダイオード側交流入力点はコンデンサ5cを介して直列接続され、電源用コンデンサ8a、9aの中間接続点は直接IGBT4aのエミッタに、また電源用コンデンサ8b、9bの中間接続点は直接IGBT4bのエミッタに各々接続された構成である。
FIG. 5 shows a fourth embodiment of the present invention. Gate drive circuits 3a and 3b for driving IGBTs 4a and 4b connected in series are connected to gate-emitters of IGBTs 4a and 4b, respectively. The power supply capacitors 8a and 9a and the diodes 6a and 7a constitute a half-bridge type diode rectifier circuit 200a, and the power supply capacitors 8a and 9a are respectively connected as a forward bias power source and a reverse bias power source for the gate drive circuit 3a. Is done. Similarly, a half-bridge type diode rectifier circuit 200b is constituted by the power supply capacitors 8b and 9b and the diodes 6b and 7b. The power supply capacitors 8b and 9b are respectively a forward bias power supply and a reverse bias power supply for the gate drive circuit 3b. Connected as
The diode side AC input points of the half-bridge type diode rectifier circuits 200a and 200b are connected in series via the capacitor 5c, and the intermediate connection point of the power supply capacitors 8a and 9a is directly connected to the emitter of the IGBT 4a, and the power supply capacitor 8b, The intermediate connection point 9b is directly connected to the emitter of the IGBT 4b.

このような回路構成において、IGBT4aがオフ状態からオンし、IGBT4bがオン状態からオフすると、直流電源1b、1a→IGBT4a→電源用コンデンサ9a→ダイオード7a→コンデンサ5c→ダイオード6b→電源用コンデンサ8b→直流電源1b、1aの経路で電流が流れ、コンデンサ5cと同時に電源用コンデンサ8b、9aを充電する。次にIGBT4aがオフし4bがオンすると、コンデンサ5cに充電されていたエネルギーにより、コンデンサ5c→ダイオード6a→電源用コンデンサ8a→IGBT4b→電源コンデンサ9b→ダイオード7b→コンデンサ5cの経路で電流が流れ、電源コンデンサ9b、8aを充電する。
このIGBT4a、4bのスイッチングによるオンオフの一連の動作により、IGBT4a、4bを駆動するための電源となる電源用コンデンサ8a、8b、9a、9bの充電が低損失で可能となる。
In such a circuit configuration, when the IGBT 4a is turned on from the off state and the IGBT 4b is turned off from the on state, the DC power source 1b, 1a → IGBT 4a → power capacitor 9a → diode 7a → capacitor 5c → diode 6b → power capacitor 8b → A current flows through the paths of the DC power supplies 1b and 1a, and the power supply capacitors 8b and 9a are charged simultaneously with the capacitor 5c. Next, when the IGBT 4a is turned off and the 4b is turned on, the current flows in the path of the capacitor 5c → the diode 6a → the power supply capacitor 8a → the IGBT 4b → the power supply capacitor 9b → the diode 7b → the capacitor 5c due to the energy charged in the capacitor 5c. The power supply capacitors 9b and 8a are charged.
By a series of on / off operations by switching of the IGBTs 4a and 4b, charging of the power supply capacitors 8a, 8b, 9a and 9b serving as a power source for driving the IGBTs 4a and 4b can be performed with low loss.

図6に、本発明の第5の実施例を示す。図5に示したハーフブリッジ形ダイオード整流回路200a、200bの各々のダイオード側交流入力点間を直列接続するコンデンサ5cと直列にリアクトル11を挿入した構成である。このリアクトル11を追加することで、IGBT4a、4bのスイッチングにより流れるコンデンサ5cの充放電電流のピーク値を小さくする事が出来るため、ダイオード6a、6b、7a、7b、及び電源用コンデンサ8a、8b、9a、9bの電流定格を下げる効果があり、装置の一層の小形化が可能となる。   FIG. 6 shows a fifth embodiment of the present invention. This is a configuration in which a reactor 11 is inserted in series with a capacitor 5c that connects in series between the diode-side AC input points of the half-bridge type diode rectifier circuits 200a and 200b shown in FIG. By adding this reactor 11, the peak value of the charge / discharge current of the capacitor 5c that flows by switching of the IGBTs 4a and 4b can be reduced, so that the diodes 6a, 6b, 7a and 7b, and the power supply capacitors 8a and 8b, This has the effect of lowering the current ratings of 9a and 9b, and enables further miniaturization of the device.

図7に、本発明の第6の実施例を示す。図6で挿入したリアクトル11の代わりに、リアクトル11a、11bを各々ハーフブリッジ形ダイオード整流回路200a、200bの直流部分に挿入した構成である。   FIG. 7 shows a sixth embodiment of the present invention. Instead of the reactor 11 inserted in FIG. 6, the reactors 11 a and 11 b are inserted into the direct current portions of the half-bridge diode rectifier circuits 200 a and 200 b, respectively.

このような回路構成において、IGBT4aがオフ状態からオンし、IGBT4bがオン状態からオフすると、直流電源1b、1a→IGBT4a→電源用コンデンサ9a→ダイオード7a→コンデンサ5c→ダイオード6b→リアクトル11b→電源用コンデンサ8b→直流電源1b、1aの経路で電流が流れ、コンデンサ5cと同時に電源用コンデンサ8b、9aを充電し、さらにリアクトル11bにもエネルギーが蓄積される。コンデンサ5cの電圧が直流電源1a、1bの合計と同じになりその充電が完了したあとも、リアクトル11bに蓄えられたエネルギーにより、リアクトル11b→電源コンデンサ8b→電源コンデンサ9b→ダイオード7b→ダイオード6b→リアクトル11bの経路で電流が還流し、このリアクトル11bのエネルギーにより電源コンデンサ8b、9bのさらなる充電が可能になる。   In such a circuit configuration, when the IGBT 4a is turned on from the off state and the IGBT 4b is turned off from the on state, the DC power source 1b, 1a → IGBT 4a → power capacitor 9a → diode 7a → capacitor 5c → diode 6b → reactor 11b → power source A current flows through the path from the capacitor 8b to the DC power supply 1b, 1a, charges the power supply capacitors 8b, 9a simultaneously with the capacitor 5c, and further stores energy in the reactor 11b. Even after the voltage of the capacitor 5c becomes the same as the sum of the DC power supplies 1a and 1b and the charging is completed, the reactor 11b → the power supply capacitor 8b → the power supply capacitor 9b → the diode 7b → the diode 6b → with the energy stored in the reactor 11b. The current flows back through the path of the reactor 11b, and the energy of the reactor 11b enables further charging of the power supply capacitors 8b and 9b.

次にIGBT4aがオフし4bがオンすると、コンデンサ5cに充電されていたエネルギーにより、コンデンサ5c→ダイオード6a→リアクトル11a→電源用コンデンサ8a→IGBT4b→電源コンデンサ9b→ダイオード7b→コンデンサ5cの経路で電流が流れ、電源コンデンサ9b、8aを充電し、さらにリアクトル11aにエネルギーが蓄積される。コンデンサ5cの電圧がゼロとなりその放電が完了した後も、リアクトル11aに蓄積されたエネルギーは、リアクトル11a→電源コンデンサ8a→電源コンデンサ9a→ダイオード7a→ダイオード6a→リアクトル11aの経路で還流し、このリアクトル11aのエネルギーで電源コンデンサ8a、9aのさらなる充電が可能となる。このため、リアクトル11a、11bの追加により、図5の回路に対し、コンデンサ5cのピーク電流を下げるだけでなく、電源コンデンサ8a、8b、9a、9bの充電効率を高める効果があり、一層の小形化が可能となる。   Next, when the IGBT 4a is turned off and the 4b is turned on, the current in the path of the capacitor 5c → the diode 6a → the reactor 11a → the power supply capacitor 8a → the IGBT 4b → the power supply capacitor 9b → the diode 7b → the capacitor 5c due to the energy charged in the capacitor 5c. Flows, charging the power supply capacitors 9b and 8a, and further storing energy in the reactor 11a. Even after the voltage of the capacitor 5c becomes zero and the discharge is completed, the energy accumulated in the reactor 11a is circulated through the path of the reactor 11a → the power supply capacitor 8a → the power supply capacitor 9a → the diode 7a → the diode 6a → the reactor 11a. The power supply capacitors 8a and 9a can be further charged with the energy of the reactor 11a. Therefore, the addition of the reactors 11a and 11b has the effect of not only reducing the peak current of the capacitor 5c but also increasing the charging efficiency of the power supply capacitors 8a, 8b, 9a, and 9b with respect to the circuit of FIG. Can be realized.

図8に、本発明の第7の実施例を示す。図8の回路は図5の回路に対し、動作開始前の初期充電を行うため、直流−直流変換回路の一つであるシリーズレギュレータ回路19を追加し、ダイオード12で整流回路200aの電源コンデンサ8aを、ダイオード13で整流回路200bの電源コンデンサ8bを充電するようにした構成である。IGBT4a、4bの起動前にシリーズレギュレータ回路19より、直流電源1a、1bの電圧を使い、2つのゲート駆動回路3a、3bの順バイアス側電源用コンデンサ8a、8bを予め充電しておくことで、IGBT4a、4bの起動が可能になる。起動後の電源は図5〜図7の説明の通り、コンデンサ5cの充放電により自給している。この結果、シリーズレギュレータ回路19は初期充電用途だけであり、極小さい容量の回路が適用可能である。   FIG. 8 shows a seventh embodiment of the present invention. The circuit of FIG. 8 adds a series regulator circuit 19 which is one of DC-DC conversion circuits to perform initial charging before the operation starts with respect to the circuit of FIG. 5, and the power supply capacitor 8 a of the rectifier circuit 200 a with the diode 12. The power supply capacitor 8b of the rectifier circuit 200b is charged by the diode 13. Before starting the IGBTs 4a and 4b, by using the voltage of the DC power supplies 1a and 1b from the series regulator circuit 19, the forward bias side power supply capacitors 8a and 8b of the two gate drive circuits 3a and 3b are charged in advance. The IGBTs 4a and 4b can be activated. The power supply after startup is self-supplied by charging and discharging the capacitor 5c as described with reference to FIGS. As a result, the series regulator circuit 19 is only used for initial charging, and a circuit with a very small capacity is applicable.

本発明は、ゲート駆動回路の駆動用電源をスイッチング素子の主端子から得る回路方式であり、インバータ、UPS、スイッチング電源など、スイッチング動作で電力変換を行う装置のほとんどに適用が可能である。   The present invention is a circuit system that obtains a driving power source for a gate driving circuit from a main terminal of a switching element, and can be applied to almost all devices that perform power conversion by a switching operation, such as an inverter, a UPS, and a switching power source.

本発明の第1の実施例を示す回路図である。1 is a circuit diagram showing a first embodiment of the present invention. 本発明の第2の実施例を示す回路図である。FIG. 6 is a circuit diagram showing a second embodiment of the present invention. 本発明の第3の実施例を示す回路図である。FIG. 4 is a circuit diagram showing a third embodiment of the present invention. 図1〜図3のゲート駆動回路の構成例を示す。An example of the configuration of the gate drive circuit of FIGS. 本発明の第4の実施例を示す回路図である。It is a circuit diagram which shows the 4th Example of this invention. 本発明の第5の実施例を示す回路図である。It is a circuit diagram which shows the 5th Example of this invention. 本発明の第6の実施例を示す回路図である。It is a circuit diagram which shows the 6th Example of this invention. 本発明の第7の実施例を示す回路図である。It is a circuit diagram which shows the 7th Example of this invention. 従来のゲート駆動電源回路例を示す。An example of a conventional gate drive power supply circuit is shown. 図5のゲート駆動回路の詳細を示す。6 shows details of the gate drive circuit of FIG.

1、1a、1b・・・直流電源 2・・・負荷
3、3a、3b、30・・・ゲート駆動回路 4、4a、4b・・・IGBT
5、5a、5b、5c、104、105・・・コンデンサ
103・・・スナバコンデンサ
6、7、6a、7a、6b、7b、101、106、107、202、401、402・・・ダイオード 直流電源・・・300
8、9、8a、9a、8b、9b・・・電源用コンデンサ
11、11a、11b・・・リアクトル
102、201、303・・・抵抗
200、200a、200b・・・ハーフブリッジ形ダイオード整流回路
301、302・・・トランジスタ
1, 1a, 1b ... DC power supply 2 ... Load
3, 3a, 3b, 30 ... gate drive circuit 4, 4a, 4b ... IGBT
5, 5a, 5b, 5c, 104, 105 ... capacitors
103 ... Snubber capacitor
6, 7, 6a, 7a, 6b, 7b, 101, 106, 107, 202, 401, 402 ... Diode DC power supply ... 300
8, 9, 8a, 9a, 8b, 9b ... power supply capacitors 11, 11a, 11b ... reactor
102, 201, 303... Resistance
200, 200a, 200b... Half-bridge diode rectifier circuits 301, 302.

Claims (5)

直流電源と、前記直流電源に並列に接続される直列接続された上下アーム対の半導体スイッチング素子と、これら上下アームをそれぞれ駆動する二つのスイッチング素子駆動回路を含むスイッチング回路において、前記上下アーム対の半導体スイッチング素子が交互にオンオフ動作することにより、前記上アーム側スイッチング素子駆動回路と、前記下アーム側スイッチング素子駆動回路との間に生じる電位差変動により充放電を繰り返すバイパスコンデンサを備え、前記バイパスコンデンサの充放電電流を前記上アーム側スイッチング素子駆動回路と前記下アーム側スイッチング素子駆動回路の各々の電源部に設けた整流回路で整流し、前記整流回路の出力を各々のスイッチング素子駆動回路用電源とすることを特徴とする半導体スイッチング素子駆動用電源回路。   In a switching circuit including a DC power supply, a semiconductor switching element of a pair of upper and lower arms connected in parallel to the DC power supply, and two switching element driving circuits for driving the upper and lower arms, respectively, A bypass capacitor that repeats charging and discharging due to potential difference fluctuations generated between the upper arm side switching element driving circuit and the lower arm side switching element driving circuit by alternately turning on and off the semiconductor switching element; Is rectified by a rectifier circuit provided in each power supply section of the upper arm side switching element drive circuit and the lower arm side switching element drive circuit, and the output of the rectifier circuit is a power supply for each switching element drive circuit A semiconductor switch characterized by Grayed element driving power source circuit. 前記バイパスコンデンサと直列にリアクトルを接続することを特徴とする請求項1に記載の半導体スイッチング素子駆動用電源回路。   2. The power supply circuit for driving a semiconductor switching element according to claim 1, wherein a reactor is connected in series with the bypass capacitor. 前記各々の電源部に設けた整流回路は、ダイオード直列回路とコンデンサ直列回路との並列回路で構成することを特徴とする請求項1に記載の半導体スイッチング素子駆動用電源回路。   2. The semiconductor switching element driving power supply circuit according to claim 1, wherein the rectifier circuit provided in each of the power supply units is configured by a parallel circuit of a diode series circuit and a capacitor series circuit. 前記各々の電源部に設けた整流回路は、ダイオード直列回路とコンデンサ直列回路とをリアクトルを介して並列接続して構成することを特徴とする請求項1に記載の半導体スイッチング素子駆動用電源回路。   2. The semiconductor switching element driving power supply circuit according to claim 1, wherein the rectifier circuit provided in each of the power supply units is configured by connecting a diode series circuit and a capacitor series circuit in parallel via a reactor. 前記直流電源から直流−直流変換回路を通して前記整流回路のコンデンサ直列回路を充電することを特徴とする請求項1〜4のいずれか1項に記載の半導体スイッチング素子駆動用電源回路。   5. The power supply circuit for driving a semiconductor switching element according to claim 1, wherein the capacitor series circuit of the rectifier circuit is charged from the DC power supply through a DC-DC conversion circuit.
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CN110896306A (en) * 2018-09-12 2020-03-20 立积电子股份有限公司 Control circuit with bypass function

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JP2012222954A (en) * 2011-04-08 2012-11-12 Shindengen Electric Mfg Co Ltd Driving circuit
CN110896306A (en) * 2018-09-12 2020-03-20 立积电子股份有限公司 Control circuit with bypass function
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