JP2012058008A - Method for converting analog signal into digital signal - Google Patents

Abstract

PROBLEM TO BE SOLVED: To facilitate malfunction detection by acquiring a frequency modulation signal from an accumulator disposed within the negative feedback control system, while eliminating a subtractor conventionally disposed outside a negative feedback control system.SOLUTION: A method for converting analog signal into digital signal performs: a frequency modulation signal(ωt+φ)which is obtained by adding phase reference (ωt) to a digital angle output (φ) is obtained by an accumulator (20) disposed between a second counter (63) and a second pseudo sine wave generation unit (64) in a negative feedback control system (6) connected to a control deviation calculation unit (4); and the digital angle output (φ) from the second counter (63) is directly output.

Description

The present invention relates to a method for converting an analog signal into a digital signal, and in particular, eliminates a subtractor provided outside the negative feedback control system, and generates a frequency modulation signal (ω _R t + φ) by an adder provided inside the negative feedback control system. The present invention relates to a novel improvement for directly outputting a digital angle output φ from a counter so that it can be covered by detecting an abnormality of a negative feedback control system without detecting an abnormality of a subtractor.

Conventionally used analog signal digital conversion methods of this type include the analog signal digital conversion method disclosed in Patent Document 1, for example.
That is, although not shown, a subtractor is provided outside the closed-loop negative feedback control system including the resolver, and the phase reference (ω _R t) is subtracted from the frequency modulation signal (ω _R t + φ) by this subtractor to obtain the digital angle Output (φ) was output.

JP 2009-168546 A

Since the conventional analog signal digital conversion method is configured as described above, the following problems exist.
That is, in the above-described conventional configuration, the digital reference output (φ _R t) is subtracted by the subtractor provided outside the negative feedback control system to obtain the digital angle output φ. Even if is normal, if the subtractor fails, a normal angle output cannot be obtained.
Therefore, when detecting an abnormality in the digital angle output φ, it is necessary to separately detect the abnormality of the negative feedback control system and the abnormality of the subtracter itself, and the abnormality detection operation requires time and cost. It was.

Digital conversion method of the analog signal according to the present invention, first and second amplitude modulation signals sin [theta · sin from the rotation detector excitation signal sin .omega _{E t} from the excitation signal source is amplitude-modulated by the input angle θ (ω _E t −α), cos θ · sin (ω _E t−α) is input to the rotation signal processing means, and the input is performed by performing signal processing to make the control deviation ε zero by negative feedback control of the rotation signal processing means. when converting the angle θ to the digital angle output phi, said excitation signal source is independent from the negative feedback control of the rotation signal processing means, and any fixed frequency excitation frequency omega _{E t} of the excitation signal sin .omega _{E t} In the digital signal conversion method, the rotation signal processing means includes the first and second amplitude modulation signals sin θ · sin (ω _E t−α), cos θ · sin (ω _E t−α). A control deviation calculation unit that outputs the control deviation ε based on the first pseudo sine wave signals sinω _R t, cos ω _R t and the second pseudo sine wave signals sin ωt, cos ωt, an oscillation unit, a first counter, and a first has a quasi-sine wave generating unit, the phase reference based on a reference clock oscillation unit outputs omega _{R t} and the first quasi-sine wave signal sin .omega _{R t,} and phase reference generation means for generating and cos .omega _{R t,} A synchronous detection unit, a loop compensator, a signal converter, a second counter, and a second pseudo sine wave generation unit, based on the control deviation ε from the control deviation calculation unit and the frequency modulation signal ω _R t + φ and the A negative feedback control system that generates the second pseudo sine wave signals sinωt and cosωt, and an adder provided between the second counter and the second pseudo sine wave generator, from the second counter. Desi Le angle output (phi) and the adds the phase reference (ω _{R t),} and enter the adder obtained in the frequency-modulated signal (ω _{R t} + φ) on the second pseudo sinusoid generator, wherein In this method, the digital angle output (φ) is directly output from the second counter.

Since the analog signal-to-digital conversion method according to the present invention is configured as described above, the following effects can be obtained.
That is, the first and second amplitude modulation signals sin [theta · sin from the rotation detector excitation signal sin .omega _E t from the excitation signal source is amplitude-modulated by the input angle _{θ (ω E t-α)} , cosθ · sin (ω _E t−α) is input to the rotation signal processing means, and the input angle θ is converted into a digital angle output φ by performing signal processing that makes the control deviation ε zero by negative feedback control of the rotation signal processing means. to case, the made independent from the negative feedback control system of the rotation signal processing means for the excitation signal source, a digital conversion method of the analog signal to the excitation frequency omega _{E t} of the excitation signal sin .omega _{E t} any fixed frequency the rotation signal processing means, said first and second amplitude modulation signals _{sinθ · sin (ω E t-} α), cosθ · sin (ω E t-α) and the first pseudo sine wave signal sin .omega _R t, has osω _{R t} and a second quasi-sine wave signal sin .omega.t, a control deviation computing unit for outputting the control deviation ε based on the cos .omega.t, oscillating unit, a first counter, and the first pseudo sinusoid generator, wherein wherein a phase reference omega _{R t} based on the reference clock oscillation unit outputs the first quasi-sine wave signal sin .omega _{R t,} and phase reference generation means for generating and cos .omega _{R t,} the synchronous detector, the loop compensator, the signal conversion And a second counter and a second pseudo sine wave generator, and based on the control deviation ε from the control deviation calculator, the frequency modulation signal ω _R t + φ and the second pseudo sine wave signals sin ωt, cos ωt, A negative feedback control system for generating a digital angle output (φ) from the second counter and the phase by an adder provided between the second counter and the second pseudo sine wave generator. Standard (ω _R t) is added, and the frequency modulation signal (ω _R t + φ) obtained by the adder is input to the second pseudo sine wave generator, and the digital angle output (φ) is output from the second counter. By directly outputting, since the digital angle output can be obtained directly from the counter without using a subtracter outside the negative feedback control system as in the conventional case, the conventional abnormality detection for the subtractor alone can be performed. Even if not, it is possible to easily determine the presence or absence of the entire abnormality by detecting the abnormality of the negative feedback control system.

It is a block diagram which shows the digital conversion method of the analog signal by this invention. It is the block diagram which showed the control deviation calculating part of FIG. 1 in detail.

The present invention eliminates the subtractor provided outside the negative feedback control system, and obtains the frequency modulation signal (ω _R t + φ) by the adder provided in the negative feedback control system, so that the digital angle output φ is output from the counter. It is an object of the present invention to provide a method for converting an analog signal into a digital signal, which can be directly covered by detecting an abnormality in a negative feedback control system without detecting an abnormality in a subtractor.

DESCRIPTION OF EXEMPLARY EMBODIMENTS Hereinafter, preferred embodiments of an analog signal digital conversion method according to the invention will be described with reference to the drawings.
FIG. 1 is a block diagram showing a method for digitally converting an analog signal according to the present invention.
In the figure, the rotation detector 1 is a one-phase excitation / 2-phase output type amplitude modulation type brushless resolver (BRX, VRX). The rotation detector 1 includes an excitation signal sinω _E from an excitation signal source 2. t is input via the excitation signal amplifier 2a. For convenience of explanation, the amplitude of the excitation signal sin .omega _E t is 1. The excitation signal source 2 is provided independently of the negative feedback control loop of the rotation signal processing means 3 to be described later, the excitation frequency omega _{E t} of the excitation signal sin .omega _{E t} is set to any fixed frequency . In this embodiment, since a resolver is used as the rotation detector 1, the excitation frequency ω _E t is set high enough to be used as a carrier wave of an amplitude modulation signal. When is used, the excitation frequency ω _E t can be set to zero.

In the rotation detector 1, the excitation signal sin ω _E t is amplitude-modulated by the input angle θ. That is, the rotation detector 1 outputs the first and second amplitude modulation signals sin θ · sin (ω _E t−α), cos θ · sin (ω _E t−α). Here, α indicates a phase shift caused by the rotation detector 1 itself, the sensor cable, and the like. The rotation detector 1 is connected to the rotation signal processing means 3, and the first and second amplitude modulation signals sin θ · sin (ω _E t−α), cos θ · sin (ω _E t−α). Is input to the rotation signal processing means 3.

The rotation signal processing unit 3 includes a control deviation calculation unit 4, a phase reference generation unit 5, and a negative feedback control system 6. The control deviation calculation unit 4 includes the first and second amplitude modulation signals sin θ · sin (ω _E t−α), cos θ · sin (ω _E t−α) from the rotation detector 1 and the phase reference. The first pseudo sine wave signal sin ω _R t, cos ω _R t from the generating means 5 and the second pseudo sine wave signal sin ω t, cos ω t from the negative feedback control system 6 are input, and control is performed based on these signals. Deviation ε [sin (θ−ωt + ω _R t) · sin (ω _E t−α)] is output. The generation of the control deviation ε will be described in detail later.

The phase reference generator 5 includes an oscillation unit 50, a first counter 51, and a first pseudo sine wave generation unit 52. The oscillating unit 50 oscillates and outputs a reference clock 50a, and the first counter 51 counts the reference clock 50a and outputs a phase reference ω _R t. In other words, the phase reference ω _R t is a periodic function that is sorted by the first counter 51 based on the reference clock 50a and has a constant rate of change. The first pseudo sinusoid generator 52, the phase reference ω on the basis of the _R t first pseudo sine wave signal sin .omega _R t, and outputs the cos .omega _R t. The pseudo sine wave signal is a staircase-like periodic function signal.

The negative feedback control system 6 includes a synchronous detector 60, a loop compensator 61, a signal converter 62, a second counter 63, and a second pseudo sine wave generator 64. The synchronous detector 60 performs synchronous detection of a control deviation ε [sin (θ−ωt + ω _R t) · sin (ω _E t−α)], and the loop compensator 61 is configured by the negative feedback control system 6. Loop compensation is performed to ensure the stability and high speed of the negative feedback control loop [Phase Locked Loop]. The signal converter 62 converts the loop-compensated control deviation ε into a digital signal. An adder 20 is provided between the second counter 63 and the second pseudo sine wave generator 64 provided in the negative feedback control system 6, and the phase reference ω _R t from the first counter 51 and the first counter The digital angle output φ from the 2 counter 63 is added by the adder 20, and the frequency modulation signal ω _R t + φ is input to the second pseudo sine wave generator 64 as a feedback signal.
Therefore, the adder 20 is arranged in the negative feedback control system 6 described above, and the digital angle output φ is not generated by using the subtracter outside the negative feedback control system 6 as in the prior art. Direct output from the counter 63 is possible.

  FIG. 2 shows the configuration of the control deviation calculation unit 4 of FIG. 1 in more detail. In FIG. 2, the pseudo sine wave generation unit is abbreviated as PSG (Pseud Sinusoid Generator). In the figure, the control deviation calculation unit 4 includes first to sixth multipliers 40 to 45, an adder 46, first and second calculators 47 and 48, and an excitation component extraction unit 49. .

First, ze is the output of the rotary detector 1 first and second amplitude modulation signals _{sinθ · sin (ω E t-} α), cosθ · sin (ω E t-α) , the first to fourth multipliers 40-43.

In the first multiplier 40, the following analog operation is performed by which the first amplitude modulation signal sin θ · sin (ω _E t−α) is multiplied by the second pseudo sine wave signal sin ωt from the second pseudo sine wave generator 64. To be implemented.
sin θ · sin (ω _E t−α) × sin ωt
= Sinθ · sin (ω _E t−α) · sinωt (1)

Similarly, the second multiplier 41 multiplies the second amplitude modulation signal cos θ · sin (ω _E t−α) by the second pseudo sine wave signal sin ωt from the second pseudo sine wave generator 64 as follows. Analog operations are performed.
cos θ · sin (ω _E t−α) × sin ωt
= Cos θ · sin (ω _E t−α) · sin ωt (2)

Similarly, the third multiplier 42 multiplies the first amplitude modulation signal sin θ · sin (ω _E t−α) by the second pseudo sine wave signal cos ωt from the second pseudo sine wave generator 64 as follows. Analog operations are performed.
sinθ · sin (ω _E t−α) × cosωt
= Sin θ · sin (ω _E t−α) · cos ωt (3)

Similarly, the fourth multiplier 43 multiplies the second amplitude modulation signal cos θ · sin (ω _E t−α) by the second pseudo sine wave signal cos ωt from the second pseudo sine wave generator 64 as follows. Analog operations are performed.
cos θ · sin (ω _E t−α) × cos ωt
= Cos θ · sin (ω _E t−α) · cos ωt (4)

Next, the adder 46 adds the output of the first multiplier 40 expressed by the above equation (1) and the output of the fourth multiplier 43 expressed by the above equation (4) as follows: To be implemented.
Equation (1) + Equation (4) = sin θ · sin (ω _E t−α) · sin ωt
+ Cos θ · sin (ω _E t−α) · cos ωt
= Cos (ωt−θ) · sin (ω _E t−α) (5)

Similarly, the first subtractor 47 subtracts the output of the third multiplier 42 expressed by the above equation (3) from the output of the second multiplier 41 expressed by the above equation (2) as follows. Is implemented.
Equation (2)-(3) = cos θ · sin (ω _E t−α) · sin ωt
−sin θ · sin (ω _E t−α) · cos ωt
= Sin (ωt−θ) · sin (ω _E t−α) (6)

Next, the fifth multiplier 44 multiplies the output of the adder 46 expressed by the above equation (5) and the first pseudo sine wave signal sinω _R t from the first pseudo sine wave generator 52 as follows. The analog operation is performed.
cos (ωt−θ) · sin (ω _E t−α) × sinω _R t
= Cos (ωt−θ) · sinω _R t · sin (ω _E t−α) (7)

Similarly, the sixth multiplier 45 multiplies the output of the first subtractor 47 expressed by the above equation (6) and the first pseudo sine wave signal cosω _R t from the first pseudo sine wave generator 52. The following analog operations are performed.
sin (ωt−θ) · sin (ω _E t−α) × cosω _R t
= Sin (ωt−θ) · cosω _R t · sin (ω _E t−α) (8)

Next, in the second computing unit 48, the following analog computation in which the output of the sixth multiplier 45 represented by the equation (8) is subtracted from the output of the fifth multiplier 44 represented by the equation (7). Is implemented.
Equation (7)-(8) = cos (ωt−θ) · sinω _R t · sin (ω _E t−α)
-Sin (ωt-θ) · cosω _R t · sin (ω _E t-α)
= Sin (θ−ωt + ω _R t) · sin (ω _E t−α)
= Ε (9)

That is, in the control deviation calculation unit 4, the analog signal processing expressed by the above formulas (1) to (9) is performed, and the first and second amplitude modulation signals sin θ ·· that are the outputs of the rotation detector 1. sin (ω _E t−α), cos θ · sin (ω _E t−α) is converted into a control deviation ε [sin (θ−ωt + ω _R t) · sin (ω _E t−α)].

The rotation signal processing means 3 of this embodiment is controlled so that the control deviation ε is always zero. This control deviation ε is input to the synchronous detection unit 60 and is synchronously detected by the excitation phase reference ω _E t-α from the excitation component extraction unit 49. That is, the control deviation ε is converted into the following equation by making the excitation component sin (ω _E t−α) an absolute value.
Ε = sin (θ−ωt + ω _R t) · | sin (ω _E t−α) | (10)

Here, the excitation phase reference ω _E t-α will be considered.
The excitation phase reference ω _E t-α serves as a synchronization reference signal for accurately demodulating the envelope (envelope) / amplitude component sin (θ = ωt + ω _R t) of the control deviation ε by the synchronous detector 60. Is. The excitation phase reference ω _{E t-α,} by adding a phase shift alpha to the exciting signal sin .omega _{E t} of the first and second amplitude-modulated signal has, is one that is to more accurately synchronous detection.
The excitation component extraction unit 49 uses the digital angle output φ to increase the amplitude from the first and second amplitude modulation signals sinθ · sin (ωt−α) · cosθ · sin (ωt−α). A signal is selected, and an excitation phase reference ω _E t-α is extracted from the selected signal.

Note that the control deviation 同期 after the synchronous detection expressed by the equation (10) can be regarded as a signal equivalently expressed by the following equation.
Ε≈sin (θ−ωt + ω _R t) (11)
As described above, the rotation signal processing means 3 according to this embodiment is controlled so that the control deviation ε expressed by the equation (9) is always zero. Since the control deviation ε can be considered by equivalently replacing it with the control deviation 示 shown by the equation (11), the following equation is established.
ε≈sin (θ−ωt + ω _R t) = 0 (12)
∴ ωt = θ + ω _R t (13)

  Next, the control deviation 同期 after synchronous detection expressed by the equation (11) is input to the loop compensator 61 and loop compensated by the loop compensator 61. Here, the loop compensator 61 performs general P1 compensation [P (proportional) +1 (integral)]. On the other hand, the control deviation Ε can be output to the outside as a monitor signal and used to grasp the behavior of the rotation signal processing means 3. By using the control deviation と し て as a monitor signal, system fail-safe such as abnormality detection and self-diagnosis can be realized.

  The output of the loop compensator 61 is input to the signal converter 62. As the signal converter 62, an A / D converter that outputs a parallel absolute value output (parallel digital output) is used, and an output 62a from the signal converter 62 is processed by a second counter 63 to output a digital angle output φ. Is output as

The frequency modulation signal ωt = ω _R t + φ output from the adder 20 is a parallel signal, and is converted into a staircase-like pseudo sine wave by the second pseudo sine wave generator 64, and the control deviation calculator Returned to 4. That is, a series of negative feedback control systems 6 is configured by feeding back the frequency modulation signal ωt = ω _R t + φ to the control deviation calculation unit 4.

  The final output in the rotation detection device of this embodiment is a digital angle output φ obtained by converting the analog input angle θ into digital. The digital angle output φ can be obtained directly from the second counter 63.

As described above, the phase reference ω _R t is divided by the first counter 51 based on the reference clock oscillated by the oscillating unit 50, and is a periodic function having a constant rate of change. The phase reference omega _R t is inputted to the first pseudo sinusoid generator 52, the first pseudo sine wave signal sin .omega _R t staircase wave is converted into cos .omega _R t. The first pseudo sine wave signals sin ω _R t and cos ω _R t are input to the fifth and sixth multipliers 44 and 45 and used for analog calculation.

  As a digital circuit including at least the first counter 51, the second counter 63, the subtractor, the first pseudo sine wave generation unit 52, the second pseudo sine wave generation unit 64, and the signal converter 62. A synchronization circuit using a single and the same reference clock 50a generated by the oscillator 50 is used.

Accordingly, the rotation detecting device of this embodiment, the excitation signal source 2 is independent from the negative feedback control loop of the rotation signal processing means 3, the excitation frequency omega _{E t} of the excitation signal sin .omega _{E t} any fixed frequency An analog signal digital conversion method is applied.

The rotation signal processing means 3 includes the first and second amplitude modulation signals sin θ · sin (ω _E t-α), cos θ · sin (ω _E t-α) and the first pseudo sine wave signal sin ω _R t. , Cos ω _R t and the second pseudo sine wave signals sin ωt, cos ωt, a control deviation calculation unit 4 that outputs the control deviation ε, an oscillation unit 50, a first counter 51, and a first pseudo sine wave generation unit 52. the a, the phase reference based on the reference clock 50a of the oscillation unit 50 outputs omega _R t and the first quasi-sine wave signal sin .omega _R t, a phase reference generation means 5 generates the cos .omega _R t, synchronous detection Unit 60, loop compensator 61, signal converter 62, second counter 63, and second pseudo sine wave generator 64, and based on the control deviation ε from the control deviation calculator 4, the frequency modulation signal ω said the _R t + _φ 2 quasi-sine wave signal sin .omega.t, is constituted by a negative feedback control system 6 for generating and cos .omega.t.

The gist of the present invention is as follows.
Excitation signal source first and second amplitude modulation signals sin [theta · sin from the rotation detector is amplitude modulated by the excitation signal sin .omega _{E t} from (2) the input angle _{θ (1) (ω E t} -α), cosθ Sin (ω _E t−α) is input to the rotation signal processing means (3), and signal processing is performed to make the control deviation ε zero by negative feedback control of the rotation signal processing means (3). When converting the input angle θ to the digital angle output φ,
Wherein said is independent from the negative feedback control of the rotation signal processing means excitation signal source (2) (3), the digital conversion of the analog signal to the excitation frequency omega _{E t} of the excitation signal sin .omega _{E t} any fixed frequency In the method
The rotation signal processing means (3)
The first and second amplitude modulation signals sin θ · sin (ω _E t-α), cos θ · sin (ω _E t-α), the first pseudo sine wave signals sin ω _R t, cos ω _R t, and the second pseudo sine wave. A control deviation calculation unit (4) for outputting the control deviation ε based on the signals sinωt and cosωt, an oscillation unit (50), a first counter (51), and a first pseudo sine wave generation unit (52). and, the oscillating unit (50) on the basis of the output to the reference clock (50a) is phase reference omega _{R t} and the first quasi-sine wave signal sin .omega _{R t,} the phase reference generator means for generating and cos .omega _{R t} (5) When,
A synchronous detector (60), a loop compensator (61), a signal converter (62), a second counter (63), and a second pseudo sine wave generator (64), and the control deviation calculator (4 A negative feedback control system (6) for generating a frequency modulation signal ω _R t + φ and the second pseudo sine wave signals sin ωt, cos ωt based on the control deviation ε from
An adder (20) provided between the second counter (63) and the second pseudo sine wave generator (64) is used to output the digital angle output (φ) and the phase from the second counter (63). A reference (ω _R t) is added, and the frequency modulation signal (ω _R t + φ) obtained by the adder (20) is input to the second pseudo sine wave generator (64), and the second counter (63) is a digital conversion method of an analog signal for directly outputting the digital angle output (φ).

DESCRIPTION OF SYMBOLS 1 Rotation detector 2 Excitation signal source 3 Rotation signal processing means 4 Control deviation calculation part 5 Phase reference generation means 6 Negative feedback control system 20 Adder 49 Excitation component extraction part 50 Oscillation part 50a Reference clock Phase reference ω _R t
51 First Counter 52 First Pseudo Sine Wave Generation Unit 60 Synchronous Detection Unit 61 Loop Compensator 62 Signal Converter 63 Second Counter 64 Second Pseudo Sine Wave Generation Unit

Claims (1)

Excitation signal source first and second amplitude modulation signals sin [theta · sin from the rotation detector is amplitude modulated by the excitation signal sin .omega _{E t} from (2) the input angle _{θ (1) (ω E t} -α), cosθ Sin (ω _E t−α) is input to the rotation signal processing means (3), and signal processing is performed to make the control deviation ε zero by negative feedback control of the rotation signal processing means (3). When converting the input angle θ to the digital angle output φ,
Wherein said is independent from the negative feedback control of the rotation signal processing means excitation signal source (2) (3), the digital conversion of the analog signal to the excitation frequency omega _{E t} of the excitation signal sin .omega _{E t} any fixed frequency In the method
The rotation signal processing means (3)
The first and second amplitude modulation signals sin θ · sin (ω _E t-α), cos θ · sin (ω _E t-α), the first pseudo sine wave signals sin ω _R t, cos ω _R t, and the second pseudo sine wave. A control deviation calculation unit (4) for outputting the control deviation ε based on the signals sinωt and cosωt, an oscillation unit (50), a first counter (51), and a first pseudo sine wave generation unit (52). and, the oscillating unit (50) on the basis of the output to the reference clock (50a) is phase reference omega _{R t} and the first quasi-sine wave signal sin .omega _{R t,} the phase reference generator means for generating and cos .omega _{R t} (5) When,
A synchronous detector (60), a loop compensator (61), a signal converter (62), a second counter (63), and a second pseudo sine wave generator (64), and the control deviation calculator (4 A negative feedback control system (6) for generating a frequency modulation signal ω _R t + φ and the second pseudo sine wave signals sin ωt, cos ωt based on the control deviation ε from
An adder (20) provided between the second counter (63) and the second pseudo sine wave generator (64) is used to output the digital angle output (φ) and the phase from the second counter (63). A reference (ω _R t) is added, and the frequency modulation signal (ω _R t + φ) obtained by the adder (20) is input to the second pseudo sine wave generator (64), and the second counter A digital conversion method for an analog signal, wherein the digital angle output (φ) is directly output from (63).

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