JP2002054948A - Resolver conversion device - Google Patents

Abstract

PROBLEM TO BE SOLVED: To provide a resolver conversion device capable of enhancing responsiveness to the variation of a rotation angle to be detected and of accurately detecting the rotation angle and rotation angular velocity with a relatively inexpensive and simple structure. SOLUTION: After a D.C. signal having a level proportional to the sine value and the cosine value of the rotation angle θ is produced from the output signals (b) and (c) of a resolver 1 by synchronous rectifying circuits 11 and 12, the signal is undergone analog-digital conversion carried out by-digital conversion circuits 13 and 14, and inputted to a digital calculation processing means 15. The digital calculation processing means 15 calculates an estimation value θp of the rotation angle and an estimation value ωp of the rotation angular velocity by using an observer 29 so as to cancel the phase difference (θ-θp) between the rotation angle θof a rotor 1a of the resolver 1 and its estimation value θp.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a resolver converter for generating data representing a rotation angle of a rotor of a resolver (specifically, an amplitude modulation type resolver) from an output signal of the resolver.

[0002]

2. Description of the Related Art A resolver has been conventionally used as a detector for detecting the rotation angle (rotational position) of various kinds of rotating bodies (for example, rotors of electric motors and generators). The resolver rotor detects the rotation angle. It is connected to the rotating body to be tried. Then, the resolver outputs a detection signal (analog signal) corresponding to the rotation angle of the rotor connected to the rotating body.

[0003] This type of resolver is classified into an amplitude modulation type and a phase modulation type in terms of the form of the detection signal, and the amplitude modulation type detects the rotation angle of the rotating body at a higher speed than the phase modulation type. In general, an amplitude modulation type resolver is used because the structure of the resolver is relatively simple and inexpensive.

With reference to FIG. 6, a description will be given of an amplitude modulation type resolver and a conventional resolver converter for detecting a rotation angle using the resolver.

In FIG. 6, reference numeral 1 denotes an amplitude modulation type resolver 1 schematically. This amplitude modulation type resolver 1 outputs two types of detection signals b and c when a sine wave excitation signal (voltage signal) a is applied to its excitation circuit (not shown). More specifically, the equation representing the voltage waveform of the excitation signal a applied to the excitation circuit of the resolver 1 is expressed as “E · sin (2πft)”.
(However, E: amplitude of excitation signal, f: frequency of excitation signal,
t: time), one detection signal b output from the resolver 1 is an amplitude modulation signal in which the excitation signal is amplitude-modulated by the sine value sin θ of the rotation angle θ of the rotor 1 a of the resolver 1,
That is, the voltage waveform is “KE · sin (2πft)
.Sin θ ”.
Another detection signal c output by the resolver 1 is:
An amplitude modulation signal in which the excitation signal is amplitude-modulated by the cosine value cos θ of the rotation angle θ of the rotor 1a of the resolver 1, that is,
The voltage waveform is an amplitude modulation signal represented by the formula “KE · sin (2πft) · cos θ”. Note that “K” in the expression of the voltage waveforms of the detection signals b and c is a transformation ratio of a rotary transformer (not shown) of the resolver 1.

The resolver conversion device relating to the amplitude modulation type resolver 1 is based on two types of amplitude modulation signals b and c generated by the resolver 1 according to the rotation angle θ of the rotor 1a as described above. This circuit generates data (digital data) representing an estimated value (detected value) of the rotation angle θ. As the resolver conversion device, a circuit configuration shown by the reference numeral 100 in FIG. Known from.

That is, the resolver conversion device 100
Represents a deviation (θ−θp) (hereinafter referred to as a phase difference) between the actual rotation angle θ of the rotor of the resolver 1 and the estimated rotation angle θp finally generated by the resolver conversion device 100 as “0”. Are converged and balanced.

More specifically, the resolver conversion device 100
Are multiplication circuits 101 and 102 for multiplying two types of detection signals (amplitude modulation signals) b and c output by the resolver 1 by a cosine value cos θp and a sine value sin θp of the current value of the estimated rotation angle θp, respectively. And these multiplication circuits 10
And an addition circuit 103 that adds the outputs of the first and second outputs. In this case, the addition circuit 103 calculates the excitation signal a (E · sin (2π · f ·) according to the addition theorem of the trigonometric function.
t)) is replaced by the sine value sin (θ−θ) of the phase difference (θ−θp).
p), a signal d whose amplitude is modulated, i.e., its voltage waveform is "KE * sin (2π * ft) * sin (θ-
θp) ”.

[0009] The resolver converter 100 inputs the output signal d of the adder circuit 103 to the synchronous rectifier circuit 104. In the synchronous rectification circuit 104, the addition circuit 10
3 is rectified in synchronization with the excitation signal a,
By removing the component of the excitation signal a, the phase difference (θ−
A DC voltage signal e having a level proportional to the sine value sin (θ−θp) of θp), specifically, a level of KE sin (θ−θp), is generated.

Further, the resolver conversion device 100 integrates the output signal of the synchronous rectifier circuit 104 by the integrator 105, and then inputs the output signal (DC voltage signal) of the integrator 105 to the voltage controlled oscillator 106, A pulse signal f corresponding to the level of the output voltage signal of 105 is generated. Then, the resolver conversion device 100 includes the voltage-controlled oscillator 106
The counter 107 counts the number of pulses of the pulse signal f output by the counter 107 to generate digital data representing the estimated value θp of the rotation angle of the rotor of the resolver 1.

In the resolver converter 100, the estimated value θp of the rotation angle represented by the digital data generated by the counter 107 is fed back to the multiplication circuits 101 and 102, so that the estimated value θp is basically obtained. In the integrator 105 in such a state that the actual rotation angle θ matches.
Output etc. are balanced. As a result, the estimated value θp of the rotation angle represented by the digital data generated by the counter 107 becomes equivalent to the detected value of the actual rotation angle θ, and the rotation angle θ is detected by the digital data of the counter 107. Can be done.

However, in the conventional resolver conversion device 100, the multiplication circuits 101 and 102, the addition circuit 103, the synchronous rectification circuit 104, the integrator 105, and the voltage controlled oscillator 106 are constituted by analog circuits. That is, the conventional resolver conversion device 100 is mainly configured with an analog circuit.

For this reason, it is easy to be affected by the drift phenomenon of each analog circuit due to a temperature change or the like, and it is difficult to secure sufficient accuracy of the estimated value θp of the rotation angle.
In particular, the multiplication circuits 101 and 102 often have an adverse effect on the accuracy of the estimated rotation angle θp due to the influence of their linearity.

Further, since the resolver conversion device 100 is mainly composed of an analog circuit, the estimated value θp for the rapid change of the rotation angle θ of the rotor 1a of the resolver 1 is obtained.
It is difficult to improve the followability (quick response) of the robot. For this reason,
For example, it is difficult to detect the rotation angle of a rotating body that rotates at a high speed such as 15000 rpm.

Further, there are many situations in which not only the rotation angle (rotation position) of the rotating body connected to the rotor 1a of the resolver 1 but also the rotation angular velocity is desired to be detected. The estimated value (detected value) of the rotational angular velocity must be separately obtained from the data of the estimated value θp of the rotational angle by arithmetic processing, which is inconvenient.

In order to eliminate such inconvenience, the output signals (amplitude modulated signals) b and c of the resolver 1 are A / D-converted, and the same processing as the processing of each analog circuit of the resolver converter 100 is performed. It is also conceivable to generate the data such as the estimated value of the rotation angle by performing the processing on software (digital processing) by a microcomputer or the like. However, since the output signals b and c of the resolver 1 include the component of the high-frequency (for example, 15 kHz) excitation signal a, high-speed sampling is required to digitize the component. For this reason, the resolver conversion device including the A / D converter and the CPU for processing digital data becomes extremely expensive.
This is disadvantageous in terms of cost. In addition, software for processing digital data is also complicated.

[0017]

SUMMARY OF THE INVENTION The present invention has been made in view of such a background, and can improve the responsiveness to a change in the rotation angle to be detected, and has a relatively inexpensive and simple configuration. It is an object of the present invention to provide a resolver conversion device that can accurately detect a rotation angle and a rotation angular velocity.

[0018]

In order to achieve the above object, a resolver converter according to the present invention has a structure in which a sine-wave excitation signal is applied to a resolver, and the resolver according to the rotation angle (θ) of its rotor. From the first and second amplitude modulation signals having amplitude levels proportional to the sine value and cosine value of the rotation angle, respectively, from the two types of amplitude modulation signals to be output. In the resolver conversion device for generating data representing the following, the sine value (si) of the rotation angle is obtained from each of the first and second amplitude modulation signals.
a first and a second DC signal generation circuit for generating a DC signal having a level proportional to the nθ) and the cosine value (cos θ);
And the output of each of the second DC signal generation circuits is A / D
A / D conversion circuit for converting, and digital data obtained by A / D conversion of the output of each of the first and second DC signal generation circuits are sampled in a predetermined control cycle, and based on the sampled digital data. And a digital arithmetic processing means for calculating an estimated value (θp) of the rotational angle of the resolver rotor and an estimated value (ωp) of the rotational angular velocity of the resolver sequentially. The digital arithmetic processing means estimates the latest value of digital data obtained by A / D converting the output of each of the first and second DC signal generation circuits by the A / D conversion circuit and the rotation angle. From the latest value, the phase difference (θ−θ) between the actual value (θ) of the rotation angle and the estimated value (θp) of the rotation angle is obtained.
θp), phase difference data generating means for sequentially generating phase difference data in the predetermined control cycle, and the estimated values (θp, ωp) of the rotation angle and the rotation angular velocity as calculation parameters, and the phase difference data represented by the phase difference data. And an observer means for calculating the calculation parameters while sequentially updating the calculation parameters according to the phase difference data so as to eliminate the phase difference (θ−θp).

According to the present invention, the first and the second
The sine value (sinθ) and the cosine value (cosθ) of the rotation angle (θ) of the resolver rotor are respectively expressed from the first and second amplitude modulation signals by the DC signal generation circuit and the A / D conversion circuit. Digital data is generated and sampled by the digital arithmetic processing means in a predetermined control cycle.

In this case, since the outputs (DC signals) of the first and second DC signal generation circuits have the components of the high-frequency excitation signal removed therefrom, the A / D conversion circuit and digital arithmetic processing means The digital data representing the sine value (sin θ) and the cosine value (cos θ) of the rotation angle (θ) can be sequentially taken into the digital arithmetic processing means without requiring the sampling at such a high speed.

Note that the first and second analog circuits
Each DC signal generation circuit may have the same circuit configuration as, for example, the synchronous rectifier circuit 104 of the above-described conventional resolver converter 100, but need not necessarily have the same circuit configuration as the synchronous rectifier circuit 104.

In the present invention, the digital arithmetic processing means includes the latest value of the sampled digital data,
That is, the latest value of the sine value (sin θ) and the cosine value (cos θ) of the rotation angle (θ) and the estimated value (θ p) of the rotation angle sequentially calculated by the digital arithmetic processing unit by the observer unit described later for each control cycle. From the latest value, phase difference data representing the phase difference (θ−θp) between the actual value (θ) of the rotation angle and the estimated value (θp) of the rotation angle, that is, estimation of the actual value (θ) of the rotation angle The phase difference data representing the error of the value (θp) is generated by the phase difference data generating means.

In this case, the phase difference data includes, for example, a sine value (sin (θ-θ) of the phase difference (θ-θp).
p)) and a tangent value (tan (θ−θp)).
The tangent value (tan (θ-θp)) is obtained by calculating the phase difference (θ-θp) ±
When the value is near π / 2, the value becomes very large.
In practice, the sine value (sin (θ−θ) of the phase difference (θ−θp)
Preferably, p)) is obtained as the phase difference data. Then, the phase difference data is converted into the phase difference (θ−θ).
p), the first sine value (sin (θ−θp))
The value obtained by multiplying the sine value (sinθ) of the actual rotation angle θ represented by the latest value of the sampling data on the DC signal generation circuit side by the cosine value (cosθp) of the latest value of the estimated rotation angle (θp) ( sin θ × cos θp) and the sine value (cos θ) of the actual rotation angle θ represented by the latest value of the sampling data on the second DC signal generation circuit side (cos θ). sinθp) multiplied by (cosθ ×
sin θp), the sine value (sin) of the phase difference (θ−θp) is calculated according to the addition theorem of the trigonometric function.
(Θ−θp)) can be calculated.

In the present invention, the digital arithmetic processing means further calculates the estimated value θp of the rotation angle and the estimated value ωp of the rotation angular velocity which is a temporal change rate (time differential value) of the rotation angle as a calculation parameter. Observer means for performing the phase difference (θ−θp)
Is calculated while sequentially updating the calculation parameters in accordance with the phase difference data so as to eliminate. Specifically, for example, the rotation angle estimation value θp and the rotation angular velocity estimation value ωp are calculated while being sequentially updated in each control cycle according to the value of the phase difference (θ−θp) represented by the phase difference data.

According to such an observer means, the estimated value θp of the rotation angle is directly adjusted by the feedback processing according to the phase difference (θ−θp), so that the estimated value θp for the change of the actual rotation angle θ is obtained. Can be improved, and even when the rotation angle θ changes at a high speed, the data of the estimated value θp of the rotation angle θ can be obtained by following the change. In addition, for each of the estimated value of the rotation angle θp and the estimated value of the rotation angular velocity ωp, a feedback gain that defines the amount of change in the value with respect to the phase difference (θ−θp) can be set separately. , The estimated value of the rotation angle θp and the estimated value of the rotational angular velocity ωp can be simultaneously and stably and accurately obtained.

Therefore, according to the present invention, the provision of the observer means in the digital arithmetic processing means makes it possible to calculate the estimated value θp of the rotation angle θ to be detected with respect to a change in the rotation angle θ. This can be performed while ensuring the quick response (followability), and the rotation angle of the rotating body that rotates at a higher speed can be detected in real time.

After the resolver output signal (amplitude modulated signal) is subjected to analog processing by the first and second DC signal generation circuits, the estimated values θp and ωp of the rotational angle and rotational angular velocity are obtained by digital processing. Therefore, it is possible to minimize the influence of circuit drift due to a temperature change or the like, and the estimated value θ of the rotation angle and the rotation angular velocity can be reduced.
The accuracy of p and ωp can be improved. Then, it is also possible to compensate for the influence of the drift of the first and second DC signal generation circuits as needed.

Further, since the digital arithmetic processing circuit is provided with digital data relating to the signal from which the frequency component of the excitation signal has been removed by the first and second DC signal generation circuits, the data is sampled at a very high speed. Can be captured and processed without the need for
As a result, the digital operation processing circuit can be configured at a low cost, and the processing can be relatively simplified.

In the present invention, as described above, the phase difference data generated by the phase difference data generating means is preferably a sine value (sin (θ-θp)) of the phase difference (θ-θp). However, in an angle range in which the absolute value of the actual phase difference (θ-θp) exceeds approximately π / 2 (90 degrees), the sine value (sin (sin ( θ-θ
The error of p)) increases. In such a situation, the observer means updates the estimated rotation angle θp and the estimated rotation angular velocity ωp by using the sine value (sin (θ−θp)) as the phase difference data as it is. In this case, the responsiveness to changes in the actual values of the estimated values θp and ωp may be reduced.

Therefore, in the present invention, the phase difference data generated by the phase difference data generating means is the phase difference (θ-θ
p), the sine value (sin (θ−θp))
The digital arithmetic processing means is configured to perform an A / D conversion of the output of each of the first and second DC signal generation circuits by the A / D conversion circuit, and obtain the latest value of digital data and the estimated value of the rotation angle. From the latest value, the phase difference (θ−θ
means for determining the cosine value (cos (θ-θp)) of the phase difference, and the phase difference data input to the observer means in accordance with the positive and negative polarities of the cosine value (cos (θ-θp)) of the phase difference. Means for limiting the value.

That is, when the cosine value (cos (θ-θp)) of the phase difference (θ-θp) is obtained, the absolute value of the actual phase difference (θ-θp) is π / 2 depending on the polarity. It can be seen whether or not the angle range is larger. Therefore, the cosine value (co
By limiting the value of the phase difference data according to the polarity of s (θ−θp)), the actual phase difference (θ
-Θp) can be prevented from becoming excessive. As a result, regardless of the angle range of the actual phase difference (θ−θp), quick response to changes in the actual values of the estimated value θp of the rotational angle and the estimated value ωp of the rotational angular velocity obtained by the observer means is ensured. be able to.

[0032]

DESCRIPTION OF THE PREFERRED EMBODIMENTS One embodiment of the present invention is shown in FIGS.
This will be described with reference to FIG. FIG. 1 is a block diagram showing a configuration of a resolver converter of the present embodiment, FIG. 2 is a circuit diagram showing a configuration of a synchronous rectifier circuit of the resolver converter, FIG. 3 is a flowchart showing processing of a limiter of the resolver converter, FIG. 4 is a diagram for explaining processing by the limiter, and FIG.
FIG. 5 is a diagram illustrating calculation data of an estimated value of an actual rotation angle by the resolver device of the embodiment.

Referring to FIG. 1, reference numeral 1 denotes the above-described amplitude modulation type resolver, and a rotor (not shown) for detecting a rotation angle and a rotation angular velocity is connected to its rotor 1a. As described above, when the sine wave excitation signal a is applied to the excitation circuit (not shown), the resolver 1 has two types of rotation angle θ (= rotation angle of the rotating body) of the rotor 1 a of the resolver 1. It generates and outputs amplitude modulation signals b and c. Here, in the present embodiment, the excitation signal a is
It is a signal having a voltage waveform of the form “E · sin (2πft)” (where E: amplitude of the excitation signal, f: frequency of the excitation signal, t: time), and the amplitude modulation signals b and c are
“KE · sin (2πft) · sinθ”, “KE · sin”
(2πft) · cos θ ”(where K is the transforming ratio of the resolver).

The resolver converter 10 according to the present embodiment for processing the amplitude modulated signals b and c output from the resolver 1
Are synchronous rectifier circuits 11 and 12 for inputting amplitude modulation signals b and c, respectively, and A / D converter circuits 13 and 14 for A / D converting the outputs (analog signals) of the synchronous rectifier circuits 11 and 12, respectively. Digital operation processing means 1 for processing the outputs (digital signals) of the A / D conversion circuits 13 and 14
And 5.

The synchronous rectifier circuits 11 and 12 are analog circuits corresponding to the first and second DC signal generating circuits in the present invention, respectively. The synchronous rectifier circuit 11 converts the amplitude modulated signal b into the rotor 1a of the resolver 1. And a circuit that generates a DC voltage signal s1 having a level proportional to the sine value sinθ of the rotation angle θ, and a level proportional to the cosine value cosθ of the rotation angle θ of the rotor 1a of the resolver 1 from the amplitude modulation signal c. Is a circuit that generates the DC voltage signal s2 of FIG.

In this case, each of the synchronous rectifier circuits 11 and 12
More specifically, for example, the circuit configuration is as shown in FIG.

That is, for example, the synchronous rectifier circuit 11 will be described. The synchronous rectifier circuit 11 includes a differential amplifying circuit 16 for inputting the amplitude modulation signal b of the resolver 1 and an output of the differential amplifying circuit 16 having positive and negative outputs. Inverting circuit 17 for inverting and outputting the polarity, differential amplifier circuit 16 and inverting circuit 1
7, an analog switch 18 for selectively outputting the output of
A comparator circuit 19 is provided which generates a rectangular wave signal synchronized with the excitation signal a of the resolver 1 from the excitation signal a and applies the signal to the analog switch 18 as a control signal for the switching operation of the analog switch 18.

The analog switch 18 is switched between the differential amplifier 16 side and the inverting circuit 17 side in synchronization with the excitation signal a by the rectangular wave signal output from the comparator circuit 19. As a result, the analog switch 18 outputs a rectified DC voltage signal of the amplitude modulation signal b.

The synchronous rectifier circuit 11 is provided with an active filter 2 for receiving the output of the analog switch 18.
0, the active filter 20 removes high-frequency components including the frequency component of the excitation signal a from the output of the analog switch 18, and outputs a level proportional to the amplitude of the amplitude modulation signal b, in other words, the rotation angle θ. To generate a DC voltage signal s1 having a level proportional to the sine value sinθ of.

The circuit configuration and operation are the same for the synchronous rectifier circuit 12. The active filter 20 provided in the synchronous rectifier circuit 12 finally outputs a signal having a level proportional to the cosine value cos θ of the rotation angle θ. A DC voltage signal s2 is generated.

In this case, in this embodiment, more specifically, the DC voltage signal s1 on the side of the synchronous rectifier 11 is "KE * sin".
θ, and the DC voltage new word s2 on the side of the synchronous rectifier 12 is a voltage signal having a level of “KE · cos θ”.

The digital operation processing means 15 is constituted by a microcomputer, a personal computer, or the like including a CPU, a RAM, and a ROM (not shown), and outputs the outputs of the A / D conversion circuits 13 and 14 according to a program installed in advance. Digital signal), that is, each of the synchronous rectifiers 1
The estimated value θp of the rotation angle θ based on the digitized data of the levels of the DC voltage signals s1 and s2 output by
(Hereinafter, referred to as a rotational angle estimated value θp) and data of an estimated value of the rotational angular velocity ωp (hereinafter, referred to as an estimated rotational angular velocity ωp) are sequentially generated in a predetermined control cycle.

The digital operation processing means 15
As a functional configuration, the level value of the DC voltage signal s1 represented by the output of the A / D conversion circuit 13 (= K · E · sin θ)
Multiplying means 21 and 22 for multiplying the cosine value cos θp and the sine value sin θp of the latest value (the value obtained in the previous control cycle) of the rotation angle estimation value θp calculated by an observer described later, and A / D conversion The sine value sinθp and the cosine value cosθp of the latest value (value obtained in the previous control cycle) of the rotation angle estimation value θp are added to the level value (= KE · cosθ) of the DC voltage signal s2 represented by the output of the circuit 14. Multiplying means 23 and 24, respectively, and the value calculated by the multiplying means 21 (= K
E · sinθ · cosθp) and the value calculated by the multiplication means 23 (= K
・ E ・ cosθ ・ sinθp) difference (= KE ・ sinθ ・ cos)
θp−KE · cosθ · sinθp) is converted to the rotor 1 of the resolver 1.
a subtracting means 25 for calculating as phase difference data DSsin representing a phase difference (θ−θp) between the actual rotation angle θ and the estimated angle estimated value θp, and a calculated value (= K · E · sin) of the multiplying means 22
θ · sin θp) and the value calculated by the multiplying means 24 (= KE · cos)
DScos (= K · E · sinθ · sinθp +) with θ · cosθp
K · E · cos θ · cos θp).

Here, the value of the phase difference data DSsin obtained by the subtracting means 25 is calculated according to the addition theorem of the trigonometric function as in the following equation (1), and the sine value sin of the phase difference (θ-θp) is obtained.
It is a value proportional to (θ−θp).

[0045]

(Equation 1)

The value DS calculated by the adding means 26 is
cos is a value proportional to the cosine value cos (θ-θp) of the phase difference (θ-θp) according to the addition theorem of the trigonometric function as in the following equation (2) (hereinafter, calculated by the adding means 26). Value DS
cos is called cosine phase difference DScos).

[0047]

(Equation 2)

The multiplying means 21 and 23 and the subtracting means 25 constitute the phase difference data generating means 27 in the present invention.

The digital arithmetic processing means 15 further controls the phase difference data DSRsin (hereinafter referred to as DSRsin) by appropriately limiting the value of the phase difference data DSsin obtained by the subtraction means 25 according to the positive or negative polarity of the value of the cosine phase difference DScos. , Actual phase difference data
DSRsin), and an observer 29 (observer means) for sequentially updating the rotation angle estimation value θp and the rotation angular velocity estimation value ωp for each control cycle using the phase difference data DSRsin obtained by the limiter 28. It has. These limiter 28 and observer 2
The more detailed processing of No. 9 will be described later.

Next, the operation of the resolver conversion device 10 of this embodiment will be described together with the detailed processing of the limiter 28 and the observer 29.

When the rotor 1a of the resolver 1 is connected to a rotating body (not shown) and an excitation signal a is applied, the rotor 1a is connected via the synchronous rectification circuit 11 and the A / D conversion circuit 13.
Level (= K ·) proportional to the sine value sin θ of the rotation angle θ of
Digital data representing the level value of the DC voltage signal of (E · sin θ) is captured (sampled) by the digital arithmetic processing unit 15 every control cycle of the digital arithmetic processing unit 15. Further, the synchronous rectifier circuit 12 and A /
Through the D conversion circuit 14, digital data representing a level value of a DC voltage signal having a level (= KEECosθ) proportional to the cosine value cosθ of the rotation angle θ of the rotor 1a is subjected to the digital calculation for each control cycle. It is taken into the processing means 15 (sampled).

Then, the digital arithmetic processing means 15 sets the value of the fetched digital data K for each control cycle.
Multiplying means 21 from E · sin θ and K · E · cos θ
As described above (see equations (1) and (2)), the phase difference data DSsi
Calculate n and cosine phase difference DScos.

Further, the digital arithmetic processing means 15 uses these phase difference data DSsin and cosine phase difference DScos to calculate
The processing shown in the flowchart of FIG.
The actual phase difference data DSRsin is obtained for each control cycle.

That is, the limiter 28 first determines whether the value of the cosine phase difference DScos is positive or negative (STEP
1). When the cosine phase difference DScos is “0” or a positive value, the value of the actually used phase difference data DSRsin is replaced with the phase difference data DSsin, and the proportionality constant (= K · E) included in the value of the phase difference data DSsin. (See equation (1)) (STEP 2). That is, the value of the actually used phase difference data DSRsin is changed to the sine value sin (θ) of the phase difference (θ−θp).
−θp) itself.

Further, the cosine phase difference DS is determined in STEP1.
When cos is a negative value, the phase difference data DSs
The positive / negative polarity of the value of in (this is equal to the positive / negative polarity of cos (θ−θp)) is determined (STEP 3). At this time, DS
When sin> 0, the value of the actually used phase difference data DSRsin is set to “1” (= sin (π / 2)) (STEP 4), and D
When Ssin <0, the value of the actually used phase difference data DSRsin is set to “−1” (= sin (−π / 2)) (STEP
5).

By performing the processing of the limiter 28, the value of the actually used phase difference data DSRsin is changed to the cosine value cos (θ-θp) of the phase difference (θ-θp) as shown by the solid line in FIG.
Is positive, that is, in the range where the absolute value of the phase difference (θ−θp) is π / 2 or less (the range of 90 degrees or less),
It is set to a sine value sin (θ−θp) proportional to the phase difference data DSsin. Also, the cosine value cos (θ) of the phase difference (θ−θp)
−θp) is negative, and in the range where the absolute value of the phase difference (θ−θp) exceeds π / 2 (over 90 degrees),
The value of the actually used phase difference data DSRsin is fixedly set to “1” or “−1”.

The reason why the actual use phase difference data DSRsin is limited by the limiter 28 is as follows. That is, the observer 29, which will be described in detail later, calculates the rotation angle estimation value θp and the rotation angular velocity estimation value ωp on the assumption that the value of the actually used phase difference data DSRsin is equal to the phase difference (θ−θp). The absolute value of (θ−θp) is π / 2
In the following range, approximately θ−θp ≒ sin (θ−θp) holds approximately. For this reason, there is no problem even if a sine value sin (θ−θp) proportional to the phase difference data DSsin is used as the actually used phase difference data DSRsin. However, the phase difference (θ−θ
When the absolute value of p) exceeds π / 2, the phase difference (θ−
As the absolute value of θp) increases, the phase difference (θ-θp)
And the error between the sine value sin (θ−θp) increases.
For this reason, in the present embodiment, when the absolute value of the phase difference (θ−θp) exceeds π / 2, the actually used phase difference data DSRs
The value of in is fixed to “1” or “−1”, thereby preventing the error between the phase difference (θ−θp) and the value of the actually used phase difference data DSRsin from becoming excessive.

Using the actually used phase difference data DSRsin obtained by performing the processing of the limiter 28 on the phase difference data DSsin, the digital arithmetic processing means 15 uses the observer 29 to calculate the following equation (3) for each control cycle. By performing the processing, the rotation angle estimation value θp and the rotation angular velocity estimation value θp are calculated while being sequentially updated.

[0059]

(Equation 3)

In the expression (3), "n" is an integer (hereinafter the same) representing the number of the control cycle of the digital processing means 15, and .DELTA.t is the cycle of the control cycle. “K1” and “K2” are the estimated rotation angle θ, respectively.
p, a feedback gain related to the estimated rotational angular velocity ωp, which is a predetermined constant.

That is, assuming that the change in the rotational angular velocity of the rotating body (not shown) connected to the rotor 1a of the resolver 1 in the control cycle is sufficiently small, θp (n + 1) = θp (n) +
The relationship of Δt · ωp (n), ωp (n + 1) = ωp (n) holds. The observer 29 then calculates the rotational speed estimated value θp and the rotational angular speed estimated value θp determined for each control cycle by the phase difference (θ−θp) (this is the actual rotation of the rotational angle estimated value θp). The latest value of the actually used phase difference data DSRsin representing the error with respect to the angle θ)
Corrected according to DSRsin (n) (the value obtained by the limiter 28 in this control cycle), a new estimated rotational speed θp (n +
1) and the rotational angular velocity estimated value ωp (n + 1) are obtained.

Accordingly, the estimated rotational speed θp and the estimated rotational angular speed ωp are controlled so as to eliminate the phase difference (θ−θp) (to converge the phase difference (θ−θp) to “0”). It is determined while being updated successively for each cycle.

Here, the feedback gains “K1” and “K2” in equation (3) will be supplementarily described.

When the actually used phase difference data DSRsin is substantially equal to the actual phase difference (θ-θp), the phase difference (θ-θp), the actual rotational angular velocity ω and the estimated rotational angular velocity ωp Deviation (ω-ωp) (hereinafter, rotation angle deviation (ω
−ωp)), the above equation (3) establishes the state equation of the following equation (4).

[0065]

(Equation 4)

The root of the characteristic equation of the system represented by the equation (4) (to which reference numeral λ is attached) is given by the following equation (5).

[0067]

(Equation 5)

Therefore, the system represented by the equation (4) is stable (the phase difference (θ−θp) and the rotation angle deviation (ω−ω
The condition for p) to stably converge to “0”) is that the root given by Expression (5), that is, the pole of the system of Expression (4) exists in a unit circle on the complex plane.

In this embodiment, the values of the feedback gains K1 and K2 are set so as to satisfy the above conditions, for example, K1 = 0.2 and K2 = K1 / (4 · Δ
t).

In order to prevent the phase difference (θ-θp) and the rotation angle deviation (ω-ωp) from converging in an oscillatory manner,
Theoretically, it is preferable to place the poles of the system of equation (4) on the real axis of the complex plane.

According to the resolver converter 10 of the present embodiment described above, both the rotation angle estimation value θp and the rotation angular velocity estimation value ωp are obtained while being successively updated by the feedback processing according to the phase difference (θ−θp). Therefore, the rotation angle estimation value θp can be obtained with high accuracy while improving the quick response (followability) of the rotation angle estimation value θp to the actual change in the rotation angle θ.

FIG. 5 shows that the rotation speed of a rotating body (not shown) connected to the rotor 1a of the resolver 1 is, for example, 15000.
By setting the waveform of the output (∝sin θ) of the synchronous rectifier circuit 11 at the time of high-speed rotation at rpm and the values of the feedback gains K1 and K2 as described above, the resolver conversion device 10
An example of how the estimated rotation angle θp obtained by the above-described method changes from the waveform of the sine value sin θp is shown.

As is apparent from FIG. 5, the resolver 1
Even when the rotor 1a rotates at a high speed of 15000 rpm, the rotation angle estimation value θp accurately follows the actual rotation angle θ with high responsiveness.

In addition, the resolver device 10 of the present embodiment
Since the analog circuit is only the synchronous rectifier circuits 11 and 12,
The influence of circuit drift due to a temperature change or the like can be minimized, and the accuracy of the rotation angle estimated value θp and the rotation angular velocity estimated value ωp can be improved. In this case, although not adopted in the present embodiment, the digital arithmetic processing means 1
By the processing of 5, it is also possible to compensate for the influence of the drift of the synchronous rectifier circuits 11, 12.

Further, since the data input to the digital arithmetic processing means 15 is obtained by removing high-frequency excitation signal components by the synchronous rectifier circuits 11 and 12 (r-level signal data proportional to sin θ and cos θ), The data required for the processing by the digital processing means 15 can be captured without requiring high-speed sampling, and the digital processing means 15 can be configured at a relatively low cost. Can be easily constructed.

Further, by using the observer 29, not only the estimated rotation angle θp but also the estimated rotation angular velocity ωp can be obtained at the same time. Therefore, the processing for separately obtaining the estimated rotation angular velocity from the estimated rotation angle θp is not necessary. This is unnecessary, and the convenience of the resolver conversion device 10 can be improved.

Further, when the arithmetic processing of the observer 29 is performed, the actual use phase difference data DSRsin obtained by limiting the value of the phase difference data DSsin by the limiter 28 is used.
, The rotation angle estimation value θp and the rotation angular velocity estimation value ωp can be accurately and stably obtained by the observer 29.

[Brief description of the drawings]

FIG. 1 is a block diagram showing a configuration of a resolver conversion device according to an embodiment of the present invention.

FIG. 2 is a circuit diagram showing a configuration of a synchronous rectifier circuit of the resolver converter of FIG.

FIG. 3 is a flowchart showing processing of a limiter of the resolver conversion device of FIG. 1;

FIG. 4 is a diagram for explaining processing of a limiter of the resolver conversion device of FIG. 1;

FIG. 5 is a diagram illustrating calculation data of an estimated value of an actual rotation angle by the resolver device of FIG. 1;

FIG. 6 is a block diagram showing a configuration of a conventional resolver conversion device.

[Explanation of symbols]

DESCRIPTION OF SYMBOLS 1 ... Resolver, 1a ... Resolver rotor, 10 ... Resolver converter, 11, 12 ... Synchronous rectification circuit (DC signal generation circuit), 13, 14 ... A / D conversion circuit, 15 ... Digital arithmetic processing means, 27 ... Phase difference data generating means, 29 ... observer.

Claims (2)

[Claims] 1. Two types of amplitude modulation signals output by a resolver according to a rotation angle (θ) of a rotor when a sine wave excitation signal is applied to the resolver. And a first and second amplitude modulation signal having an amplitude level proportional to a cosine value, and a resolver conversion device for generating data representing an estimated value (θp) of the rotation angle, wherein the first and second amplitudes are First and second DC signal generation circuits for generating, from the modulation signal, DC signals having levels proportional to the sine value (sin θ) and the cosine value (cos θ) of the rotation angle, respectively; An A / D conversion circuit for A / D converting the output of the DC signal generation circuit; and an A / D conversion circuit for outputting the output of each of the first and second DC signal generation circuits.
The converted digital data is sampled in a predetermined control cycle, and based on the sampled digital data, an estimated value (θp) of the rotational angle of the rotor of the resolver and an estimated value (ωp) of the rotational angular velocity of the rotor are obtained. Digital arithmetic processing means for calculating while sequentially updating the digital arithmetic processing means, wherein the digital arithmetic processing means A / D converts the outputs of the first and second DC signal generation circuits by the A / D conversion circuit, respectively. The phase difference (θ) between the actual value of the rotation angle (θ) and the estimated value of the rotation angle (θp) is calculated from the latest value of the digital data obtained and the latest value of the estimated value of the rotation angle.
−θp), the phase difference data generating means for sequentially generating the phase difference data in the predetermined control cycle, and the estimated values (θp, ωp) of the rotation angle and the rotation angular velocity are used as calculation parameters, and the phase difference data represents A resolver conversion device comprising: an observer means for calculating the calculation parameters while sequentially updating the calculation parameters according to the phase difference data so as to eliminate the phase difference (θ−θp). 2. The phase difference data generated by the phase difference data generating means includes a sine value (sin (θ) of the phase difference (θ−θp).
-Θp)), and the digital arithmetic processing means performs the A / D conversion of the output of each of the first and second DC signal generation circuits by the A / D conversion circuit to obtain the latest value of digital data. And the latest value of the estimated rotation angle, the cosine value (cos (θ−θ) of the phase difference (θ−θp)
p)) and a cosine value of the phase difference (cos (θ−θ)
2. The resolver conversion device according to claim 1, further comprising means for restricting the value of the phase difference data input to said observer means according to the positive or negative polarity of p)).