JP5181123B2 - Analog signal digital conversion method - Google Patents

Description

The present invention relates to an analog signal digital conversion method for converting an analog input angle θ into a digital angle output φ. In particular, the excitation signal source is made independent of the negative feedback control loop of the rotation signal processing means, and the excitation signal sin ω _E t. By setting the excitation frequency ω _E t to an arbitrary fixed frequency, the influence on the output voltage and output impedance due to the change in the excitation frequency of the rotation detector can be suppressed, and dynamic angle detection accuracy and temperature characteristics are improved. It relates to a new improvement to make it possible.

As a digital conversion method of this kind of analog signal that has been conventionally used, for example, a method disclosed in Patent Document 1 is used. That is, conventionally, a rotation signal processing means including a rotation detector and an excitation signal source in a negative feedback control loop (closed loop) is used, and the phase modulation signal ω _R t + φ is input to the excitation signal source, and the excitation signal source is output. The rotation detector is excited by the excitation signal sin (ω _R t + φ).

JP 2006-17659 A

In the conventional analog signal digital conversion method as described above, since the rotation detector and the excitation signal source are included in the negative feedback control loop of the rotation signal processing means, the excitation signal sin is determined according to the rotation speed of the rotation detector. The excitation frequency ω _R t + φ of (ω _R t + φ) changes. This affects various characteristics of the rotation detector using the excitation frequency as a parameter. In other words, not only the signal frequency of the rotation detector due to the rotation speed but also the output voltage and output impedance are changed, and as a result, the dynamic angle detection accuracy and temperature characteristics are affected.
Further, since the rotation detector is excited by the excitation signal sin (ω _R t + φ), when the rotation detector and other components are physically separated by a sensor cable or the like, the frequency fluctuation or impedance change There arises a problem that the conversion is likely to become unstable due to the occurrence of a signal mismatch caused by the transmission or the transmission problem such that the signal is easily subjected to noise.
In addition, since the excitation frequency changes, a great deal of effort is required to create the excitation phase reference introduced into the synchronous detection unit, which causes an increase in manufacturing cost.

  The present invention has been made to solve the above-described problems, and its purpose is to suppress the influence of the excitation frequency on the output voltage and output impedance, and provide dynamic angle detection accuracy, temperature characteristics, etc. It is to provide a digital conversion method of an analog signal that can improve the frequency.

Digital conversion method of the analog signal according to the present invention, the excitation signal sin .omega _{E t} is the input angle first and second amplitude-modulated signal from the amplitude-modulated rotation detector by θ sinθ · sin (ω _E from the excitation signal source t−α), cos θ · sin (ω _E t−α) is input to the rotation signal processing means, and signal processing is performed to make the control deviation ε zero by the negative feedback control loop of the rotation signal processing means. In an analog signal digital conversion method for converting an input angle θ into a digital angle output φ, the excitation signal source is made independent of the negative feedback control loop of the rotation signal processing means, and the excitation frequency ω of the excitation signal sinω _E t is obtained. _E t is an arbitrary fixed frequency, and the rotation signal processing means includes the first and second amplitude modulation signals sin θ · sin (ω _E t−α), cos θ · sin (ω _E t−α), and A control deviation calculation unit that outputs the control deviation ε based on the first pseudo sine wave signals sinω _R t, cos ω _R t and the second pseudo sine wave signals sin ωt, cos ωt, an oscillation unit, a first counter, and a first has a quasi-sine wave generating unit, the phase reference based on a reference clock oscillation unit outputs omega _{R t} and the first quasi-sine wave signal sin .omega _{R t,} and phase reference generation means for generating and cos .omega _{R t,} A synchronous detection unit, a loop compensator, a signal converter, a second counter, and a second pseudo sine wave generation unit, based on the control deviation ε from the control deviation calculation unit and the frequency modulation signal ω _R t + φ and the the second quasi-sine wave signal sin .omega.t, and a negative feedback control system that generates and cos .omega.t, the frequency-modulated signal ω _{R t} + φ from the subtraction unit for calculating a phase reference omega _{R t} the digital angle output phi by subtracting the Is constituted by the signal converter is configured by an A / D converter for outputting a parallel absolute value output, the second counter has an integration function of the digital values.

Also, the first counter, the first pseudo sinusoid generator, the signal converter, said second counter, the second pseudo sinusoid generator, and as including at least digital circuits the subtraction unit, the reference clock A synchronization circuit synchronized with is used.

According to digital conversion method of the analog signals of the present invention, the excitation signal source is independent from the negative feedback control loop of the rotation signal processing means, since the excitation frequency omega _{E t} of the excitation signal sin .omega _{E t} be any fixed frequency The influence of the excitation frequency on the output voltage and output impedance of the rotation detector can be suppressed, and the dynamic angle detection accuracy and temperature characteristics can be improved. In particular, it is possible to solve the problem that the conversion is likely to become unstable due to a transmission mismatch such as occurrence of signal mismatch due to frequency fluctuation or impedance change or noise on the signal. . In addition, the excitation phase reference introduced into the synchronous detection unit can be created relatively easily, and an increase in manufacturing cost can be prevented.
Further, the excitation frequency ω _E t can be set to zero, that is, the excitation signal can be set to DC, and not only a resolver and a synchro but also an AC excitation power source is not required, such as a magnetic sensor It can also be applied to a DC analog signal in the form of a two-phase sine wave. In other words, the present invention can be applied to, for example, a magnetic sensor without greatly changing the circuit configuration, and the range of applications can be greatly expanded.
In particular, the present invention is used in a field where a resolver and a synchro, which are reliable rotation detectors, are used, particularly in an automotive field where there is a tendency of electric motive power using a brushless motor, or in a drive part address of an industrial robot. This is effective in the field of factory automation (FA), current home appliances mainly using sensorless control, and the field of humanoid robots that are expected to develop in the future.

In addition, the rotation signal processing means includes the control deviation calculation unit, the phase reference generation unit, the negative feedback control system, and the subtraction unit, so that signal processing can be performed appropriately. Thus, the input angle θ can be more reliably converted to the digital angle output φ. In particular, in the process of converting the input angle θ into the digital angle output φ, the control deviation ε and the digital speed output dφ / dt (dot φ) can be obtained, and the control deviation ε and the digital speed output dφ / dt are obtained by the control system. It can be used for abnormal diagnosis and redundant output. Also, the digital speed output dφ / dt can be directly used as the rotation speed output.
In addition, the conventional method generally uses a voltage controlled oscillator (VCO) as a signal converter. Depending on the linearity of the voltage controlled oscillator, the limit of the oscillation frequency range, etc., conversion accuracy, tracking speed, resolution, etc. Although the system performance (R / D conversion performance) is supposed to be restricted, the analog signal digital conversion method of the present invention is an A / D converter that outputs a parallel absolute value output as a signal converter. Therefore, the output of the A / D converter can be used as a real-time digital speed output, so that the signal conversion can be speeded up and the resolution can be increased. Further, since the second counter has a digital value integration function, the output of the A / D converter, that is, the digital speed output can be reliably integrated, and the second pseudo sine wave signals sinωt and cosωt to be fed back to the control deviation calculation unit The frequency modulation signal ω _R t + φ for generation can be reliably obtained from the output of the A / D converter.
Furthermore, the reference clock is a digital circuit including at least the first counter, the first pseudo sine wave generation unit, the signal converter, the second counter, the second pseudo sine wave generation unit, and the subtraction unit. Therefore, the signal conversion accuracy can be improved and the reliability can be improved.

The best mode for carrying out the present invention will be described below with reference to the drawings.
Embodiment 1 FIG.
FIG. 1 is a block diagram showing a rotation detection device according to Embodiment 1 of the present invention, and shows a rotation detection device to which the analog signal digital conversion method of the present invention is applied.
In the figure, the rotation detector 1 is a one-phase excitation / 2-phase output type amplitude modulation type brushless resolver (BRX, VRX). The rotation detector 1 includes an excitation signal sinω _E from an excitation signal source 2. t is input via the excitation signal amplifier 2a. For convenience of explanation, the amplitude of the excitation signal sin .omega _E t is 1. The excitation signal source 2 is provided independently of the negative feedback control loop of the rotation signal processing means 3 to be described later, the excitation frequency omega _{E t} of the excitation signal sin .omega _{E t} is set to any fixed frequency . In this embodiment, since a resolver is used as the rotation detector 1, the excitation frequency ω _E t is set high enough to be used as a carrier wave of an amplitude modulation signal. When is used, the excitation frequency ω _E t can be set to zero.

In the rotation detector 1, the excitation signal sin ω _E t is amplitude-modulated by the input angle θ. That is, the rotation detector 1 outputs the first and second amplitude modulation signals sin θ · sin (ω _E t−α), cos θ · sin (ω _E t−α). Here, α indicates a phase shift caused by the rotation detector 1 itself, the sensor cable, and the like. The rotation detector 1 is connected to the rotation signal processing means 3, and the first and second amplitude modulation signals sin θ · sin (ω _E t−α), cos θ · sin (ω _E t−α). Is input to the rotation signal processing means 3.

The rotation signal processing unit 3 includes a control deviation calculation unit 4, a phase reference generation unit 5, a negative feedback control system 6, and a subtraction unit 7. The control deviation calculation unit 4 includes the first and second amplitude modulation signals sin θ · sin (ω _E t−α), cos θ · sin (ω _E t−α) from the rotation detector 1 and the phase reference. The first pseudo sine wave signal sin ω _R t, cos ω _R t from the generating means 5 and the second pseudo sine wave signal sin ω t, cos ω t from the negative feedback control system 6 are input, and control is performed based on these signals. Deviation ε [sin (θ−ωt + ω _R t) · sin (ω _E t−α)] is output. The generation of the control deviation ε will be described in detail later.

The phase reference generator 5 includes an oscillation unit 50, a first counter 51, and a first pseudo sine wave generation unit 52. The oscillating unit 50 oscillates and outputs a reference clock 50a, and the first counter 51 counts the reference clock 50a and outputs a phase reference ω _R t. In other words, the phase reference ω _R t is a periodic function that is divided by the first counter 51 based on the reference clock 50a and has a constant rate of change. The first pseudo sinusoid generator 52, the phase reference ω on the basis of the _R t first pseudo sine wave signal sin .omega _R t, and outputs the cos .omega _R t. The pseudo sine wave signal is a staircase-like periodic function signal.

The negative feedback control system 6 includes a synchronous detector 60, a loop compensator 61, a signal converter 62, a second counter 63, and a second pseudo sine wave generator 64. The synchronous detector 60 performs synchronous detection of a control deviation ε [sin (θ−ωt + ω _R t) · sin (ω _E t−α)], and the loop compensator 61 is configured by the negative feedback control system 6. Loop compensation is performed to ensure the stability and high speed of the negative feedback control loop [Phase Locked Loop]. The signal converter 62 converts the loop-compensated control deviation ε into a digital signal. The second counter 63 integrates the output 62a of the signal converter 62 to generate a frequency modulation signal ω _R t + φ. The second pseudo sine wave generator 64 outputs the second pseudo sine wave signals sin ωt and cos ωt based on the frequency modulation signal ω _R t + φ.

The subtracting unit 7 calculates a digital angle output φ corresponding to the input angle θ by subtracting the phase reference ω _R t from the frequency modulation signal ω _R t + φ. That is, the rotation signal processing means 3 converts the analog input angle θ into a digital angle output φ.

  Next, FIG. 2 is a block diagram showing the control deviation calculation unit 4 of FIG. 1 in more detail. In FIG. 2, the pseudo sine wave generator is abbreviated as PSG (Pseud Sinusoid Generator). In the figure, the control deviation calculation unit 4 includes first to sixth multipliers 40 to 45, an adder 46, first and second subtractors 47 and 48, and an excitation component extraction unit 49. .

First, the a output of the rotary detector 1 first and second amplitude modulation signals _{sinθ · sin (ω E t-} α), cosθ · sin (ω E t-α) , the first to fourth multipliers 40-43.

In the first multiplier 40, the following analog operation is performed by which the first amplitude modulation signal sin θ · sin (ω _E t−α) is multiplied by the second pseudo sine wave signal sin ωt from the second pseudo sine wave generator 64. To be implemented.
sin θ · sin (ω _E t−α) × sin ωt
= Sin θ · sin (ω _E t−α) · sin ωt (1)

Similarly, the second multiplier 41 multiplies the second amplitude modulation signal cos θ · sin (ω _E t−α) by the second pseudo sine wave signal sin ωt from the second pseudo sine wave generator 64 as follows. Analog operations are performed.
cos θ · sin (ω _E t−α) × sin ωt
= Cos θ · sin (ω _E t−α) · sin ωt (2)

Similarly, the third multiplier 42 multiplies the first amplitude modulation signal sin θ · sin (ω _E t−α) by the second pseudo sine wave signal cos ωt from the second pseudo sine wave generator 64 as follows. Analog operations are performed.
sinθ · sin (ω _E t−α) × cosωt
= Sin θ · sin (ω _E t−α) · cos ωt (3)

Similarly, the fourth multiplier 43 multiplies the second amplitude modulation signal cos θ · sin (ω _E t−α) by the second pseudo sine wave signal cos ωt from the second pseudo sine wave generator 64 as follows. Analog operations are performed.
cos θ · sin (ω _E t−α) × cos ωt
= Cos θ · sin (ω _E t−α) · cos ωt (4)

Next, the adder 46 adds the output of the first multiplier 40 expressed by the above equation (1) and the output of the fourth multiplier 43 expressed by the above equation (4) as follows: To be implemented.
(1) Formula + (4) Formula = sin θ · sin (ω _E t−α) · sin ωt
+ Cos θ · sin (ω _E t−α) · cos ωt
= Cos (ωt−θ) · sin (ω _E t−α) (5)

Similarly, the first subtractor 47 subtracts the output of the third multiplier 42 expressed by the above equation (3) from the output of the second multiplier 41 expressed by the above equation (2) as follows. Is implemented.
(2) Formula- (3) Formula = cos θ · sin (ω _E t−α) · sin ωt
−sin θ · sin (ω _E t−α) · cos ωt
= Sin (ωt−θ) · sin (ω _E t−α) (6)

Next, the fifth multiplier 44 multiplies the output of the adder 46 expressed by the above equation (5) and the first pseudo sine wave signal sinω _R t from the first pseudo sine wave generator 52 as follows. The analog operation is performed.
cos (ωt−θ) · sin (ω _E t−α) × sinω _R t
= Cos (ωt−θ) · sinω _R t · sin (ω _E t−α) (7)

Similarly, the sixth multiplier 45 multiplies the output of the first subtractor 47 expressed by the equation (6) by the first pseudo sine wave signal cosω _R t from the first pseudo sine wave generator 52. The following analog operations are performed.
sin (ωt−θ) · sin (ω _E t−α) × cosω _R t
= Sin (ωt−θ) · cosω _R t · sin (ω _E t−α) (8)

Next, the second subtracter 48 subtracts the output of the sixth multiplier 45 represented by the above equation (8) from the output of the fifth multiplier 44 represented by the above equation (7). Is implemented.
(7) Formula- (8) Formula = cos (ωt−θ) · sinω _R t · sin (ω _E t−α)
-Sin (ωt-θ) · cosω _R t · sin (ω _E t-α)
= Sin (θ−ωt + ω _R t) · sin (ω _E t−α)
= Ε (9)

That is, in the control deviation calculation unit 4, the analog signal processing expressed by the above formulas (1) to (9) is performed, and the first and second amplitude modulation signals sin θ ·· that are the outputs of the rotation detector 1. sin (ω _E t−α), cos θ · sin (ω _E t−α) is converted into a control deviation ε [sin (θ−ωt + ω _R t) · sin (ω _E t−α)].

The rotation signal processing means 3 of this embodiment is controlled so that the control deviation ε is always zero. This control deviation ε is input to the synchronous detection unit 60 and is synchronously detected by the excitation phase reference ω _E t-α from the excitation component extraction unit 49. That is, the control deviation ε is converted into the following equation by making the excitation component sin (ω _E t−α) an absolute value.
E = sin (θ−ωt + ω _R t) · | sin (ω _E t−α) | (10)

Here, the excitation phase reference ω _E t-α will be considered.
The excitation phase reference ω _E t-α is a synchronization reference signal for accurately demodulating the envelope (envelope) / amplitude component sin (θ−ωt + ω _R t) of the control deviation ε by the synchronous detector 60. Is. The excitation phase reference ω _{E t-α,} by adding a phase shift alpha to the exciting signal sin .omega _{E t} of the first and second amplitude-modulated signal has, is one that is to more accurately synchronous detection.
Since the patent application has been filed separately, the details are omitted, but the excitation component extraction unit 49 uses the digital angle output φ to generate the first and second amplitude modulation signals sinθ · sin (ωt−α), cosθ. A signal having a large amplitude is selected from sin (ωt−α), and the excitation phase reference ω _E t−α is extracted from the selected signal.

Note that the control deviation E after the synchronous detection expressed by the equation (10) can be regarded as a signal equivalently expressed by the following equation.
E≈sin (θ−ωt + ω _R t) (11)
As described above, the rotation signal processing means 3 according to this embodiment is controlled so that the control deviation ε expressed by the equation (9) is always zero. Since the control deviation ε can be considered by equivalently replacing it with the control deviation E expressed by the equation (11), the following equation is established.
ε≈sin (θ−ωt + ω _R t) = 0 (12)
∴ ωt = θ + ω _R t (13)

  Next, the control deviation E after the synchronous detection expressed by the equation (11) is input to the loop compensator 61 and loop compensated by the loop compensator 61. Here, the loop compensator 61 performs general PI compensation [P (proportional) + I (integral)]. On the other hand, the control deviation E can be output to the outside as a monitor signal and used for grasping the behavior of the rotation signal processing means 3. By using the control deviation E as a monitor signal, system fail-safe such as abnormality detection and self-diagnosis can be realized.

Since the output of the loop compensator 61 is arranged in the preceding stage of the second counter 63 described later, it can be considered physically a signal having a speed weight (dimension). Here, in general, a voltage controlled oscillator (VCO) is used as the signal converter 62. The output of the loop compensator 61 is converted into a digital pulse train by a voltage controlled oscillator, and then integrated by a second counter 63 in the next stage to be converted into a frequency modulation signal ωt = ω _R t + φ. However, since the output of the voltage controlled oscillator includes a component of the phase reference ω _R t, it is somewhat difficult to directly use this digital pulse train as a speed signal.
As described above, although it is common to use a voltage controlled oscillator as the signal converter 62, in this embodiment, as the signal converter 62, an A / B that outputs a parallel absolute value output (parallel digital output). A D converter is used. Since the output of the A / D converter is a parallel absolute value output, it is more suitable for use as a speed signal than the above-described digital pulse train output.

Also, since the performance of the control system depends on the characteristics of the voltage controlled oscillator itself, such as linearity and oscillation frequency, when using the voltage controlled oscillator, the resolution and responsiveness of the system are often limited. The performance of the A / D converter, such as the resolution, the conversion accuracy, and the conversion speed, can sufficiently withstand the application to the rotation detection device of this embodiment at the present time. Furthermore, since the voltage controlled oscillator itself cannot be easily manufactured, the cost can be reduced by using an A / D converter.
However, as in the case of using the voltage controlled oscillator, in order to obtain the frequency modulation signal ωt = ω _R t + φ, it is necessary to consider the frequency offset corresponding to the phase reference ω _R t. A simple A / D converter can be used as the signal converter 62 by performing numerical integration by adding an offset amount to the A / D conversion value. That is, in this embodiment, the second counter 63 has a digital value integration function. The output of the A / D converter can be physically considered as a signal having speed weights (dimensions). That is, the output of the A / D converter can be used as the digital speed output dφ / dt (dot φ). Using an A / D converter as the signal converter 62 is very effective for the rotation detection device of this embodiment.

The frequency modulation signal ωt = ω _R t + φ is a parallel digital signal and is converted by the second pseudo sine wave generation unit 64 into a pseudo sine wave having a staircase waveform based on 8-level (8 levels) / 16 sampling. Then, it is fed back to the control deviation calculator 4. That is, the frequency modulation signal ωt = ω _R t + φ is fed back to the control deviation calculation unit 4 to constitute a series of negative feedback control loops.

Here, the use of the staircase-like pseudo sine wave is intended to simplify the analog multiplication circuit and reduce the cost. In the case of 8 values (8 levels) / 16 sampling, the harmonic component is theoretically generated in a higher range than the fifteenth order, and when the characteristics of the loop compensator 61 are taken into consideration, even if it is a pseudo sine wave, anything It can be considered that it does not affect system performance. That is, for example, when the frequency of the frequency modulation signal is 10 kHz, the harmonic component is generated at 150 kHz or more, but the band limited by the loop compensator 61 is generally 5 kHz or less, which has no effect. It is not an effect.
Note that the actual analog multiplication is realized as a simple sine wave multiplication circuit by stepwise switching the amplifier gain of one signal instead of directly using the staircase-like pseudo sine wave for multiplication. In other words, rational analog signal processing can be realized with a simple circuit, the chip area can be kept small in the case of monolithic IC, and commercialization (mass production) with high reliability, small size, and low price can be made possible.

The final output in the rotation detection device of this embodiment is a digital angle output φ obtained by converting the analog input angle θ into digital. In order to obtain this digital angle output φ, the subtraction unit 7 may subtract the phase reference ω _R t from the frequency modulation signal ω _R t + φ.

As described above, the phase reference ω _R t is divided by the first counter 51 based on the reference clock oscillated by the oscillating unit 50, and is a periodic function having a constant rate of change. The phase reference ω _R t is also input to the first pseudo sine wave generator 52, and the first pseudo sine wave signal sin ω _R t, cos ω having a staircase waveform based on 8-level (8 levels) / 16 sampling is used. It is converted to _R t. The first pseudo sine wave signals sin ω _R t and cos ω _R t are input to the fifth and sixth multipliers 44 and 45 and used for analog calculation.

  As a digital circuit including at least the first counter 51, the second counter 63, the subtractor, the first pseudo sine wave generation unit 52, the second pseudo sine wave generation unit 64, and the signal converter 62. A synchronization circuit using a single and the same reference clock 50a generated by the oscillator 50 is used.

Accordingly, the rotation detecting device of this embodiment, the excitation signal source 2 is independent from the negative feedback control loop of the rotation signal processing means 3, the excitation frequency omega _{E t} of the excitation signal sin .omega _{E t} any fixed frequency An analog signal digital conversion method is applied.

The rotation signal processing means 3 includes the first and second amplitude modulation signals sin θ · sin (ω _E t-α), cos θ · sin (ω _E t-α) and the first pseudo sine wave signal sin ω _R t. , Cos ω _R t and the second pseudo sine wave signals sin ωt, cos ωt, a control deviation calculation unit 4 that outputs the control deviation ε, an oscillation unit 50, a first counter 51, and a first pseudo sine wave generation unit 52. the a, the phase reference based on the reference clock 50a of the oscillation unit 50 outputs omega _R t and the first quasi-sine wave signal sin .omega _R t, a phase reference generation means 5 generates the cos .omega _R t, synchronous detection Unit 60, loop compensator 61, signal converter 62, second counter 63, and second pseudo sine wave generator 64, and based on the control deviation ε from the control deviation calculator 4, the frequency modulation signal ω said the _R t + _φ 2 quasi-sine wave signal sin .omega.t, a negative feedback control system 6 for generating and cos .omega.t, a subtracting unit 7 that calculates the digital angle output phi by subtracting the phase reference omega _{R t} from the frequency-modulated signal ω _{R t} + φ It is comprised by.
Further, the signal converter 62 is composed of an A / D converter that outputs a parallel absolute value output, and the second counter 63 has a digital value integration function.
Furthermore, the digital including at least the first counter 51, the first pseudo sine wave generator 52, the signal converter 62, the second counter 63, the second pseudo sine wave generator 64, and the subtractor 7. As a circuit, a synchronization circuit synchronized with the reference clock 50a is used.

It is a block diagram which shows the rotation detection apparatus by Embodiment 1 of this invention. It is the block diagram which showed the control deviation calculating part of FIG. 1 in detail.

Explanation of symbols

DESCRIPTION OF SYMBOLS 1 Rotation detector 2 Excitation signal source 3 Rotation signal processing means 4 Control deviation calculation part 5 Phase reference generation means 6 Negative feedback control system 7 Subtraction part 50 Oscillation part 50a Reference clock 51 1st counter 52 1st pseudo sine wave generation part 60 Synchronous detector 61 Loop compensator 62 Signal converter 63 Second counter 64 Second pseudo sine wave generator

Claims (2)

Excitation signal source first and second amplitude modulation signals sin [theta · sin from the rotation detector is amplitude modulated by the excitation signal sin .omega _E t from (2) the input angle _{θ (1) (ω E t} -α), cosθ Sin (ω _E t−α) is input to the rotation signal processing means (3), and signal processing is performed to make the control deviation ε zero by the negative feedback control loop of the rotation signal processing means (3). In the analog signal digital conversion method for converting the input angle θ into the digital angle output φ,
Said said is independent from the negative feedback control loop of the excitation signal source (2) the rotation signal processing means (3), the excitation frequency omega _{E t} of the excitation signal sin .omega _{E t} is an arbitrary constant frequency,
The rotation signal processing means (3)
The first and second amplitude modulation signals sin θ · sin (ω _E t-α), cos θ · sin (ω _E t-α), the first pseudo sine wave signals sin ω _R t, cos ω _R t, and the second pseudo sine wave. A control deviation calculator (4) for outputting the control deviation ε based on the signals sin ωt and cos ωt;
It has an oscillation unit (50), a first counter (51), and a first pseudo sine wave generation unit (52), and is based on a phase reference ω _R t based on a reference clock (50a) output from the oscillation unit (50). and said first pseudo sine wave signal and sin .omega _R t, the phase reference generator means for generating and cosω _R t (5),
A synchronous detector (60); a loop compensator (61); a signal converter (62); a second counter (63); and a second pseudo sine wave generator (64). A negative feedback control system (6) for generating a frequency modulation signal ω _R t + φ and the second pseudo sine wave signals sin ωt, cos ωt based on the control deviation ε from
A subtractor (7) for calculating the digital angle output φ by subtracting the phase reference ω _R t from the frequency modulation signal ω _R t + φ ;
It consists of
The signal converter (62) includes an A / D converter that outputs a parallel absolute value output, and the second counter (63) has a digital value integration function. A method for digital conversion of analog signals. The first counter (51), the first pseudo sine wave generator (52), the signal converter (62), the second counter (63), the second pseudo sine wave generator (64), and the a digital circuit including subtraction unit (7) at least, a digital conversion method according to claim 1 analog signal, wherein the uses synchronization circuit which is synchronized by the reference clock (50a).

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