JP2011243255A - Optical disk drive - Google Patents

Abstract

PROBLEM TO BE SOLVED: To realize separation between a reproduction signal and a high frequency component that hinder a high-speed reproduction of an optical disk.SOLUTION: An optical disk drive comprises means (11) for converting a pulse reproduction signal into a temporal discretized signal at a channel clock frequency, means (71) for suppressing a bright-line spectrum resulted from a pulse drive light-emission from a laser source in the temporal discretized signal, and means (72) for suppressing a high frequency component of the reproduction signal in the temporal discretized signal.

Description

  The present invention relates to an optical disk drive reproduction technique, and more particularly to an optical disk drive excellent in high-speed reproduction.

  The scope of application of the present invention is not limited to Blu-ray Disc (BD), but in the following description, BD is assumed, and the terminology is based on what is used in BD.

  Many of the current optical disk devices including BD employ a high-frequency superposition method in order to suppress noise generated by a laser diode used as the light source. Since this technique is disclosed in Non-Patent Document 1 and is well known to those skilled in the art, only matters necessary for the present invention will be described below, and further details will not be described in detail.

  When the laser beam reflected by the disk enters the oscillating laser diode, the oscillation state becomes unstable, resulting in significant laser noise. In order to avoid this, a high-frequency superposition method is used. This is called a high-frequency superposition method because a laser emits pulses by superimposing a high-frequency signal on a laser diode drive signal. A schematic diagram of the light emission waveform at that time is shown in FIG. That is, light emission and quenching are repeated alternately. Here, the ratio (duty ratio) between the interval between laser pulses (modulation period) and the light emission period with respect to the interval is a parameter adjusted so that laser noise is minimized. That is, the frequency and the duty ratio are selected so that the laser pulse reflected from the disk does not enter the laser diode during laser oscillation.

  Since the emission waveform of the laser has a shape as shown in FIG. 2, assuming that there is no band limitation due to the reproduction photodiode and the current-voltage conversion amplifier, the reproduction signal waveform has the shape as shown in FIG. become. Hereinafter, a signal including such a reproduction pulse train is referred to as a pulse reproduction signal. Here, the broken line in FIG. 3 is a reproduced signal waveform obtained when the laser is continuously oscillated with the same output as the peak of the laser pulse at the time of high frequency superposition. That is, the shape of the upper envelope of the pulse reproduction signal is a reproduction waveform by continuous light. Therefore, a desired reproduction waveform can be obtained by passing the pulse reproduction signal through an envelope detection, that is, a low-pass filter having a cutoff frequency sufficiently lower than the frequency of the superimposed high-frequency current. In most of the current optical disk apparatuses, this is realized by the band limitation of the system including the photodetector and the current-voltage conversion amplifier and the analog equalizer.

  An example of the spectrum of the pulse reproduction signal is shown in FIG. The reproduction speed is 8 times, the frequency of the superimposed high frequency signal is 400 MHz, and the duty ratio of the pulse is 0.2. A component at 0 to 132 MHz is a reproduction signal. In addition, since the pulsed reproduction signal is a kind of amplitude modulation, the bright line spectrum of the superimposed high-frequency signal and the reproduction signal component modulated in the vicinity thereof are observed. As shown in FIG. 4, the spectrum of the pulse reproduction signal is composed of a reproduction signal component and a harmonic component of the reproduction signal accompanied by the bright line spectrum of the high frequency superimposed signal. Hereinafter, the high-frequency superimposed signal will be referred to as a carrier in accordance with the custom in modulation / demodulation technology. Unless otherwise specified, a combination of the reproduction signal harmonic component and the carrier harmonic is simply referred to as a harmonic component.

  The appropriate carrier frequency is determined solely by the optical path length of the pickup. In an evaluation machine that conforms to the BD standard, it is 400 MHz. However, a higher frequency can be used if the optical path length of the pickup is designed to be sufficiently short.

  When the playback speed is increased, the playback signal band naturally increases. In the case of BD, the upper limit of the reproduction signal band is almost the same as the repetition signal frequency (hereinafter referred to as 2T frequency) of 2T mark-2T space (T: channel clock period), which is 1/4 of the channel clock frequency. Since the channel clock frequency of BD 1 × speed is 66 MHz, the 2T frequency reaches 16.5 MHz at 1 × speed, 132 MHz at 8 × speed, and 165 MHz at 10 × speed. That is, if the speed is 8 times or more, the upper limit of the reproduction signal band and the carrier frequency are considerably close. As a result, it becomes extremely difficult to sufficiently suppress the carrier emission line spectrum due to the band limitation of the system including the photodetector and the current-voltage conversion amplifier and the analog equalizer.

  As a technique for solving this difficulty, there is a technique for sampling the apex of each pulse of a pulse reproduction signal as disclosed in Patent Documents 1 and 2. In order to implement this method, there is a mounting difficulty that the operation band of the photodetector and the current-voltage conversion amplifier is required several times that of the conventional method.

  Further, as another technique for facilitating separation of a reproduction signal and a carrier, there is a method of frequency-converting the bright line spectrum of a carrier into an attenuation region of an adaptive equalizer as disclosed in Patent Document 3. Here, Patent Document 3 is incorporated into the present application.

The main points of this technique will be described below with reference to FIG. FIG. 5 shows the conditions of an appropriate carrier frequency for solving the above-mentioned problem. Here, f _HF , f _clk , and f _2T represent a carrier frequency, a channel clock frequency, and a 2T frequency, respectively. Here, the 2T frequency is regarded as the upper end of the reproduction signal band. Thereafter, the 2T frequency is regarded as the upper end of the reproduction signal band unless otherwise required. In FIG. 5, 101 represents a reproduction signal band, and 102 schematically represents the amplitude transfer characteristic of the adaptive equalizer. Also, in FIG. 5, the carrier bright line spectrum 103 and the bright line spectrum 104 of the alias signal generated by sampling are both schematically shown as a half line with an arrow. As shown in FIG. 5, the above problem can be solved by setting the carrier frequency within the range specified by (Equation 1).

At this time, it is assumed that the configuration of the signal processing system includes a digital low-pass filter or a digital equalizer having a low-pass characteristic. The adaptive equalizer corresponds to this. An example of such a signal processing system is shown in FIG. FIG. 6 shows only the configuration of the signal processing system after AD conversion of the reproduction signal in the optical disk drive, and other parts are omitted. The input signal is AD converted by the AD converter 11, equalized by the adaptive equalizer 32, and decoded by the Viterbi decoder 53. The adaptive equalizer is a FIR (finite impulse response) filter. The decoding result is input to an LMS (least mean square) controller 54. An adaptive equalizer output is also input to the LMS controller. The LMS controller compares the decoding result with the target waveform obtained from the PR (partial response) class of the Viterbi decoder and the output of the adaptive equalizer according to the LMS algorithm, so that the difference between the two is minimized. Increase or decrease each tap coefficient. Since the LMS algorithm and the adaptive equalizer using the LMS algorithm are techniques widely known among those skilled in the art, further explanation is omitted.

  The output of the adaptive equalizer is also input to the phase comparator 22. The phase comparator compares the driving clock of the signal processing system with the phase of the reproduction signal. The result is smoothed by the loop filter 23 and input to the DA converter 12. The DA converter output is input to the voltage controlled oscillator 24 as a frequency control signal. The voltage controlled oscillator oscillates at a frequency according to the instruction of the frequency control signal. The output of the voltage controlled oscillator is used as a drive clock for the AD converter. Although not shown, it is also used as a drive clock for each digital element in FIG. The above closed loop constitutes a PLL (phase locked loop), and the clock of the input signal and the drive clock of the signal processing system can be synchronized.

  It is natural that the carrier frequency satisfies the laser noise suppression condition (determined by the optical distance from the laser diode in the pickup to the disk) by the high frequency superimposition method.

If the carrier frequency is set in the range specified by Equation 1, it is aliased because it is higher than half the channel clock frequency. Briefly explaining this, since the carrier frequency signal is a sine wave, its phase increases at 2πf _HF t with time t, whereas this increases with the channel clock frequency (phase increases with 2πf _clk t). Since sampling is performed, the phase at which the carrier is sampled changes by 2π (f _clk -f _HF ) t. That is, the data after sampling is equivalent to a sine wave having a frequency f _clk -f _HF . Therefore, when the carrier frequency is set within the range specified by Equation 1, the carrier frequency is converted into the frequency range represented by Equation 2 by sampling. If the alias frequency of the carrier is f _aliased ,

Here, as described above, the transfer characteristic of the adaptive equalizer is a low-pass filter that passes the reproduction signal. In other words, the alias of the carrier signal comes to the stop band of the adaptive equalizer. Therefore, the alias of the carrier signal is attenuated by the adaptive equalizer. As a result, it is possible to compensate for carrier attenuation, which is insufficient with the low-pass characteristics of the photodetector and the analog equalizer.

  That is, the carrier can be attenuated by the adaptive equalizer by actively converting the carrier frequency to the stop band of the adaptive equalizer by actively using aliasing.

  The reason why the phase comparator is placed after the adaptive equalizer in FIG. 6 is to attenuate the carrier before performing the phase comparison. The adaptive equalizer equalizes the reproduced signal so as to approach the waveform required by the Viterbi decoder. In this case, since the carrier is an unnecessary component, the adaptive equalizer acts in a direction to suppress this.

JP2007-073147 (corresponding US2007 / 0053262) JP 2008-217913 (corresponding US2008 / 0219132) JP 2009-187593 (corresponding US2009 / 0196156)

Arimoto Akira, et al. “Reduction of laser noise in video disc players equipped with semiconductor lasers by high-frequency current superposition method”, Optics Vol.14, No.5, paragraphs 377-383 Floyd M. Gardner, “Interpolation in Digital Modems-Part I: Fundamentals”, IEEE Transactions on Communications, Vol. 41, pp. 501-507 (1993). Lars Erup, “Interpolation in Digital Modems-Part II: Implementation and Performance”, IEEE Transactions on Communications, Vol. 41, pp.998-1008 (1993).

  The problem to be solved by the present invention is to realize separation of a reproduction signal and a harmonic component that hinder high-speed reproduction of an optical disc as described in the background section.

  Among the techniques described in the background art section, the technique disclosed in Patent Document 3 focuses on suppressing carrier aliases (hereinafter simply referred to as aliases). Actually, since the amplitude of the alias overwhelms the amplitude of the reproduction signal, it is essential to suppress this in order to realize high-speed reproduction. However, in the technique disclosed in Patent Document 3, no consideration is given to the harmonic component of the reproduction signal accompanying the emission line spectrum of the carrier. As can be seen from FIG. 4, the harmonic spectrum of the reproduction signal is accompanied around the bright line spectrum of the carrier. If these are also not attenuated sufficiently and are input to the AD converter, components higher than the Nyquist frequency (1/2 of the sampling frequency) are frequency-converted in the same manner as the carrier. If the converted previous frequency is within the band of the reproduction signal (baseband), the reproduction result is significantly adversely affected. Even if the converted frequency is not within the band of the reproduction signal, the reproduction performance is greatly deteriorated because it is the same as the addition of noise.

  Also, when reproducing in the CAV mode, since the channel clock frequency differs depending on the radius of the disc being reproduced, in the technique disclosed in Patent Document 3, the alias frequency differs depending on the radius of reproduction. For this reason, there has been annoyance that it is necessary to change the carrier frequency depending on the radius to be reproduced.

In order to solve the above-described problems, in the present invention, conversion means for converting a pulse reproduction signal into a temporally discretized signal at a channel clock frequency, and a bright line derived from pulse-driven emission of a laser light source in the temporally discretized signal. An optical disc drive having means for suppressing a spectrum and means for suppressing a harmonic component of a reproduction signal in the temporally discretized signal is provided.

As described above, by providing means for suppressing the emission line spectrum and means for suppressing the harmonic component of the reproduction signal in the temporally discretized signal, separation of the reproduction signal and the harmonic component can be realized. Thus, an optical disk drive capable of high speed reproduction can be provided.

The figure which shows an example which implemented this invention. The figure which shows an example of the pulse light emission waveform of a laser. The model explanatory drawing of a pulse reproduction signal waveform. An example of the spectrum of a pulse reproduction signal Explanatory drawing of the method of suppressing a carrier using frequency conversion at the time of AD conversion and an adaptive equalizer The figure explaining an example of a structure of the reproduction | regeneration system of an optical disk drive. (a) An example of an analog equalizer output spectrum, (b) An example of a spectrum after AD conversion (a) Example of amplitude characteristics of notch filter, (b) Example of spectrum of notch filter output, (c) Example of eye pattern of notch filter output (a) Some examples of pass amplitude characteristics of a filter that suppresses harmonics of the reproduction signal, (b) PLL operation evaluation results when the filter of the design shown in (a) is applied Spectrum and eye pattern of low-pass filter output when design value with cutoff frequency of 0.28fs is applied Simplified configuration diagram of an example configuration of an optical disk drive Example of calculation result of minimum and maximum optical path length according to reproduction speed The figure which shows another example which implemented this invention.

  One embodiment of the present invention is shown in FIG. In the system shown here, the output of the voltage controlled oscillator 24 is used as a drive clock for digital signal processing elements including the AD converter 11, that is, a channel clock. Further, the channel clock is frequency-divided by a half by the frequency divider 80 and used as a drive clock for the laser driver 14. The laser driver 14 generates a pulse current signal that drives the laser diode 6 at the input clock frequency. That is, the carrier frequency is half the channel clock frequency. At this time, the laser driver 14 generates a laser driving current having a waveform such that a desired average laser power and duty ratio can be obtained. The output light intensity of the laser diode changes with time as shown in FIG. The pulse shape does not necessarily have to be as shown in the example of FIG. 2, and may be, for example, a rectangle.

  The laser light emitted from the laser diode is converted into parallel light by the collimator lens 5, passes through the polarization beam splitter 4 and the quarter wavelength plate 3, and then focused on the recording film surface of the optical disk 1 by the objective lens 2. . The laser beam is reflected on the recording film surface. At this time, since the reflectance varies depending on whether the place irradiated with each pulse is a recording mark or a space, a reflected pulse laser array in which the pulse intensity is modulated accordingly. The temporal change in the laser light intensity is similar to that of the pulse reproduction signal, as schematically shown in FIG. When the reflected pulse laser array returns to the polarization beam splitter 4 through the original path, the reflected pulse laser string is reflected and collected on the photodiode 8 by the focusing lens 7 and converted into a current signal. The current signal is converted into a voltage signal by the current amplifier 10, equalized by the analog equalizer 9, and then converted into a digital signal by the AD converter 11. At this time, the higher-order harmonics are attenuated due to the band limitation of the photodiode and the analog equalizer. The AD converter 11 uses the output of the voltage controlled oscillator VCO 24 as a drive clock, and converts the pulse reproduction signal into a temporal discretization signal. Here, the feature of this embodiment is that the pulse interval of the pulse reproduction signal is twice the channel clock period.

  Now, since the carrier frequency is exactly half the channel clock frequency, that is, the Nyquist frequency, the carrier frequency remains unchanged even after AD conversion. Further, the harmonics of the reproduction signal located in a frequency band higher than the Nyquist frequency are frequency-converted to baseband by undersampling. This situation will be described in detail later, and the description of the operation of the system will be continued.

  The output of the AD converter 11 is input to a notch filter 71 having an attenuation pole at the Nyquist frequency, and the carrier is removed. The output of the notch filter is input to the low pass filter 72. This low-pass filter has a characteristic of allowing a reproduction signal to pass and suppressing harmonics of the reproduction signal.

  The output of the low-pass filter 72 is input to the phase comparator 22. The phase comparator compares the phase of the input signal with the channel clock and outputs the difference. The phase comparison result is smoothed by the loop filter 23 and then converted into a voltage signal by the DA converter 12. This DA converter output is used as a frequency control signal of the voltage controlled oscillator. That is, the AD converter, the notch filter, the low-pass filter, the phase comparator, the loop filter, the DA converter, and the voltage-controlled oscillator constitute a closed loop. This closed loop operates as a PLL. The operation of the PLL is widely known and will not be described in further detail here.

  The output of the low-pass filter 72 is also input to the adaptive equalizer 32 and is input to the Viterbi decoder 53 after undergoing adaptive equalization. The Viterbi decoder decodes a bit string from the input signal. Note that the input signal to the adaptive equalizer may be the output of the notch filter 71.

  Next, the state of the input signal in the signal processing process will be described based on the simulation result. In the signal processing simulation, a pseudo reproduction signal synthesized based on the optical simulation was used. The length of the signal is 1 RUB (recording unit block), which is a 25 GB BD format. The reproduction signal amplitude ratio was superimposed with pseudo disk noise of -19.4 dB and white noise of -29.4 dB. The oversample ratio of the evaluation waveform was 16, and the high-frequency superimposed wave was a rectangular wave with a duty ratio of 1/4. The band-limiting characteristics of the photodiode are assumed to be a second-order Butterworth low-pass filter with a cutoff frequency of 0.32 fs. The band limiting characteristic of the analog equalizer is assumed to be a fifth order Butterworth low-pass filter with a cutoff frequency of 0.64 fs. Here, fs is an AD conversion frequency, that is, a channel clock frequency.

  FIG. 7 (a) shows the spectrum of the analog equalizer output. The horizontal axis of the graph is displayed at a frequency normalized by the channel clock frequency fs. Since the carrier frequency is 0.5 fs, carrier harmonics are recognized at a frequency that is an integral multiple of 0.5 fs. In addition, the harmonics of the reproduction signal are associated with the upper and lower sidebands of the carrier and its harmonics. Each harmonic is band-limited by a photodiode and an analog equalizer except for a part of the first harmonic, but the carrier and the emission line amplitude of the harmonic are overwhelmingly larger than the reproduction signal amplitude. . Therefore, it can be easily understood that the reproduction system used in the current optical disc drive cannot normally process a signal having a spectrum as shown in FIG.

  FIG. 7B shows a spectrum after AD conversion of the analog equalizer output. As described above, since the carrier frequency matches the Nyquist frequency, the carrier emission line spectrum can be seen at a frequency of 0.5 fs. In addition, the result of frequency conversion of the higher harmonics of the carrier is also converted into a bright line spectrum at a frequency of 0.5 fs. Further, the rise of the spectrum between the frequencies 0.25 fs and 0.5 fs is the first harmonic component of the reproduction signal. Here, the high-order harmonic component of the reproduction signal and the upper sideband component of the first-order harmonic are also frequency-converted and added. As described above, the reproduction system used in the current optical disk drive cannot perform normal processing.

  The notch filter used has 2 taps and its coefficient is 0.5. FIG. 8A shows the pass characteristic of the notch filter. There is an attenuation pole at a frequency of 0.5 fs. FIG. 8B shows the spectrum of the notch filter output. It can be seen that the bright line spectrum is removed by the notch filter. FIG. 8C shows an eye pattern of the notch filter output. It is clearly different from the normal BD eye pattern. The PLL cannot operate normally only by removing the emission line spectrum of the carrier.

  A person skilled in the art can easily understand that the form of the notch filter is not limited to the above. However, among these, an FIR filter having an even number of tap coefficients can be easily configured because the frequency 0.5 fs becomes a stop band. In particular, when the tap coefficients are symmetric, the frequency 0.5 fs becomes an attenuation pole, so that a large attenuation can be obtained with certainty. When the number of taps is 4 or more, it is possible to avoid attenuation in the reproduction signal band as shown in FIG. Even an FIR filter with an odd number of tap coefficients can be used as long as the attenuation at a frequency of 0.5 fs can be increased. For example, when the coefficients of the FIR filter are 0.5, 0.5, and -0.05 in order, an attenuation of 25 dB or more can be expected at a frequency of 0.5 fs.

  The reason why the signal quality of the notch filter output is low is that the harmonics of the reproduction signal remain as described above. Therefore, it is necessary to attenuate the remaining harmonics of the reproduction signal before being input to the phase comparator. The low-pass filter 72 is responsible for this. This low-pass filter is required to pass the reproduction signal and suppress harmonics of the reproduction signal. Such a filter can be realized using, for example, an FIR filter.

An actual design example is shown below. The type of filter was an equiripple filter, and the number of taps was fixed at 9. The frequency at the upper end of the passband (attenuation 0 dB) was 0.25 fs. This is the frequency of the repetitive pattern of 2T mark and 2T space, and can be regarded as the maximum frequency component included in the BD playback signal. A plurality of filters (5 types of 0.28, 0.30, 0.32, 0.34, and 0.36 fs) were designed using the cutoff frequency as a parameter. FIG. 9 (a) shows their pass amplitude characteristics. The numbers in the legend represent the cutoff frequency (attenuation 3 dB) normalized by fs. When these were applied as the low-pass filter 72, stable operation of the PLL could be realized in all design examples. FIG. 9B summarizes the attenuation amount of these stop bands and the evaluation results of the PLL when applied as the low-pass filter 72. For the PLL operation evaluation, the rms (root mean square) value of the output phase error signal is used in the phase comparator output diagram. Fig. 9 (b) shows the case of processing a pseudo-waveform in which the rms value of the phase error signal between 8 x 10 ⁵ bits and 9 x 10 ⁵ bits measured from the beginning of the signal is not superposed with high frequency. The value normalized by the value of is shown.

  When the cut-off frequency is 0.28 fs, the phase error of the PLL is 1.44 times larger than when no high frequency is superimposed. This is thought to be because the attenuation in the stopband of the residual harmonic suppression filter is as small as 7.0 dB. Increasing the stopband attenuation to 14.8 dB reduces the PLL phase error to 1.05 times that without high-frequency superposition. It can be said that this can suppress the influence of the residual harmonics. However, even if the attenuation is further increased, the phase error increases again. As can be seen from FIG. 9 (a), the cutoff frequency has to be separated from the upper end of the pass band in order to increase the attenuation due to the filter design limitation. This is thought to be due to the fact that the attenuation between the frequencies is smaller.

  FIG. 10A shows the spectrum of the low-pass filter output when a design value with a cutoff frequency of 0.28 fs is applied. FIG. 10B shows an eye pattern of the low-pass filter output in that case. Compared with the eye pattern of the notch filter output in FIG.

  In the example shown in FIG. 1, the laser driver sets a pulse duty ratio and a pulse amplitude in advance, and supplies a pulsed light emission clock thereto to obtain a desired laser driving current signal. In addition to the method using such a laser driver, for example, a laser driver mainly having only a signal amplification function and no waveform generation function is used, and a laser driving waveform is generated on the system LSI side. It is also possible to use a method of amplifying this waveform as necessary.

  In the above example, the attenuation pole frequency of the notch filter and the alias frequency of the carrier are both 0.5 fs. Since the highest frequency component of the BD playback signal is 0.25 fs, it is difficult to consider other options as long as BD playback is considered. However, this is not the case when the performance of the optical system is equivalent to that of BD and recording is performed at a higher linear recording density than BD. For example, when the shortest recording mark and space is 2T and the length is about 120 nm, the highest frequency component of the reproduction signal is approximately 0.2 fs. In this case, for example, the same effect can be obtained by designing the attenuation pole frequency of the notch filter to 0.4 fs and setting the carrier frequency to 0.6 fs.

  The above contents are established at a standardized frequency normalized by the channel clock frequency. The actual reproducible reproduction speed depends on the optical path length of the pickup. The range of the carrier frequency and pulse duty ratio that can avoid the return optical noise of the semiconductor laser is determined by the optical path length, that is, the optical distance between the light source and the disk. In this method, since the carrier frequency is fixed to 1/2 of the channel clock frequency, the range of reproduction speed that can be actually used is determined by the optical path length of the pickup.

  Generally, CAV (constant angular velocity) mode is applied to high-speed playback. In CAV mode, the playback speed differs by about 2.4 times between the inner circumference and the outermost circumference. In other words, when this method is applied to the entire area, the carrier frequency must also change by the same amount.

  If the pulse duty ratio is d, the carrier frequency is f, and the speed of light is vc, the one-way optical path length L that can avoid return optical noise is given by the following (Equation 3).

FIG. 11 shows a plot of the maximum and minimum optical path lengths of equation (1) for each reproduction speed between 6 and 16 times speed with a pulse duty ratio of 1/5. The higher the reproduction speed, the smaller the absolute value and range of the allowable optical path length. From FIG. 11, it can be seen that if the optical path length is set from 151 to 227 mm, it is possible to cover from this speed to 6 times speed to 16 times speed. This is almost equivalent to the speed range for 16x CAV playback (16x at the outermost circumference and 6.7x at the innermost circumference).

  FIG. 12 is a diagram showing a schematic configuration when the present invention is applied to an optical disk drive. However, elements that are not necessary for the following description are omitted. The pickup 201 includes an optical system necessary for reproduction and recording, a semiconductor laser, a laser driver, a photoelectric conversion system, and the like. On the main board 202, a system LSI 203, a power supply, a spindle driver, and the like on which many functions necessary for the optical disk drive are integrated are mounted. The firmware 204 is software for controlling the operation of the optical disk drive, and is built in a memory built in the system LSI. The system LSI also incorporates elements after the analog equalizer 9 in the configuration diagram shown in FIG.

  By doing so, it is possible to separate the reproduction signal and the harmonic component that hinder high-speed reproduction of the optical disk, and this is particularly effective at high-speed reproduction of 12 times or more.

FIG. 13 shows another example in which the present invention is implemented. This example is characterized in that an interpolator type PLL is used as a reproduction type PLL. The carrier clock is generated by an oscillator 90. The oscillator is independent of the channel clock. However, the carrier frequency is controlled by the firmware so as to be about 5% higher than the channel clock. In FIG. 13, the notch filter and the AD converter use the output of the oscillator 90 as a clock. On the other hand, each element arranged on the right side of the interpolator uses the output of the numerically controlled oscillator 91 as a clock. The output of the numerically controlled oscillator 91 is a channel clock.

  The output clock of the oscillator is frequency-divided by ½ by the frequency divider 80 and used as a drive clock for the laser driver 14. The laser driver 14 generates a pulse current signal that drives the laser diode 6 at the input clock frequency. That is, the carrier frequency is ½ of the oscillation frequency of the oscillator. At this time, the laser driver 14 generates a laser driving current having a waveform such that a desired average laser power and duty ratio can be obtained. The output light intensity of the laser diode changes with time as shown in FIG. The pulse shape does not necessarily have to be as shown in the example of FIG. 2, and may be, for example, a rectangle.

  The laser light emitted from the laser diode is converted into parallel light by the collimator lens 5, passes through the polarization beam splitter 4 and the quarter wavelength plate 3, and then focused on the recording film surface of the optical disk 1 by the objective lens 2. . The laser beam is reflected on the recording film surface. At this time, since the reflectance varies depending on whether the place irradiated with each pulse is a recording mark or a space, a reflected pulse laser array in which the pulse intensity is modulated accordingly. The time change of the laser beam intensity coincides with the pulse reproduction signal shown in FIG. When the reflected pulse laser array returns to the polarization beam splitter 4 through the original path, the reflected pulse laser string is reflected and collected on the photodiode 8 by the focusing lens 7 and converted into a current signal. The current signal is converted into a voltage signal by the current amplifier 10, equalized by the analog equalizer 9, and then converted into a digital signal by the AD converter 11. At this time, the higher-order harmonics are attenuated due to the band limitation of the photodiode and the analog equalizer.

  In this example, the carrier frequency is approximately 5% higher than half the channel clock frequency. Here, when setting the carrier frequency higher than half the channel clock frequency, the degree is set to 5%. However, it is generally sufficient if the carrier frequency is 1/2 or more of the channel clock frequency. The analog equalizer output is sampled at a frequency that is exactly twice the carrier frequency as in the previous embodiment. Accordingly, since the carrier frequency is the Nyquist frequency for AD conversion, as in the example of FIG. 1, the emission line spectrum of the carrier and its alias can still be suppressed by the notch filter operating with the oscillator output as the clock.

  The output of the AD converter 11 is input to a notch filter 71 having an attenuation pole at the Nyquist frequency, and the carrier is removed. The output of the notch filter is input to the interpolator 92. A numerically controlled oscillator output, that is, a channel clock is also input to the interpolator. This interpolator converts an input sampled pulse reproduction signal into a signal resampled at a frequency and phase indicated by another input clock. Such an interpolator is widely known and can change the sampling rate of an input signal in sufficiently small steps. Specific configuration examples are disclosed in, for example, Non-Patent Documents 2 and 3, and will not be described in detail here. Since the output signal of the interpolator and the channel clock are not synchronized as they are, they are synchronized using a PLL.

  The output of the interpolator is input to the low pass filter 72. This low-pass filter has the characteristic of passing the reproduced signal and suppressing the harmonics of the reproduced signal, as in the case of the first embodiment.

  The output of the low-pass filter 72 is input to the phase comparator 22. The phase comparator compares the phase of the input signal with the channel clock and outputs the difference. The phase comparison result is smoothed by the loop filter 23 and then input to the numerically controlled oscillator 91. The numerically controlled oscillator oscillates at a frequency corresponding to the input signal. That is, the interpolator, the low-pass filter, the phase comparator, the loop filter, and the numerically controlled oscillator constitute a closed loop. This closed loop operates as a PLL. The operation of the PLL is widely known and will not be described in further detail here.

  The output of the low-pass filter 72 is also input to the adaptive equalizer 32 and is input to the Viterbi decoder 53 after undergoing adaptive equalization. The Viterbi decoder decodes a bit string from the input signal.

  In the example of FIG. 13, since the carrier frequency and the channel clock are independent, it is necessary to change the oscillation frequency of the oscillator in accordance with the reproduction speed. This can be easily implemented by firmware or hardware. When reproducing in the CAV mode, it is necessary to change the oscillation frequency of the oscillator according to the reproduction radius. This can also be easily implemented by firmware or hardware. The carrier frequency may be approximately 5% higher than the channel clock frequency, and high accuracy is not necessary. For example, it is sufficient if the approximate channel clock frequency is calculated from the address and rotation speed of the block to be reproduced and multiplied by 1.05.

  In the above embodiment, the description has been made using notch filters and low-pass filters for convenience. However, the present invention is not limited thereto, and a device having frequency characteristics that suppress the harmonics of the bright line spectrum and the reproduction signal may be used. In the above embodiment, the low-pass filter 72 is disposed after the notch filter, but this order may be reversed.

1: optical disk, 2: objective lens, 3: 1/4 wavelength plate, 4: polarizing beam splitter, 5: collimator lens, 6: laser diode, 7: focusing lens, 8: photodiode, 9: analog equalizer, 11: AD converter, 14: laser driver, 22: phase comparator, 24: voltage controlled oscillator, 32: adaptive equalizer, 53: Viterbi decoder, 54: LMS controller, 71: notch filter, 72: low Band pass filter, 80: 2 frequency divider, 90: oscillator, 91: numerically controlled oscillator, 92: interpolator,
101: reproduction signal band, 102: adaptive equalizer amplitude transfer characteristic, 103: bright line spectrum, 104: carrier alias signal bright line spectrum,
201: Pickup, 202: Main board, 203: System LSI, 204: Firmware

Claims (8)

A laser light source;
A light source driving unit that drives the laser light source at a frequency based on a channel clock frequency; and
An optical system for irradiating an optical disc with laser light generated from the laser light source;
A photodetector for receiving the laser beam reflected from the optical disc;
Means for converting the output of the photodetector into an electrical pulse regeneration signal;
Means for converting the pulse reproduction signal into a temporally discrete signal at a channel clock frequency;
Means for suppressing an emission line spectrum derived from pulse-driven emission of the laser light source in the temporal discretization signal;
Means for suppressing harmonic components of the reproduction signal in the temporally discretized signal. A laser light source;
A light source driving unit that drives the laser light source at a frequency of 1/2 or more of a channel clock frequency;
An optical system for irradiating an optical disc with laser light generated from the laser light source;
A photodetector for receiving the laser beam reflected from the optical disc;
Means for converting the output of the photodetector into an electrical pulse regeneration signal;
Means for converting the pulse reproduction signal into a temporally discretized signal at a frequency twice the pulse driving frequency;
Means for suppressing an emission line spectrum derived from pulse-driven emission of the laser light source in the temporal discretization signal;
Means for suppressing harmonic components of the reproduction signal in the temporally discretized signal. In the optical disk drive according to claim 1,
An optical disc drive characterized in that a pulse driving frequency of the laser light source is ½ of the channel clock frequency. In the optical disk drive according to claim 3,
An optical disk drive characterized in that the means for suppressing the emission line spectrum is a notch filter having an attenuation pole at a frequency of 0.5 fs. In the optical disk drive according to claim 1 or 2,
An optical disc drive characterized in that the means for suppressing the emission line spectrum derived from the pulse-driven emission of the laser light source in the temporally discretized signal is an FIR filter having an even number of tap coefficients. In the optical disk drive according to claim 1 or 2,
An optical disc drive characterized in that the means for suppressing the emission line spectrum derived from the pulsed light emission of the laser light source in the temporally discretized signal is an FIR filter having an even number of tap coefficients and symmetrical tap coefficients. In the optical disk drive according to claim 1 or 2,
The optical distance from the laser light source to the recording layer of the optical disc is not less than the minimum distance allowed by the high-frequency superposition method at the minimum reproduction speed and not more than the maximum distance allowed by the high-frequency superposition method at the maximum reproduction speed. Features an optical disk drive. A laser light source;
A light source driving unit that drives the laser light source at a frequency based on a channel clock frequency; and
An optical system for irradiating an optical disc with laser light generated from the laser light source;
A photodetector for receiving the laser beam reflected from the optical disc;
Means for converting the output of the photodetector into an electrical pulse regeneration signal;
A carrier clock oscillator,
Means for converting the pulse reproduction signal into a temporally discrete signal at a carrier clock frequency oscillated by the carrier clock oscillator;
Means for suppressing an emission line spectrum derived from pulse-driven emission of the laser light source in the temporal discretization signal;
A channel clock generator for generating a channel clock;
An interpolator that converts the temporally discretized signal in which the emission line spectrum is suppressed into a sampled pulse reproduction signal using the frequency of the channel clock;
An optical disc drive comprising: means for suppressing harmonic components of an output signal from the interpolator.