JP2011166924A - Motor driving device for driving three-phase variable speed motor - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To reduce a step-up ratio of a motor driving device for driving a three-phase variable speed motor having a wide torque range and a wide speed range. <P>SOLUTION: A DC/DC converter 8 supplies a maximum interphase voltage (Vx), having a three-phase full-wave rectified waveform, to an inverter 4. The three-phase inverter 4 drives a variable speed motor 6. An upper-arm switch of one leg of the inverter 4 is turned on. A lower-arm switch of another leg of the inverter 4 is turned on. The remaining one leg of the inverter 4 is PWM-switched. The leg to be PWM-switched is sequentially changed. It is configured to adopt an existing PWM method for switching a plurality of phases instead of one-phase PWM switching, preferably, in a small-torque and low-speed state. <P>COPYRIGHT: (C)2011,JPO&INPIT

Description

  The present invention relates to a motor drive device for driving a three-phase variable speed motor, preferably a permanent magnet motor for driving a vehicle.

  A vehicle drive motor requires a wide rotational speed range of, for example, 0 to 10,000 rpm. The rotational speed of the vehicle drive motor is proportional to the vehicle speed. The stator winding of the motor induces a generated voltage (EMF) proportional to the motor rotation speed by the electromagnetic induction principle. In particular, a permanent magnet type synchronous motor induces a large EMF during high-speed rotation. The motor torque is proportional to the motor current. Therefore, the inverter must apply a high voltage to the vehicle drive motor during high speed rotation in order to maintain the motor current.

  A chopper type step-up DC / DC converter that applies a step-up voltage to an inverter can drive a permanent magnet motor during high-speed rotation. However, the chopper type boost DC / DC converter outputs a pulse-shaped boost voltage. The duty ratio of the boosted voltage decreases when the boosted voltage value increases. That means that the boosted power supplied to the motor decreases during high-speed rotation. In order to increase the boost voltage of the motor drive device, it is desired to reduce the boost ratio of the boost voltage.

  A single-phase PWM switching method (SPSM) of a grid-connected voltage-type three-phase inverter has been proposed in the following non-patent literature.

  This SPSM method controls the DC input power of the converter to be constant. The SPSM method shown in FIG. 1 has a step-up DC / DC converter 80 that applies a periodic change voltage to a three-phase inverter 40 connected to a grid instead of a known constant DC link voltage. The inverter 40 outputs a three-phase voltage to the grid network 200 through the three-phase low-pass filter 100. The maximum interphase voltage of boost converter 80 is applied to inverter 40. The smoothing capacitor 8D connected to the DC link bus line has a small capacity. The DC power source 70 applies a DC voltage to the reactor 90 of the chopper type boost converter 80.

  Only one phase leg of the three-phase inverter is PWM switched for a predetermined period. The upper arm switch of the other phase leg and the lower arm switch of the remaining one phase leg remain on during this period. The period-change voltage applied to the inverter from the step-up DC / DC converter has a three-phase full-wave rectified waveform. Furthermore, the legs that are PWM controlled are changed in order.

  In the above description regarding the SPSM method, the DC input power supplied to the grid network from the DC power source through the step-up DC / DC converter and the grid connection inverter is controlled to be constant. The instantaneous value of the maximum interphase voltage of the grid network is larger than the voltage value of the DC power supply. After all, it is difficult to adopt the SPSM method for the motor drive device for variable speed motor drive. Variable speed motors, especially vehicle motors, have a wide rotational speed range and a wide motor torque range. As a result, the three-phase voltage applied to the variable speed motor must have a wide amplitude range.

  Switching loss reduction of single-phase PWM control method, IEEJ, 4-029, 2009.

An object of the present invention is to reduce the step-up ratio of a motor driving device for driving a three-phase variable speed motor having a wide torque range and a wide speed range. Another object of the present invention is to provide a motor driving device for driving a three-phase variable speed motor having a wide torque range and a wide speed range and excellent efficiency. Another object of the present invention is to provide a motor driving device for driving a three-phase variable speed motor having a wide torque range and a wide speed range, excellent efficiency and low surge power.
In the present invention, the three-phase variable speed motor is operated by a step-up voltage source inverter driven by the SPSM method. In the high-speed rotation region, the motor current decreases due to the increase of the induced motor voltage. Therefore, the boost ratio must be increased during high speed rotation. However, since the output period of the boost converter is reduced, the motor current supplied by the boost DC / DC converter inverter is reduced. As a result, the motor torque proportional to the motor current decreases during high speed rotation.

  It has been found that the step-up ratio of the motor driving device of the present invention can be reduced as compared with known motor driving devices. As a result, more electric power can be supplied to the variable speed motor during high speed rotation. In other words, the motor driving device of the present invention can have a lower step-up ratio than known motor driving devices.

  The boost converter of the present invention applies the maximum interphase voltage (Vx) to the three-phase inverter. The interphase voltage means a voltage between two phase voltages of the three phase voltages. The maximum interphase voltage (Vx) is less than twice the single phase voltage. As a result, the boost ratio of the converter can be reduced by about 15-20%.

  Since the boost ratio decreases, the voltage output period of the boost converter increases. The motor torque of the variable speed motor is proportional to the current supplied to the motor. After all, the variable speed motor of the present invention can generate a strong torque during high speed rotation by adopting the SPSM method.

  Additional explanation of the SPSM method is described below. The maximum interphase voltage (Vx) is changed every 60 degrees. Only one-phase half-bridge of the three-phase inverter that is one leg is PWM-switched. The legs that are PWM switched are changed in turn every 60 degrees of electrical angle. The other two fixed legs, each half bridge, are not PWM switched.

  The three-phase inverter of the present invention outputs a symmetric three-phase voltage, preferably a sinusoidal three-phase voltage, by one-phase half-bridge PWM switching. As a result, the switching loss of the three-phase inverter is reduced. In one PWM carrier cycle, four switching operations are performed instead of the traditional inverter's 12 switching operations. The boost converter outputs a boost voltage having a three-phase full-wave rectified waveform.

  In order to control the PWM leg, the duty ratio increase mode and the duty ratio decrease mode are alternately executed at every electrical angle of 60 degrees. In the duty ratio increase mode, the PWM duty ratio continuously increases from 0% to 100%. In the duty ratio reduction mode, the PWM duty ratio continuously decreases from 100% to 0%. The pair of legs for outputting the maximum interphase voltage (Vx) is changed in turn every 60 degrees of electrical angle.

  According to a preferred embodiment, the three-phase variable speed motor has a permanent magnet fixed to the rotor. In the single leg switching mode, the converter applies a maximum interphase voltage (Vx) larger than the motor induced voltage to the inverter. As a result, the motor drive device can execute the single leg switching mode, that is, the SPSM method even if the motor rotates at high speed.

  According to a preferred embodiment, the motor driving device of the present invention driven by the SPSM method changes the duty ratio of the boost converter according to the received torque command value (Tr) of the motor and the detected rotational speed (ω) of the motor. . As a result, the motor drive device driven by the SPSM method can smoothly generate the required torque even if the rotational speed and the torque command value change over a wide range.

  According to a preferred embodiment, the controller has a table holding a relationship between the maximum interphase voltage (Vx), the rotor rotation angle (θ), the torque command value (Tr), and the rotation speed (ω). The controller determines the maximum interphase voltage (Vx) based on the detected values of the rotor rotation angle (θ), torque command value (Tr), and rotation speed (ω). As a result, the variable speed motor using the SPSM method smoothly generates the required torque at the required rotational speed.

  According to a preferred embodiment, the converter has a reactor and a half bridge. The half bridge has an upper arm switching element and a lower arm switching element connected in series. One end of each of the upper arm switching element and the lower arm switching element is connected to the power supply device through a reactor. The other end of the upper arm switching element is connected to the inverter through a high potential bus line. The other end of the lower arm switching element is connected to the power supply device and the inverter through a low potential bus line. The above-described chopper type boost converter has a simple configuration and can be operated as a bidirectional converter. As a result, the motor can charge the power supply during deceleration.

  According to a preferred embodiment, the controller has a single leg switching mode and a multi-phase leg switching mode for control of the converter and inverter. The multi-phase leg switching mode is essentially the same as the traditional PWM switching operation of a three-phase motor. For example, one of the multi-phase leg switching modes is traditional three-phase PWM switching operation. The three legs of the three-phase inverter are PWM-switched by traditional three-phase switching operation. Another one of the multi-phase leg switching modes is a known two-phase modulation method or a known instantaneous vector method. The two legs of the three-phase inverter are PWM switched by the two-phase modulation method or the instantaneous vector method.

  The single leg switching mode is adopted when the command value of the maximum interphase voltage (Vx) is larger than the voltage of the power supply device. The multi-phase leg switching mode is employed when the command value of the maximum interphase voltage (Vx) is smaller than the voltage of the power supply device. Therefore, the motor driving device operated by the SPSM method can be driven by a known multi-phase leg switching mode even if the motor voltage is smaller than the power supply voltage. As a result, the motor driving device using the SPSM method can apply the three-phase voltage to the variable speed motor even if the three-phase voltage is small.

  According to a preferred embodiment, in single leg switching mode, the upper arm switching element of one switched leg is turned off within a period in which the converter does not output a boosted voltage. According to a preferred embodiment, in single leg switching mode, the upper arm switching element of one switched leg is turned on within a period in which the converter outputs a boosted voltage. As a result, PWM-switched legs are operated with excellent efficiency.

  Further details are described below. One drawback of the SPSM method is the large potential fluctuation of the high potential bus line between the converter and the inverter. The traditional high potential bus line between the converter and the inverter is connected to a large capacity smoothing capacitor. In the present invention, since the maximum interphase voltage (Vx) is rapidly changed, a large-capacity smoothing capacitor cannot be connected to a high potential bus line. As a result, the voltage ripple on the high-potential bus line adversely affects the small-amplitude interphase voltage (Vy) applied from the PWM-switched leg of the three-phase inverter.

  In one example, the PWM-switched leg performs PWM switching to output the small amplitude interphase voltage (Vy) only while the boost converter outputs the maximum interphase voltage (Vx). The turn-on period of the upper arm switching element of the PWM-switched leg is shorter than the output period of the boost converter that outputs the maximum interphase voltage (Vx). Therefore, the leg that is PWM-switched is not output when the high-potential bus line drops. As a result, the voltage ripple of the small amplitude interphase voltage (Vy) is reduced.

  According to a preferred embodiment, the upper arm switching element of one PWM switched leg is turned off in a period that overlaps the turn on of the lower arm switch of the converter. As a result, voltage noise can be reduced. According to a preferred embodiment, the upper arm switching element of one PWM switched leg is turned on in a period that overlaps with the turn-off of the lower arm switch of the converter. As a result, voltage noise can be reduced.

  According to a preferred embodiment, the converter and the inverter are connected through a high potential bus line (100) made of a conductor plate. Each upper arm switching element and each upper arm freewheeling diode of the inverter and converter are fixed to this conductive plate. The upper arm switching element and freewheeling diode of the converter are surrounded by each upper arm switching element and each freewheeling diode of the inverter. As a result, surge noise of the high potential bus line (100) caused by switching of the inverter is reduced. This is because the inductance of the short high potential bus line is reduced.

FIG. 1 is a block circuit diagram of a known grid-connected three-phase inverter operated by the SPSM method. FIG. 2 is a block circuit diagram showing a motor drive device for driving a three-phase variable speed motor. FIG. 3 is a schematic connection diagram showing six switching states of the three-phase inverter. FIG. 4 is a diagram illustrating six gate voltage patterns of the three-phase inverter. FIG. 5 is a timing chart showing one PWM carrier period of the three-phase inverter. FIG. 6 is a timing chart showing the three-phase voltage applied to the motor by the three-phase inverter. FIG. 7 is a timing chart showing the maximum interphase voltage applied to the inverter by the converter. FIG. 8 is a block diagram of a controller that controls the three-phase inverter converter. FIG. 9 is a vector diagram of the maximum interphase voltage and the small amplitude interphase voltage. FIG. 10 is a circuit diagram showing a state where the converter outputs the maximum interphase voltage and the switched phase leg of the three-phase inverter is turned off. FIG. 11 is a circuit diagram showing a state where the converter outputs the maximum interphase voltage and the switched phase leg of the three-phase inverter is turned on. FIG. 12 is a circuit diagram showing a state in which the converter does not output the maximum interphase voltage and the switched phase leg of the three-phase inverter is turned off. FIG. 13 is a timing chart of the converter and the legs to be switched. FIG. 14 is a timing chart of the converter and the legs to be switched. FIG. 15 is another timing chart of the converter and the legs to be switched. FIG. 16 is a timing chart showing an error tracking PWM method as an example of the PWM method. FIG. 17 is a circuit diagram for implementing the error tracking PWM method. FIG. 18 is a flowchart showing the operation control of the variable speed motor. FIG. 19 is a timing chart showing various waveform patterns from the converter. FIG. 20 is a plan view showing the arrangement of each switch and each freewheeling diode element in the motor drive device. FIG. 21 is a schematic cross-sectional view of the motor drive device shown in FIG. 22 is a cross-sectional view of the motor drive device shown in FIG.

  The SPSM drive motor drive apparatus of this embodiment will be described with reference to the drawings. FIG. 1 is a circuit diagram of an SPSM drive type motor drive device. This motor drive device has a three-phase inverter 4, a smoothing capacitor 5, a chopper type step-up DC / DC converter 8, a controller 9, and a three-phase smoothing capacitor circuit 10. The DC voltage of the battery 7 is applied to the converter 8. The inverter 4 outputs a three-phase voltage to a three-phase synchronous motor 6 for driving the vehicle. The motor 6 has a U-phase winding 6U, a V-phase winding 6V, and a W-phase winding 6W that are star-connected. Each capacitor of the three-phase smoothing capacitor circuit 10 is individually connected to each terminal of the three-phase winding of the motor 6.

The three-phase inverter 4 has a U-phase leg 1, a V-phase leg 2, and a W-phase leg 3. Each leg is constituted by a half bridge. The U-phase leg 1 has an upper arm switch 11 and a lower arm switch 12 connected in series. The V-phase leg 2 has an upper arm switch 21 and a lower arm switch 22 connected in series. The W-phase leg 3 has an upper arm switch 31 and a lower arm switch 32 connected in series. Each switch comprises a transistor and a freewheeling diode connected in parallel with each other. The three-phase inverter 4 drives a three-phase AC motor 6.

  The step-up DC / DC converter 8 changes the battery voltage Vb to a DC link voltage Vx that varies periodically. The converter 8 outputs the DC link voltage Vx to the three-phase inverter 4 through the high potential bus line 100 and the low potential bus line 101. The converter 8 periodically changes the amplitude of the DC link voltage Vx applied to the inverter 4.

  The SPSM operation of the inverter 4 will be described with reference to FIG. FIG. 3 has stages A-F. The electrical angle of 360 degrees is divided into six stages A-F. Stages A-F each having an electrical angle of 60 degrees are selected in order according to the detected rotor angle. In each of the stages A-F, the four switches of the two half bridges maintain a certain state. In each of the stages A-F, only two switches of one half bridge are PWM switched to form a sinusoidal small amplitude interphase voltage (Vy). Hereinafter, a half bridge that is PWM-switched is referred to as a PWM switch leg. The other two half bridges are called fixed legs.

  V-phase leg 2 is a PWM switch leg in stages A and D. W-phase leg 3 is a PWM switch leg in stages B and E. U-phase leg 1 is a PWM switch leg in stages C and F. The upper arm switch and lower arm switch of the PWM switch leg are PWM-switched.

  In stage A, the upper arm switch 11 of the U-phase leg 1 and the lower arm switch 32 of the W-phase leg 3 are turned on. As a result, the maximum interphase voltage (Vx) is applied to the U-phase winding 6U and the W-phase winding 6W. In stage A, the U-phase leg 1 and the W-phase leg 3 are fixed legs.

  Stages A-F are determined according to the rotor angle detected by the rotor angle sensor. Instead of detecting the rotor angle, the stage A-F can be determined by the induced phase voltages of the phase windings 6U, 6V and 6W. If the induced U-phase voltage is maximum, the stage is A or D. If the induced V-phase voltage is maximum, the stage is B or E. If the induced W-phase voltage is maximum, the stage is C or F.

  In FIG. 3, the U-phase leg 1 outputs a U-phase voltage Vu. V-phase leg 2 outputs V-phase voltage Vv. The W-phase leg 3 outputs a W-phase voltage Vw. A voltage between two phase voltages selected from among the three phase voltages Vu, Vv, and Vw is called an interphase voltage. The interphase voltage having the maximum amplitude is called the maximum interphase voltage (Vx).

  FIG. 4 shows the states of the switches 11-12, 21-22 and 31-32 of the three-phase inverter 4 in the stage A-F. The gate voltage UU is applied to the switch 11. The gate voltage UL is applied to the switch 12. The gate voltage VU is applied to the switch 21. The gate voltage VL is applied to the switch 22. The gate voltage WU is applied to the switch 31. The gate voltage WL is applied to the switch 32. Each switch of the inverter 4 is PWM-switched for a period of 60 degrees. In the next 120 degree period, each switch is turned off and cooled. Therefore, the temperature rise of each switch is suppressed.

  FIG. 5 shows waveforms of gate voltages applied to the six switches of the inverter 4 in one PWM carrier period TP of the stage A. In the PWM carrier period TP, the switches 11 and 32 are turned on and the switches 12 and 31 are turned off. The V-phase leg switches 21 and 22 are PWM-switched.

  One of the two switches of the PWM switch leg has a duty ratio that continuously varies from 0% to 100% in a 60 degree period. The other one of the two switches of the PWM switch leg has a duty ratio that continuously varies from 100% to 0% over a 60 degree period.

  The SPSM operation of the converter 8 will be described with reference to FIGS. FIG. 6 shows a three-phase sine wave voltage applied to the motor 6. The converter 8 outputs a DC link voltage Vx which is the maximum interphase voltage (Vx) to the inverter 4.

  The boosting operation itself of the chopper type boosting converter 8 is well known. When the lower arm switch 8F is turned on, the reactor 8C accumulates magnetic energy. When the lower arm switch 8F is turned off, the boosted voltage is applied to the high potential bus line 100 through the switch 8E.

  The smoothing capacitor 5 connects the high potential bus line 100 and the positive terminal of the battery 7. When the upper arm switches 11, 21, and 31 are turned off, the smoothing capacitor 5 absorbs surge energy. Further, the smoothing capacitor 5 absorbs the voltage ripple of the high potential bus line 100. However, the large-capacity smoothing capacitor 5 regulates the change in the maximum interphase voltage (Vx).

  The controller 9 calculates the duty ratio Dx of the converter 8 in order to output the sine wave maximum interphase voltage (Vx) based on the motor torque command value and the detected rotor angle value. The controller 9 can control the converter 8 by the PWM feedback control method. Further, the controller 9 calculates the duty ratio Dy of the inverter 4 in order to output a sine wave small amplitude interphase voltage (Vy) based on the motor torque command value and the detected rotor angle value. The maximum interphase voltage (Vx) and the small amplitude interphase voltage (Vy) are shown in FIG.

  As shown in FIG. 6, the maximum interphase voltage (Vx) is changed in turn every 60 degrees of electrical angle. In stage A from 30 degrees to 90 degrees, the maximum interphase voltage (Vx) is the interphase voltage Vu-Vw. In stage B from 90 degrees to 150 degrees, the maximum interphase voltage (Vx) is the interphase voltage Vu-Vv. In stage C from 150 degrees to 210 degrees, the maximum interphase voltage (Vx) is the interphase voltage Vw-Vv.

  In stage D from 210 degrees to 270 degrees, the maximum interphase voltage (Vx) is the interphase voltage Vw-Vu. In stage E from 270 degrees to 330 degrees, the maximum interphase voltage (Vx) is the interphase voltage Vv-Vu. In the stage F from 330 degrees to 30 degrees, the maximum interphase voltage (Vx) is the interphase voltage Vv-Vw.

  The maximum interphase voltage (Vx) has the waveform shown in FIG. The waveform of the maximum interphase voltage (Vx) is equal to the full-wave rectified waveform of the three-phase voltage. If the maximum value of the one-phase voltage is 1, the value of the maximum interphase voltage (Vx) is 1.5-1.73.

  Therefore, the step-up ratio of the converter 8 is 75-86.5% of the step-up ratio of the existing motor driving device having the converter and the inverter. As a result, the upper arm switch 8E of the converter 8 can have a higher duty ratio than the existing motor drive device. For example, when the battery voltage Vb is 250V, the converter of the existing motor driving device outputs 700V. The step-up ratio is 2.8. On the other hand, the converter of the motor drive device of this embodiment may output 525-605V. The maximum interphase voltage (Vx) of both motor drives is equal. Thus, both motor drives have the same maximum interphase voltage (Vx). Therefore, the converters of both motor drives have equal output current values.

  In each stage A-F, the maximum interphase voltage (Vx) has a partial sinusoidal waveform. As a result, as shown in FIG. 6, only one phase is PWM switched to output a sinusoidal waveform. The value of the small amplitude interphase voltage (Vy) is alternately changed from 0% to 100% and from 100% to 0% of the maximum interphase voltage (Vx).

  The memory of the controller 9 has a map that holds the relationship between the relative duty ratio Dz and the rotor rotation angle in order to determine the small amplitude interphase voltage (Vy). The relative duty ratio Dz is equal to Dy / Dx. The relative duty ratio Dz indicates a relative amplitude ratio between the small amplitude interphase voltage (Vy) and the maximum interphase voltage (Vx). The half bridge constituting the PWM leg is PWM-switched with this PWM duty ratio Dz.

  The controller 9 reads the relative duty ratio Dz from the map based on the detected rotor angle θ. The map stores the relative duty ratio Dz for each rotor angle. The upper arm switches 11, 21 and 31 of the PWM leg are PWM-switched with a relative duty ratio Dz. The lower arm switches 12, 22 and 32 of the PWM leg are PWM-switched with a relative duty ratio 1-Dz. Thus, the three-phase voltages Vu, Vv, and Vw are determined only by the PWM switching of the one-phase leg of the three-phase inverter 4.

  FIG. 8 shows a part of a block diagram of the controller 9. The controller 9 includes a stage determination circuit 10A, waveform generation circuits 10B and 10D, and PWM signal generation circuits 10C and 10E. The stage determination circuit 10A determines the current stage based on the rotor angle θ. The map shown in FIG. 4 in memory is used for this determination. The waveform generation circuit 10B generates a PWM signal having a relative duty ratio Dz for the PWM phase leg based on the rotor angle θ. The PWM signal generation circuit 10C generates the PWM gate voltages UU, UL, VU, VL, WU and WL every 60 degree period based on the determined stage and the determined PWM signal.

  The waveform generation circuit 10D generates a duty ratio Dx of the DCDC converter 8B. The waveform of the maximum interphase voltage (Vx) changes as shown in FIG. The maximum interphase voltage (Vx) is determined based on the detected rotor angle θ and torque command value Ti. The PWM signal generation circuit 10E generates a PWM gate voltage for the DCDC converter 8B.

The voltages Vx and Vy are displayed as vectors in FIG.
Vu = Vm sin ωt
Vv = Vm (sin ωt-2π / 3)
Vw = Vm (sin ωt + 2π / 3)
Vx = Vu-Vw = 1.73 * Vm * sin (ωt-2π / 3) = 1.73 * Vm * Dx
Vy = Vv-Vw = 1.73 * Vm * sin (ωt-π / 2) = 1.73 * Vm * Dy
The PWM duty ratio Dx of the maximum interphase voltage (Vx) is represented by a sine wave function sin (ωt-2π / 3). The PWM duty ratio Dy of the small-amplitude interphase voltage (Vy) is represented by a sine wave function sin (ωt−π / 2). Therefore, the relative duty ratio Dz = Dy / Dx can be obtained by calculating the following equation.
Dz = sin (ωt-π / 2) / sin (ωt-2π / 3)

  The previously calculated duty ratios Dx and Dz are described in the memory map. Accordingly, the duty ratios Dx and Dy are extracted from the map by using the rotor angle θ = ωt. The command value for the maximum interphase voltage (Vx) is 1.73 * Vm. This command value is calculated based on the motor torque command value. The command value of the calculated maximum interphase voltage (Vx) is compared with the detected DC link voltage Vx.

  The duty ratio of the converter 8 can be feedback controlled by the comparison result. Furthermore, the upper arm switches 11, 21, and 31 of the PWM leg are PWM-switched with a relative duty ratio Dz. The lower arm switches 12, 22, and 32 of the PWM leg are PWM-switched with a relative duty ratio 1-Dz.

  An example of this SPSM motor driving apparatus will be described with reference to FIGS. FIG. 10 to FIG. 12 show circuit diagrams of motor driving devices for driving a three-phase motor, respectively. The device has a three-phase inverter 4 and a converter 8. The three-phase inverter 4 and the boost converter 8 are the same as the inverter 4 and the converter 8 shown in FIG.

  The converter 8 has an upper arm switch 8E and a lower arm switch 8F connected in series. The connection point of the switches 8E and 8F is connected to the positive terminal of the battery 8A through the reactor 8C. The smoothing capacitor 8D connects between the two DC link lines 100 and 101. The well-known chopper type converter 8 is a bidirectional step-up / step-down converter that outputs the step-up voltage Vx to the three-phase inverter 4 and the step-down voltage Vb to the battery 8A.

  The operation of the motor drive device will be described below. 10 to 12 show the operation in stage A. FIG. The U-phase leg 1 and the W-phase leg 3 are fixed legs. V-phase leg 2 is a PWM leg. In FIG. 10, switches 8E, 11, 22, and 32 are turned on. The boosted voltage Vx supplies a current I to the switch 11. Current I corresponds to U-phase current Iu. The switch 22 passes a V-phase current that is a freewheeling current. In FIG. 11, switches 8E, 11, 21 and 32 are turned on. The boosted voltage Vx supplies the current I to the switches 11 and 21. Therefore, current I corresponds to the sum of U-phase current Iu and V-phase current Iv.

  In FIG. 12, switches 8F, 11, 22 and 32 are turned on. Reactor 8C stores magnetic energy. Smoothing capacitor 8D passes U-phase current Iu. However, the V-phase upper arm switch 21 is turned off when the switch 8F is turned on. The smoothing capacitor 8D does not need to pass the V-phase current Iv. Therefore, the voltage drop of the smoothing capacitor 8D is reduced.

  FIG. 13 is a timing chart showing the operation of the PWM switch leg of the inverter 4 and the converter 8. The upper arm switch 21 of the PWM switch leg is turned on during the period from the time point t1 to t2 in the output period of the converter 8 between the time points t3 and t2. Therefore, the smoothing capacitor 8D becomes small. Further, the upper arm switch 21 of the PWM switch leg is turned off at the same time as the upper arm switch 8E of the converter 8 is turned off. Thereby, voltage ripple can be reduced.

  A modification of the SPSM motor driving apparatus will be described with reference to FIG. In this embodiment, the upper arm switch 21 of the PWM switch leg is turned on when the upper arm switch 8E is turned on. Therefore, the turn-on period of the upper arm switch 21 of the PWM switch leg partially overlaps the turn-on period of the upper arm switch 8E. In FIG. 14, the turn-on period of the upper arm switch 21 of the PWM switch leg overlaps with the odd-numbered turn-on period of the upper arm switch 8E. If the duty ratio of the upper arm switch 21 of the PWM switch leg is small, the gate pulse P3 'is further canceled. If the upper arm switch 21 of the PWM switch leg has a larger duty ratio, a gate pulse voltage VU that overlaps with the even-numbered gate pulse voltage CU is further added.

  A modification of the SPSM motor driving apparatus will be described with reference to FIG. FIG. 15 is a timing chart showing a relative time relationship between the gate voltages of the inverter and the converter. The gate voltage CU is applied to the upper arm switch 8E. The gate voltage CL is applied to the lower arm switch 8F. The maximum interphase voltage (Vx) is changed by the gate voltage CL. The upper arm switch 21 selects one of the two gate voltages VU1 and VU2.

The gate voltage VU1 rises at substantially the same timing as the fall timing of the gate voltage CL. As a result, the voltage ripple on line 100 is reduced. This is because the increased voltage Vx of the line 100 due to the turn-off of the lower arm switch 8F is reduced by the turn-on of the upper arm switch 21. The gate voltage VU2 drops at substantially the same timing as the rising timing of the gate voltage CL. As a result, the voltage ripple on line 100 is reduced. This is because the decrease voltage Vx of the line 100 due to the turn-on of the lower arm switch 8F is reduced by the turn-off of the upper arm switch 21.

  After all, in this modification, the upper arm switches 11, 21, and 31 of the inverter 4 are turned on simultaneously with the turn-off timing of the lower arm switch 8F. In another modification, the upper arm switches 11, 21, and 31 of the inverter 4 are turned off simultaneously with the turn-on timing of the lower arm switch 8F. As a result, the noise on the line 100 and the switching loss of the upper arm switch of the inverter 4 are reduced.

  Another variant of the SPSM motor drive is described with reference to FIGS. FIG. 16 shows the principle of the error tracking PWM method. It is also called the hysteresis band type PWM method. The error tracking PWM method can be adopted instead of the traditional PWM method of constant frequency PWM carrier signal to form the maximum interphase voltage (Vx) and the small amplitude interphase voltage (Vy).

  In FIG. 16, the broken line indicates the command value of the maximum interphase voltage (Vx). Two solid lines Vx + ΔV and Vx−ΔV are formed on both sides of the broken line Vx. The DCDC converter 8 outputs the maximum interphase voltage (Vx) within the range of two voltages Vx + ΔV and Vx−ΔV.

  FIG. 17 shows a comparator circuit of the error tracking type PWM method. The detected value of the maximum interphase voltage (Vx) is compared with Vx + ΔV and Vx−ΔV by the comparators 91 and 92. The AND gate 93 controls the upper arm switch 8E shown in FIG. Similarly, the detected value Vyd of the small amplitude interphase voltage (Vy) is compared with Vy + ΔV and Vy−ΔV by the comparators 94 and 95. The AND gate 96 controls the upper arm switch 21 of the PWM phase leg 2 shown in FIG.

  Another variant of the SPSM motor drive is described with reference to FIGS. Boost converter 8 outputs a maximum interphase voltage (Vx) greater than battery voltage Vb. However, boost converter 8 cannot output the maximum interphase voltage (Vx) smaller than battery voltage Vb. This means that the SPSM method cannot be applied to the motor drive device when the motor torque or the rotational speed is lower than a predetermined value.

  In other words, the maximum interphase voltage (Vx) is a function value of the motor torque command value Tr and the rotational speed ω. The maximum interphase voltage (Vx) is proportional to the motor torque command value Tr and the rotational speed ω. After all, the SPSM method cannot be operated when the calculated maximum interphase voltage (Vx) is smaller than a predetermined value. For this reason, a motor drive device such as an EV motor that often outputs a small torque at a low speed cannot be operated by the SPSM method. This solution is described below.

  FIG. 18 is a flowchart showing the control operation of the SPSM motor driving device. In FIG. 18, the torque command value Tr, the rotor angle θ, and the rotational speed ω are detected in step S100. Next, the maximum interphase voltage (Vx) is calculated based on step S102 based on the torque command value Tr, the rotor angle θ, and the rotational speed ω.

  The controller 9 has a table showing the relationship among the maximum interphase voltage (Vx), the torque command value Tr, the rotor angle θ, and the rotational speed ω. Further, in step S102, it is determined whether or not the maximum interphase voltage (Vx) is larger than the battery voltage Vb. When the maximum interphase voltage (Vx) is not greater than the battery voltage Vb, the multi-phase leg switching mode is selected. In the multi-phase leg switching mode, a traditional PWM switching method in which two or three legs are PWM switched is executed in S104.

  In step S102, when the maximum interphase voltage (Vx) is larger than the battery voltage Vb, the single leg switching mode is selected. Single leg switching mode control is executed in steps S106 and S108. In step S106, one of the stages A to F shown in FIG. 6 is selected based on the detected rotor angle θ. The controller 9 has a table representing the relationship between the stage A-F and the rotor angle θ.

  Next, in step S108, the gate signals S1 and S2 are calculated based on the torque command value Tr, the rotor angle θ, and the rotational speed ω. The gate signal S1 indicates the duty ratio of the PWM leg of the inverter 4. The gate signal S2 indicates the duty ratio of the converter 8. The controller has a table showing the relationship between the signals S1 and S2, the torque command value Tr, the rotor angle θ, and the rotational speed ω.

  The duty ratio of the PWM leg of the inverter 4 is further explained. The PWM leg outputs a small amplitude interphase voltage (Vy). In order to output the small amplitude phase voltage (Vy), the relative duty ratio Dz of the PWM leg is calculated based on the detected rotor angle θ and the relationship between the rotor angle θ and the relative duty ratio Dz. The relative duty ratio Dz is proportional to the ratio between the value Vy and the value Vx. The determined relative duty ratio Dz of the PWM phase is given to the upper arm switch of the PWM phase. The determined relative duty ratio 1-Dz of the PWM phase is given to the lower arm switch of the PWM phase. The lower arm switch of the PWM phase performs the opposite operation of the upper arm switch of the PWM phase.

  FIG. 19 shows several waveforms of the maximum interphase voltage (Vx). In FIG. 19, the maximum interphase voltages Vx1, Vx2, Vx3, Vx4, Vx5 and Vx6 have different amplitudes. These waveforms of maximum interphase voltage are essentially equal to the maximum interphase voltage waveform shown in FIG. However, the maximum interphase voltages Vx1, Vx2, Vx3, Vx4, Vx5 and Vx6 are different from each other because the motor torque command values are different from each other. A period Ty indicates a period in which the voltage Vx1 is higher than the battery voltage V1. A period Tx indicates a period in which the voltage Vx1 is lower than the battery voltage V1. The SPSM method is executed in the period Ty. The traditional PWM method is executed in the period Tx.

  For example, the SPSM method is performed when the maximum interphase voltage (Vx) is higher than the battery voltage (230V) (231V-700V). As a result, the motor driving device of this aspect can drive the motor even if the torque and the rotational speed are small.

  One problem with the SPSM method is the voltage ripple on the high potential bus line 100 connecting the converter 8 and the inverter 4. The SPSM method has a large voltage ripple on the high potential bus line 100 because the smoothing capacitor 5 is smaller than the traditional smoothing capacitor. 20-22 show the structure of a motor drive with small electromagnetic radiation. FIG. 20 is a plan view of the motor drive device. FIG. 21 is a schematic cross-sectional view of the apparatus shown in FIG. FIG. 22 is a vertical sectional view of the apparatus shown in FIG.

  In FIG. 21, eight switches 11, 21, 31, 12, 22, 32, 8E and 8F of the inverter 4 and converter 8 and eight freewheeling diodes FWD are formed of copper plates 100, 101, 100U, 100V, 100W and 100I. It is connected to the. Copper plates 100U, 100V, and 100W, which are AC lines, connect the inverter 4 and the motor 6. A copper plate 100, which is a high potential bus line, connects the inverter 4 and the converter 8. A copper plate 101, which is a low potential bus line, connects the inverter 4 and the converter 8 to the battery 7.

  Upper arm switches 11, 21, 31, and 8 E and four freewheeling diodes FWD are arranged on copper plate 100. It is important that the upper arm switch and freewheeling diode of the inverter 4 are fixed around the upper arm switch 8E and freewheeling diode FWD of the converter 8. As a result, the inductance of the line 100 between the converter 8 and the inverter 4 is reduced.

  20 and 22 show a water cooling device. The copper plate 100U, which is an AC line, is composed of bus plates 100UA and 100UB. The copper plate 100V, which is an AC line, is composed of bus plates 100VA and 100VB. The copper plate 100W, which is an AC line, is composed of bus plates 100WA and 100WB.

  In FIG. 22, the upper arm switch assembly 600 and the lower arm switch assembly 700 are sandwiched by cooling pipes 300, 400 and 500. The coolant C.L. flows through the cooling pipes 300, 400 and 500. The insulating sheet 200 is disposed between the cooling pipes 300, 400 and 500 and the silicon chip. The silicon chip is composed of a switch chip and a freewheeling diode chip. As a result, since the copper plate 100 is sandwiched between the other copper plate and the cooling pipe, the radiation noise of the copper plate 100 is reduced.

Claims (10)

A voltage-type three-phase inverter that applies a three-phase voltage to a three-phase motor; and
A step-up DC / DC converter that receives a DC voltage from the power supply device and applies the step-up voltage to the three-phase inverter;
In a motor driving device for driving a three-phase variable speed motor comprising the converter and a controller for controlling the inverter,
The motor has a three-phase winding with star connection,
The controller has a single leg switching mode to control the converter and inverter,
The inverter has one switch leg that is switched according to the rotation angle and two fixed legs in the single leg switching mode,
In the single leg switching mode, the converter outputs a boost voltage having a magnitude equal to the maximum interphase voltage (Vx) to the switch leg,
The maximum interphase voltage (Vx) applied to the motor has a maximum voltage value compared to the voltage between the other two phases in the single leg switching mode,
One upper arm switching element and one lower arm switching element of two fixed legs output the maximum interphase voltage (Vx) in the single leg switching mode,
One switch leg is PWM controlled in single leg switching mode and outputs a small amplitude phase voltage (Vy) smaller than the maximum phase voltage (Vx). apparatus. The three-phase variable speed motor has a permanent magnet fixed to the rotor,
The motor drive device for driving a three-phase variable speed motor according to claim 1, wherein the converter applies a maximum interphase voltage (Vx) having a value larger than an induced voltage of the motor in the single leg switching mode.  2. The three-phase variable speed motor drive according to claim 1, wherein the controller changes the PWM duty ratio of the converter based on the received torque command value (Tr) and the detected motor rotation angle (ω) in the single leg switching mode. Motor drive device. The controller has a relationship table between maximum interphase voltage (Vx), rotation angle (θ), torque command value (Tr) and rotation speed (ω),
4. The three-phase variable speed motor drive according to claim 3, wherein in the single leg switching mode, the maximum interphase voltage (Vx) is determined based on the rotation angle (θ), the torque command value (Tr), and the motor rotation angle (ω). Motor drive device. The converter has a half bridge and a reactor having an upper arm switching element and a lower arm switching element connected in series with each other,
One end of each of the upper arm switching element and the lower arm switching element of the converter is connected to a power supply device through a reactor,
The other end of the converter's upper arm switching element is connected to the inverter through a high-potential bus line.
The motor driving device for driving a three-phase variable speed motor according to claim 1, wherein the other end of the lower arm switching element of the converter is connected to an inverter and a power supply device through a low potential bus line. Furthermore, the controller has a multi-phase leg switching mode for controlling the converter and the inverter,
Multiple inverter legs are PWM switched in multiple phase leg switching mode,
The controller selects the single-leg switching mode when the maximum interphase voltage (Vx) is greater than the voltage of the power supply, and the multi-phase leg switching mode when the maximum interphase voltage (Vx) is less than the voltage of the power supply. 6. The motor driving device for driving a three-phase variable speed motor according to claim 5.   6. The upper arm switching element of one switch leg is turned off in a period in which the converter does not output a boosted voltage and is turned on in a period in which the converter outputs a boosted voltage in the single leg switching mode. Motor drive device for three-phase variable speed motor drive.   6. The motor driving device for driving a three-phase variable speed motor according to claim 5, wherein in the single leg switching mode, the turn-off operation of the upper arm switching element of one switch leg overlaps with the turn-on operation of the lower arm switching element of the converter.   6. The motor driving device for driving a three-phase variable speed motor according to claim 5, wherein in the single leg switching mode, the turn-on operation of the upper arm switching element of one switch leg overlaps with the turn-off operation of the lower arm switching element of the converter. The converter and the inverter are connected through a high potential bus line 100 made of a conductor plate,
The upper arm switching element and the upper arm free-wheeling diode of the inverter and converter are fixed to the conductor plate,
The motor driving device for driving a three-phase variable speed motor according to claim 1, wherein the upper arm switching element and the upper arm freewheeling diode of the converter are surrounded by the upper arm switching element and the upper arm freewheeling diode of the inverter.

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