JP2010038951A - Optical modulator - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide a compact optical modulator greatly improved in characteristic impedance, electric crosstalk and an optical modulation band further. <P>SOLUTION: The optical modulator includes a substrate 1, an optical waveguide 3 formed on the substrate and a traveling wave electrode comprising center electrodes 4a'-4d' and grounding electrodes 5a'-5e' for applying a voltage, and the optical waveguide has a plurality of interacting optical waveguides 3a-3d in an interaction part which is an area where a refractive index is changed by applying the voltage to the traveling wave electrode, the plurality of center electrodes and grounding electrodes are provided respectively, and the electrode of an input side feed part connected to the traveling wave electrode of the interaction part comprises the plurality of center electrodes and grounding electrodes respectively. The optical modulator is structured such that at least two adjacent ones of the equivalent refractive indexes of a plurality of high frequency electric signals propagated through the plurality of center electrodes and grounding electrodes of the input side feed part are different from each other. <P>COPYRIGHT: (C)2010,JPO&INPIT

Description

  The present invention relates to an optical modulator that modulates light incident on an optical waveguide and emits it as an optical signal using an electro-optic effect or a thermo-optic effect.

  There is an optical modulator using a dielectric material as a typical optical modulator. In recent years, high-speed and large-capacity optical communication systems have been put into practical use, and development of high-performance optical modulators to be incorporated into such high-speed and large-capacity optical communication systems is required.

As an optical modulator that meets such demands, a light modulator such as lithium niobate (LiNbO ₃ ) is used for a substrate having a so-called electro-optical effect (hereinafter abbreviated as an LN substrate) whose refractive index changes by applying an electric field. There is a traveling wave electrode type lithium niobate optical modulator (hereinafter abbreviated as an LN optical modulator) in which a waveguide and a traveling wave electrode are formed. This LN optical modulator is applied to a large capacity optical communication system of 2.5 Gbit / s and 10 Gbit / s because of its excellent chirping characteristics. Recently, application to a 40 Gbit / s ultra-high capacity optical communication system is also being studied.

  Hereinafter, characteristics and problems of a conventional LN optical modulator using the electro-optic effect of lithium niobate will be considered.

(First prior art)
In recent years, a 40 Gbit / s ultra-high capacity optical communication system that has been developed uses a phase modulator type LN optical modulator such as DPSK or DQPSK whose principle is disclosed in Patent Document 1, for example. Yes.

  For example, FIGS. 9 and 10 show schematic structural diagrams of an LN optical modulator that embodies a method for generating a DQPSK signal disclosed in Patent Document 1. FIG. Here, only the optical waveguide is shown in FIG. 9 for easy understanding. As described above, the optical waveguide of the DQPSK LN optical modulator is composed of two small Mach-Zehnder interference systems (or Mach-Zehnder optical waveguides) and one large Mach-Zehnder interference system that synthesizes the two small Mach-Zehnder interference systems.

In the figure, 1 is a z-cut LN substrate, 2 is a SiO ₂ buffer layer, and 3 is an optical waveguide. Reference numerals 3a, 3b, 3c, and 3d denote optical waveguides in which light interacts with a high-frequency electrical signal propagating through a traveling-wave electrode described in FIG. 10, and this interacting region I is defined as an interaction region (or interaction unit). ), The optical waveguides 3a, 3b, 3c, and 3d in this region are called interaction optical waveguides.

10 includes, in addition to the optical waveguide 3 shown in FIG. 9, light propagating through the interaction optical waveguides 3a, 3b, 3c, and 3d, and high-frequency electrical signals Sa, Sb, Sc, interacting in the interaction region I. A traveling wave electrode for propagating Sd is also shown. For simplification of explanation, the Si conductive layer for suppressing temperature drift is omitted. The thickness of the SiO ₂ buffer layer 2 is about 1 μm.

  As a traveling wave electrode applied to an LN optical modulator, a CPW (coplanar line or coplanar-waveguide) structure disclosed in Patent Document 2 is generally widely used. The interaction part of this CPW type traveling wave electrode requires a ground electrode in addition to the center electrode (or the center electrode), but in order to make the explanation easy to understand, in FIG. 10, four center electrodes 4a and 4b are used. Only 4c and 4d are shown, and the ground electrode is omitted (the ground electrode is shown in detail in the next FIG. 11). Reference numeral 7 denotes a center electrode of the π / 2 shift electrode, and similarly the ground electrode is omitted. The thickness of the four center electrodes 4a, 4b, 4c and 4d is about 5 to 50 μm.

  When the high-frequency electrical signals Sa, Sb, Sc, Sd are input to the LN optical modulator, these high-frequency electrical signals reach the interaction region I via the input-side feed section shown as II in FIG. Interacts with light.

  Now, taking the conventional DQPSK type LN optical modulator shown in FIG. 10 as an example, the problems of the conventional LN optical modulator having a plurality of center electrodes will be considered. The length of the LN optical modulator chip is as long as about 50 mm, but in order to obtain many chips from one wafer, its width (perpendicular to the length direction of the interaction optical waveguides 3a, 3b, 3c, and 3d). And the length in the direction parallel to the surface of the z-cut LN substrate 1) is as short as 2 mm to 5 mm. For this reason, the width of this region that can be assigned to the input-side feed unit II in the direction parallel to the surface of the z-cut LN substrate 1 is also relatively narrow.

  11 shows a detailed cross-sectional structure of the input-side feed section II in FIG. 10 (perpendicular to the length direction of the interaction optical waveguides 3a, 3b, 3c, and 3d and parallel to the surface of the z-cut LN substrate 1). Direction sectional view).

  In FIG. 11, the ground electrode omitted in FIG. 10 is also shown. Here, the width of the center electrodes 4a, 4b, 4c, 4d is S, the gap between the center electrodes 4a, 4b, 4c, 4d and the ground electrodes 5a, 5b, 5c, 5d, 5e is W, the ground electrode 5b, Let G be the width of 5c and 5d. Hereinafter, FIG. 11 will be considered in detail.

  11 can be said to be a planar (or planar) CPW structure in which the center electrodes 4a, 4b, 4c, and 4d are opposed to each other via the ground electrodes 5b, 5c, and 5d. In this planar CPW electrode structure, the ground electrodes 5a and 5b with respect to the center electrode 4a, the ground electrodes 5b and 5c with respect to the center electrode 4b, the ground electrodes 5c and 5d with respect to the center electrode 4c, and the center electrode 4d with respect to the center electrode 4d. The ground electrodes 5d and 5e are in pairs. In the prior art, the gap between each center electrode and the ground electrode is also equal to W.

As a result, high-frequency electrical signals Sa, Sb, Sc, each of the equivalent refractive index _n a of _Sd, n _b, n c, between the _{n d} consists relationship _{n a = n b = n c} = n d Yes. Thus, when the high-frequency electrical signals propagating through the traveling wave electrodes have the same equivalent refractive index, the high-frequency electrical signals are likely to be coupled to each other, and the high-frequency electrical signals are likely to propagate and electrical crosstalk is likely to occur. The equivalent refractive index of the high-frequency electrical signal may be expressed as the equivalent refractive index of the traveling wave electrode.

  The crosstalk between the center electrodes 4a, 4b, 4c, and 4d (or between the high-frequency electrical signals Sa, Sb, Sc, and Sd) is suppressed by making the width G of the ground electrode wider than the gap W. it can. That is, the ratio between the width G and the gap W of the ground electrodes 5b, 5c, and 5d greatly affects the electrical crosstalk between high-frequency electrical signals. In general, it is desirable that the width G of the ground electrode be 4 to 5 times the gap W in order to sufficiently reduce the electrical crosstalk. Hereinafter, the problems of the prior art will be considered in order.

In FIG. 12, the horizontal axis represents the gap W between the center electrodes 4a, 4b, 4c and 4d and the ground electrodes 5a, 5b, 5c, 5d and 5e, and the left vertical axis represents the characteristic impedance of the CPW traveling wave electrode. Z represents the width G of the ground electrodes 5b, 5c, and 5d on the right vertical axis. As can be seen, the characteristic impedance Z of the traveling wave electrode gap W ₁ in the ground electrode 5b becomes 50Ω matching the characteristic impedance of the external circuit (not shown), 5c, the width G of 5d becomes narrow. In addition, when the width G of the ground electrodes 5b, 5c, and 5d is increased, the gap W between the center electrodes 4a, 4b, 4c, and 4d and the ground electrodes 5a, 5b, 5c, 5d, and 5e becomes narrow (for example, at that time) When W is W ₂ , W ₂ <W ₁ ), and the characteristic impedance Z of the traveling wave electrode is significantly lower than 50Ω.

In FIG. 13, the horizontal axis indicates the gap W between the center electrodes 4a, 4b, 4c, and 4d and the ground electrodes 5a, 5b, 5c, 5d, and 5e, and the vertical axis indicates the high-frequency electrical signals Sa, Sb, and Sc from the traveling wave electrode. , Sd represents an electric power reflection coefficient (or electric power reflectivity) S ₁₁ that returns to an external electric circuit (not shown). As described in FIG. 12, since the characteristic impedance Z of the traveling wave electrode in the case of the gap W ₁ is 50Ω, the electric power reflection coefficient S ₁₁ becomes extremely small. On the other hand, the center electrode 4a, 4b, 4c, 4d and the ground electrode 5a, 5b, 5c, 5d, the gap between 5e is narrow and _{W 2,} the characteristic impedance Z of the traveling-wave electrode is much lower than 50 [Omega (e.g., 30 [Omega Therefore, as shown in FIG. 13, the electric power reflection coefficient S ₁₁ becomes large at about −10 dB or more, which is not preferable as a traveling wave electrode for an LN optical modulator that performs high-speed optical modulation.

In FIG. 14, the horizontal axis indicates the gap W between the center electrodes 4a, 4b, 4c and 4d and the ground electrodes 5a, 5b, 5c, 5d and 5e, and the left vertical axis indicates the center electrode 4a of the CPW traveling wave electrode. 4b shows electrical crosstalk with 4b, 4c, 4d (or high-frequency electrical signals Sa, Sb, Sc, Sd). Similarly to FIG. 12, the right vertical axis indicates the width G of the ground electrodes 5b, 5c, and 5d. 12 As mentioned in traveling wave characteristic impedance of the electrodes is 50Ω gap _{W 1} in the ground electrode 5b, 5c, the width G of 5d becomes narrow.

For example, when the width G of the ground electrodes 5b, 5c, and 5d is reduced to the same extent as the widths of the center electrodes 4a, 4b, 4c, and 4d, the electrical crosstalk between the center electrodes 4a, 4b, 4c, and 4d is −10 dB. Deteriorating to a degree or more, it is not preferable as a traveling wave electrode. Although electrical crosstalk in the gap W ₂ which is a lower characteristic impedance than 50Ω shown in FIG. 12 is the superior and -20dB or less, since the electric power reflection coefficient S ₁₁ is deteriorated, which is also traveling It is not preferable as a wave electrode.

As described above, the width that can be assigned to the input-side feed unit II (in a direction perpendicular to the length direction of the interaction optical waveguides 3a, 3b, 3c, and 3d and parallel to the surface of the z-cut LN substrate 1) The width to which the input side feed unit II can be allocated is limited. Therefore, the center electrode 4a described in the above, 4b, 4c, and you suppress the deterioration of the crosstalk and electric power reflection coefficient _{S 11} between 4d, narrow center electrode 4a, 4b, 4c, the width S of 4d It is possible.

However, the center electrode 4a, 4b, 4c, the width S of 4d narrow (e.g., in Figure 15 and _{S 1)} Then, a high frequency electric signal Sa, as shown in FIG. 15, Sb, Sc, the propagation loss α of Sd large Therefore, as shown in FIG. 16, the power modulation index | m | ² of the LN optical modulator is significantly deteriorated with the frequency (f) (in other words, the 3 dB optical modulation band is narrowed), and 40 Gbit / s. As an LN optical modulator, it is extremely undesirable.

For example, the center electrode 4a, 4b, 4c, when a 5μm width S of 4d, suppress central electrode 4a, 4b, 4c, both electrical crosstalk and electrical power reflection coefficient _{S 11} between 4d to -20dB or less However, since the propagation loss of the high-frequency electrical signal in this feed portion increases remarkably, the 3 dB band as the LN optical modulator deteriorates to about 10 GHz, and it is difficult to apply to 40 Gbit / s optical transmission. .
JP-T-2004-516743 Japanese Patent No. 2126214

In order to suppress electrical reflection to an external circuit connected to the LN modulator, it is necessary to set the characteristic impedance (most preferably 50Ω) appropriate for the input side feed section. In order to realize this, it is necessary to widen the gap (W in FIG. 11) between the center electrode and the ground electrode. As a result, the width G of the ground electrodes 5b, 5c, and 5d shown in FIG. Become. In the prior art, the input side feed portion of the traveling wave electrode of the LN optical modulator has the same CPW structure for several central electrodes. That is, the equivalent refractive indexes of the high-frequency electrical signals propagating through the respective center electrodes are equal to each other, and as a result, electrical crosstalk is likely to occur between the high-frequency electrical signals. As described above, since the gap G between the center electrode and the ground electrode is widened, the width G of the ground electrode is narrowed. Therefore, a large electrical cross is generated between the high-frequency electrical signals propagating through the traveling wave electrode. There was a talk. Further, if the width G of the ground electrodes 5b, 5c, and 5d is increased in order to suppress this electrical crosstalk, the gap (W in FIG. 11) between the center electrode and the ground electrode is reduced. As a result, the characteristic impedance Z of the traveling-wave electrode is lowered significantly, the electrical power reflection coefficient S ₁₁ is deteriorated. Further, in order to suppress both the degradation of the characteristic impedance and the deterioration of the electrical crosstalk between the high-frequency electrical signals, the characteristic impedance of the traveling wave electrode is set close to 50Ω and the gap between the center electrode and the ground electrode If the width G of the ground electrodes 5b, 5c, and 5d shown in FIG. 11 is set wider than (W in FIG. 11), the gap W is inevitably narrowed. As a result, the center electrode of the traveling wave electrode The width of became narrower. When the width of the center electrode is narrowed, the propagation loss of the high frequency electric signal is increased. In other words, if a feed portion having a conventional planar CPW structure is designed to suppress both a reduction in characteristic impedance and a deterioration in electrical crosstalk between high-frequency electrical signals, the propagation loss of high-frequency electrical signals is large. Therefore, high-speed light modulation is difficult as an LN light modulator.

  The present invention has been made in view of such circumstances, and an object thereof is to provide an optical modulator that is small in size, has low electrical crosstalk and electrical power reflectivity of a high-frequency electrical signal, and has a wide optical modulation band. And

  In order to solve the above problems, an optical modulator according to claim 1 of the present invention includes a substrate, an optical waveguide formed on the substrate, a traveling wave electrode including a center electrode and a ground electrode for applying a voltage, The optical waveguide includes a plurality of interaction optical waveguides in an interaction portion, which is a region where a refractive index changes when a voltage is applied to the traveling wave electrode, and the center electrode and the ground electrode each A plurality of centers of the input side feed section, wherein the input side feed section electrodes connected to the traveling wave electrode of the interaction section are each composed of a plurality of center electrodes and ground electrodes. Of the plurality of high-frequency electrical signals propagating through the electrode and the ground electrode, at least two adjacent refractive indexes are different from each other.

  The optical modulator according to claim 2 of the present invention is characterized in that at least two of the plurality of gaps between the plurality of center electrodes and the plurality of ground electrodes in the input-side feed section are different from each other. To do.

  The optical modulator according to claim 3 of the present invention is characterized in that the magnitude relationship between at least two adjacent gaps among the plurality of gaps is switched in the longitudinal direction of the input side feed section.

  The optical modulator according to claim 4 of the present invention is characterized in that, among the plurality of center electrodes in the input-side feed section, at least two adjacent center electrodes have different widths.

  The optical modulator according to claim 5 of the present invention is such that, among the plurality of center electrodes in the input-side feed portion, the magnitude relationship regarding the width of at least two adjacent center electrodes is in the longitudinal direction of the input-side feed portion. It is characterized by being replaced.

  In the present invention, in the traveling wave electrode constituting the input side feed portion of the traveling wave electrode, by making the equivalent refractive index of the high frequency electrical signal propagating through the adjacent center electrode out of the plurality of center electrodes different from each other, The width of the ground electrode between the center electrodes can be narrowed while suppressing the electrical crosstalk between the center electrodes. And this means that the gap between the center electrode and the ground electrode can be widened, that is, by making the characteristic impedance as a traveling wave electrode approximately 50Ω, the electric power of the high-frequency electric signal returning to the external circuit is reduced. It means that reflection can be suppressed. Furthermore, since the width of the ground electrode of the input side feed portion can be reduced, the width of the center electrode of the input side feed portion can be increased. Accordingly, the propagation loss of the high-frequency electrical signal is reduced, and high-speed optical modulation can be realized.

  Hereinafter, embodiments of the present invention will be described. However, since the same reference numerals as those in the related art shown in FIGS. 9 to 16 correspond to the same functional units, description of the functional units having the same reference numerals is omitted here. To do.

(Embodiment)
FIG. 1 shows a top view of one embodiment of the optical modulator of the present invention. Since DQPSK is taken as an example of the optical modulation system, the structure of the optical waveguide is the same as that of the prior art shown in FIG.

  Further, corresponding to FIG. 11, FIG. 2 shows a cross-sectional view of the input side feed section III of FIG. 1 (perpendicular to the length direction of the interaction optical waveguides 3a, 3b, 3c, 3d and z-cut LN 1 is a cross-sectional view in a direction parallel to the surface of the substrate 1. FIG. 2 also shows the ground electrode omitted in FIG. In order to simplify the explanation, the Si conductive layer for suppressing temperature drift is omitted.

The principle of the present invention will be described in detail with reference to FIG. The center electrode 4 a ', 4b', 4c ', respectively S _a width of _{4d'', S b ', S} c', S d ', the center electrode 4a', 4b', 4c', 4d' the ground electrode 5a ′, 5b ′, 5c ′, 5d ′, and 5e ′ have gaps W _a and W _a ′, W _b and W _b ′, W _c and W _c ′, W _d and W _d ′, and ground electrodes, respectively. The widths of 5b ′, 5c ′, and 5d ′ are G _b , G _c , and G _d . Here, in order to simplify the explanation of the principle of the present invention, it is assumed here that W _a = W _a ′, W _b = W _b ′, W _c = W _c ′, and W _d = W _d ′. In this discussion, it goes without saying that at least one of W _a ≠ W _a ′, W _b ≠ W _b ′, W _c ≠ W _c ′, and W _d ≠ W _d ′ may hold.

In the present invention, it is essential that at least one of W _a ≠ W _b , W _b ≠ W _c , and W _c ≠ W _d holds. However, can effect is most exhibited as the present invention is when all these conditions are satisfied (in other words, high-frequency electric signal Sa in the present invention, Sb, Sc, the equivalent refractive index n _a of Sd _', n _{b ',} n c' _{'on, n a', n d ≠} n b ', n b' ≠ n c ', n c' but it is essential that any one of true of ≠ _{n d} ', as the present invention This is most effective when all of these conditions are met).

  Hereinafter, the principle of the present invention will be described. For the sake of clarity, attention is paid to the center electrodes 4a ′ and 4b ′ in the electrode structure of the input side feed portion shown in FIG. That is, consider the two CPW electrodes whose center electrode and ground electrode are 4a ′ and 5a ′, 5b ′, respectively, and 4b ′, 5b ′, and 5c ′, respectively. The same holds true for other CPW electrodes consisting of 4c 'and 5c', 5d ', and 4d', 5d 'and 5e', respectively.

The absolute value of the difference between the center electrode 4a' and 4b '' and _{n b} 'high frequency electric signals Sa and Sb of the equivalent refractive index _{n a} of propagating and in FIG. _{3 (| Δn ab | = |} n a'-n b The vertical axis represents the degree of coupling between the high-frequency electrical signals Sa and Sb when '|) is taken on the horizontal axis. The width _{G b} of the ground electrode 5b' was 30 [mu] m. As can be seen from this figure, as the difference between the equivalent refractive indexes n _a ′ and n _b ′ of the high-frequency electric signals Sa and Sb propagating through the center electrodes 4a ′ and 4b ′ increases, the high-frequency electric signals Sa and Sb The degree of coupling between them decreases rapidly.

The Figure 4 5 a 'and the center electrode 4a' the horizontal axis, the gap _{W a} and the center electrode 4b' with 5b' 5b ', the absolute value of the difference between the gap _{W b} of 5c' (| ΔW _ab | = | W _a −W _b |), and the vertical axis represents the absolute value (| Δn) of the difference between the equivalent refractive indices n _a ′ and n _b ′ of the high-frequency electrical signals Sa and Sb propagating through the center electrodes 4a ′ and 4b ′. _ab | = | n _a ′ −n _b ′ |). As can be seen from the figure, when the absolute value (| ΔW _ab | = | W _a −W _b |) of the difference between the gaps W _a and W _b increases, the equivalent refractive indexes n _a ′ and n of the high-frequency electric signals Sa and Sb are increased. The difference from _b ′ increases.

FIG. 5 shows the absolute value (| ΔW _ab | = |) of the difference between the gap W _a between the center electrodes 4a ′ and 5a ′ and 5b ′ and the gap W _b between the center electrodes 4b ′ and 5b ′ and 5c ′ on the horizontal axis. W _a −W _b |), and the vertical axis represents the crosstalk between the high-frequency electrical signals Sa and Sb. As shown in FIG. 4, the absolute value of the difference between the gap _{W a} and _{W b (| ΔW ab | =} | W a -W b |) when increases, the equivalent refractive index of the high frequency electric signals Sa and Sb _{n a} The absolute value of the difference between ′ and n _b ′ (| Δn _ab | = | n _a ′ −n _b ′ |) increases, and as a result, as shown in FIG. 3, the high-frequency electric signals Sa and Sb The degree of coupling between them becomes small, and finally, crosstalk between these high-frequency electric signals Sa and Sb is suppressed.

FIG. 6 shows a cross-talk between the high-frequency electrical signals Sa and Sb in the case of a variable width G _b of the ground electrode 5b '. As can be seen from the figure, when the gaps W _a and W _b are equal (that is, 65 μm each in the prior art), and when the width G _b of the ground electrode 5b ′ is small, the cross between the high-frequency electrical signals Sa and Sb since talk is degraded, it is necessary to increase the width G _b of the ground electrode 5b '. However, as in the present invention, by making the gaps W _a and W _b different (that is, by making the equivalent refractive indexes n _a ′ and n _b ′ of the high-frequency electric signals Sa and Sb different), the high-frequency electric signal Sa. And Sb can be suppressed. In this example, the gaps W _a and W _b were set to 65 μm and 90 μm, respectively.

As described above, it is possible compared to the prior art to reduce the width G _b of the ground electrode 5b ', it is possible not only to widen the gap W _a and W _b, the center electrode 4 a' (or 4b It is also possible to increase the width S _a ′ (or S _b ′, S _c ′, S _d ′) of ′, 4 _c ′, 4 _d ′). Therefore, as shown as S ₁ ′ in FIG. 7, by using a wide center electrode, a high-frequency electric signal (here, the high-frequency electric signal Sa is taken up as compared with the conventional technique shown as S ₁ in FIG. 15 will be described. The propagation loss can be greatly reduced. When the width of the center electrode is increased (for example, S ₁ ′ = 45 μm) according to the present invention, the propagation loss of the high-frequency electric signal is reduced, and therefore, as shown in FIG. 8, as compared with the prior art shown in FIG. A greatly improved frequency characteristic of the modulation index can be realized (ie the 3 dB band of modulation is improved).

(Each embodiment)
In the above description, the DQPSK optical modulator has been described as an example. However, since the present invention is effective for an LN optical modulator having a plurality of center electrodes in the input side feed section, the DPSK is not limited to DQPSK, but includes a single Mach-Zehnder optical waveguide, It goes without saying that the present invention can also be applied to other phase modulation systems having more Mach-Zehnder optical waveguides than DQPSK, and further to a two-electrode type intensity modulator.

  Further, the present invention is also applicable to a so-called ridge structure in which a z-cut LN substrate is etched and ridge-processed by sandwiching all the center electrodes of the input side feed portion in the traveling wave electrode. Further, only a part of the center electrode may be obtained by combining the ridge type and the planar type electrode structure by ridge processing the z-cut LN substrate in the vicinity thereof. In addition, regarding the structure of the planar type, the ridge type, or the combination of the planar type and the ridge type, the thickness of the buffer layer below each center electrode may be partially made different at the input side feed portion. Since the difference in the equivalent refractive index of the high frequency electrode propagating through the traveling wave electrode of the input feed portion becomes large, electrical crosstalk can be further suppressed.

In general, when the equivalent refractive indexes n _a ′ and n _b ′ of the high-frequency electric signals Sa and Sb propagating through the center electrodes 4a ′ and 4b ′ are made different, the characteristic impedance as the traveling wave electrode is also different.

  For example, when the characteristic impedance of the input side feed portion of the traveling wave electrode is set to be somewhat lower than 50Ω, the gap between the center electrode and the ground electrode can be narrowed, so that the width of the ground electrode can be set wide. It becomes possible, and the crosstalk between the high frequency electric signals can be improved. Conversely, if the characteristic impedance of the input-side feed part of the traveling wave electrode is set higher than 50Ω, the current value of the high-frequency electrical signal propagating through the input-side feed part will be reduced, so that attenuation due to Joule heat will be reduced. As a result, high-speed optical modulation becomes possible.

Furthermore, the buffer layer may be partially thickened at the input side feed section. Alternatively, even if the equivalent refractive indexes n _a ′ and n _b ′ are made different by combining a planar type and a ridge type as described above, by setting the width of each center electrode and each gap appropriately, It is also possible for the characteristic impedances of the electrodes to be approximately the same.

  Furthermore, until now, in the input-side feed part III of FIG. 1, the magnitude relationship of the equivalent refractive index of the high-frequency electrical signal propagating through each center electrode may be switched in the propagation direction of the high-frequency electrical signal (longitudinal direction of the traveling wave electrode). It has been described that there is no such thing, but it goes without saying that the magnitude relationship of each equivalent refractive index may of course be switched in the propagation direction of the high-frequency electrical signal.

In other words, the widths S _a ′, S _b ′, and S of the center electrodes 4a ′, 4b ′, 4c ′, and 4d ′ of FIG. _c ′, S _d ′, gap widths between the central electrodes 4a ′, 4b ′, 4c ′, 4d ′ and the ground electrodes 5a ′, 5b ′, 5c ′, 5d ′, 5e ′, W _a , W _a ′, W _b , W _b ′, W _c , W _c ′, W _d , and W _d ′ have been described as having no structural change in the propagation direction of the high-frequency electrical signal. It can also be said that it may be switched in the propagation direction of the high-frequency electrical signal in part III. The fact that the magnitude relationship may be changed in the propagation direction of the high-frequency electric signal in the input side feed section III is true for the center electrodes 4a ′, 4b ′, 4c ′, and 4d ′.

Further, although the z-cut LN substrate has been described, an LN substrate of other cuts such as an x-cut, a y-cut, or a cut obtained by mixing them may be used, or other substrates such as a semiconductor substrate may be used. Although the buffer layer has been described as SiO ₂ , it goes without saying that other materials such as Al ₂ O ₃ , SiN, or SiO _X may be used.

  As the electrode configuration, a configuration using a CPW electrode having a symmetrical structure has been described. However, a CPW electrode having an asymmetrical structure may be used. good.

  As described above, according to the present invention, it is possible to provide an optical modulator that is significantly improved with respect to characteristic impedance, that is, electrical power reflectivity, and electrical crosstalk, and also propagation loss of high-frequency electrical signals, that is, optical modulation band. .

1 is a top view including a traveling wave electrode constituting an optical modulator according to an embodiment of the present invention. Cross section of the input side feed section in FIG. The figure explaining the principle of this invention The figure explaining the principle of this invention The figure explaining the principle of this invention The figure explaining the principle of this invention The figure explaining the principle of this invention The figure explaining the principle of this invention Top view showing configuration of optical waveguide for DQPSK optical modulator according to prior art Top view including traveling wave electrodes for a prior art DQPSK optical modulator Cross section of the input side feed section in FIG. Diagram explaining the problems of the prior art Diagram explaining the problems of the prior art Diagram explaining the problems of the prior art Diagram explaining the problems of the prior art Diagram explaining the problems of the prior art

Explanation of symbols

1: z-cut LN substrate (substrate)
2, 9: SiO ₂ buffer layer 3, 30: Mach-Zehnder optical waveguide (optical waveguide)
3a, 3b, 3c, 3d: interaction optical waveguides 4a, 4b, 4c, 4d, 4a ′, 4b ′, 4c ′, 4d ′, 7: center electrodes 5a, 5b, 5c, 5d, 5e, 5a ′, 5b '5c', 5d ', 5e': Ground electrode

Claims (5)

  A substrate, an optical waveguide formed on the substrate, and a traveling wave electrode composed of a center electrode and a ground electrode for applying a voltage, the optical waveguide being refracted by applying a voltage to the traveling wave electrode The interaction portion, which is a region where the rate changes, includes a plurality of interaction optical waveguides, and the center electrode and the ground electrode each include a plurality, and an input side connected to the traveling wave electrode of the interaction portion In an optical modulator in which each of the electrodes of the feed section is composed of a plurality of center electrodes and ground electrodes, a plurality of equivalent refractive indexes of a plurality of high-frequency electric signals propagating through the plurality of center electrodes and the ground electrodes of the input-side feed section An optical modulator characterized in that at least two adjacent ones have different structures.   The optical modulator according to claim 1, wherein at least two of the plurality of gaps between the plurality of center electrodes and the plurality of ground electrodes in the input-side feed unit are different from each other.   3. The optical modulator according to claim 2, wherein a size relationship between at least two adjacent gaps among the plurality of gaps is switched in a longitudinal direction of the input-side feed unit.   2. The optical modulator according to claim 1, wherein among the plurality of center electrodes in the input-side feed section, at least two adjacent center electrodes have different widths.   5. The size relationship of the widths of at least two adjacent center electrodes among the plurality of center electrodes in the input-side feed portion is switched in the longitudinal direction of the input-side feed portion. The light modulator described.

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