JP2008295220A - Sensorless control unit of permanent magnet synchronous electric motor - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To solve the problem of step-out in a current lead-in method that exists in a position sensorless control system of an AC synchronous electric motor. <P>SOLUTION: A high-frequency voltage coordinate conversion part coordinate-converts a high-frequency voltage component from a single-phase high-frequency voltage generating part, and the converted high-frequency voltage is combined with a voltage command from a current control part and is outputted to a reverse rotating coordinate converting part as a voltage command value. A single-phase axis phase signal is detected from detected current components of two axes, and a phase signal of high-frequency voltage is generated. The phase signal is set to be a phase signal at the coordinate conversion and is outputted to a stabilized frequency correcting part, and a correction signal is calculated, and a sum with a frequency command is set to be a reference phase signal. <P>COPYRIGHT: (C)2009,JPO&INPIT

Description

  The present invention relates to a variable speed control using a synchronous motor or a generator position sensorless / speed sensorless using a permanent magnet as a magnetic field source, and more particularly to a sensorless control device that can be driven even in a low speed region.

  Regarding the sensorless control method, Patent Documents 1 to 3 and the like are publicly known. In Patent Document 1, as shown in FIG. 11, a single-phase voltage harmonic voltage Vdc is added in the rotating coordinate system (32a, 32b), and the generated current is converted into a parallel component (Idc) and an orthogonal component (Iqc). )). Then, a phase error detector 33 detects a phase error (θ) of a voltage high-frequency current with respect to a high-frequency voltage from a component obtained by separating the current into two axes, and corrects the estimated magnetic pole phase θ0. Thus, the phase error is estimated after the single-phase high-frequency voltage is input and the high-frequency current is detected on the same coordinates as the single-phase component. In addition, 30 is an electric motor and 31 is an inverter. Further, by utilizing the axial errors of the single-phase high-frequency voltage component and the single-phase high-frequency current component of FIG. 11, an integrator 34 is provided as shown in FIG. 12, and the magnetic pole phase is estimated by performing integration using this integrator 34. Yes.

This method uses a difference in inductance between the d-axis and the q-axis (magnetic saliency) and cannot separate the N-pole and S-pole on the d-axis. For this reason, the phase of the N pole is estimated in advance by another method, and when the motor rotates with respect to the initial phase, the phase is sequentially estimated from the phase difference between the high-frequency voltage component and the high-frequency current component. The magnetic pole phase is corrected to follow.
Also. It is composed of a block of a function of injecting a single-phase high-frequency voltage into an appropriate phase, a function of detecting a generation axis of the generated single-phase high-frequency phase, and two parts for estimating the magnetic pole axis from these phase differences. It is a feature.
In the method of estimating the magnetic pole phase directly by the integrator 34 in FIG. 12, a deviation occurs in the magnetic pole estimation phase when the speed is accelerated in a ramp shape, and step-out easily occurs due to this magnetic pole estimation error.

  Patent Document 2 and Patent Document 3 not only propose a specific high-frequency separation method, but also add an integral operation to integrate the phase error component of the high-frequency voltage and current and calculate the speed once. Then, it is integrated to calculate the estimated phase. By adding this speed estimation integral, an estimation error does not occur as a steady-state deviation of the magnetic pole phase even with respect to a ramp change in speed, so that step-out is difficult.

In addition to the high-frequency method described above, a method that has been used for stepping motors, etc., is a method that forcibly generates a constant amplitude current and forcibly rotates the current phase to force the magnetic pole to follow. However, here, this method is called a current drawing method.
An example of a control block diagram of the current drawing method is shown in FIG.
In the following description, it is assumed that the rotating field type synchronous motor uses a permanent magnet as a field, and the armature is assumed to be a stator or fixed coordinates (a, b axes), and the magnetic pole side is the rotor. Or it is called a rotation coordinate (d, q axis). Further, the phase of the current command generated for current drawing is defined as the γ axis, and the axis orthogonal thereto is defined as the δ axis.
Further, in order to distinguish a fundamental wave component used for current drawing from a high frequency component voltage or current, a subscript “h” is added to the variable of the high frequency component.

In FIG. 13, reference numeral 1 denotes a γ-axis current command which is a current amplitude command value, and the current amplitude command value | I1 ^* | is set to the γ-axis component (iγ ^* ) of the orthogonal two-axis components of the rotating coordinate system. . Reference numeral 2 denotes a δ-axis current command which is a current amplitude command value, and sets zero to the δ-axis component (iδ ^* ) of the orthogonal two-axis components of the rotating coordinate system.
The amplitude command value (iγ ^* , iδ ^* ) of each current and the biaxial current component (iγ, iδ) from the rotation coordinate conversion unit 12 are input to the current control unit 5, and the current control unit 5 receives the current command. Output voltage (Vγ ^* , Vδ ^* ) is output so that the actual current follows. The current control unit 5 is generally configured by a difference calculation capital between current command and current detection, proportional integral (PI) control, and the like.

Reference numeral 3 denotes a current rotation speed command, which sets a frequency (speed) command ω1 ^* for forcibly rotating the drawing current. Reference numeral 4 denotes a phase integrator that outputs a reference phase θ that integrates the speed command 3 to generate a current. This θ is a phase angle with respect to the armature winding.
The reference phase θ is output to the reverse rotation coordinate conversion unit 6 together with the output voltages (Vγ ^* , Vδ ^* ) from the current control unit 5, and the reverse rotation coordinate conversion unit 6 uses voltage commands ( Vγ ^* , Vδ ^* ) is converted into a biaxial voltage component (Va ^* , Vb ^* ) in a fixed coordinate system by performing reverse rotation coordinate conversion with the output phase θ.

Reference numeral 7 denotes a two-phase / three-phase conversion unit, which is a three-phase AC voltage (Vu ^* , Vv ^* , Vw ^* ) having a 120 ° phase difference obtained by converting the orthogonal biaxial voltage component of fixed coordinates, which is the output of the reverse rotation coordinate conversion unit 6. And the PWM amplifier 8 performs power amplification using a pulse width modulation (PWM) method.
Here, the motor 9 is assumed to be a PM motor using a permanent magnet as a field source. The target is one having magnetic saliency with different inductance between the d-axis and the q-axis. The input current to the motor 9 is detected by the current detector 10, and the three-phase detection current (iu, iv, iw) is output to the three-phase / two-phase converter 11 to generate the orthogonal biaxial current component (ia, ib). Is converted to The detected current may not actually be three phases, but two phases can be detected and the remaining one phase can be estimated from these two phases by calculation.
The rotation coordinate conversion unit 12 performs rotation coordinate conversion on the orthogonal biaxial current component (ia, ib), which is the output of the three-phase / two-phase conversion unit 11, with the phase θ of the integrator 4, and the biaxial current in the rotation coordinate system. Convert to components (iγ, iδ). The detected current after conversion is used for current control of the current control unit 5.

The above is the configuration of the high current drawing method. With this configuration, a current having a current amplitude command value | I1 ^* | and a frequency (speed) command ω1 ^* is generated, and the rotor of the motor rotates following this current.
JP 7-245981 A JP2003-153582 JP2003-348896

(1) Problems with the high-frequency method The high-frequency method has two parts: a part that detects the phase error of the voltage and current of a single-phase high-frequency component, and a part that estimates the magnetic pole phase from the phase error information using an integrator. Explained that it exists. Here, since the high frequency method uses magnetic saliency in principle, there is no discrimination function between the N pole and the S pole. When a single-phase high-frequency voltage is injected into the d-axis, the magnetic pole phase is estimated (followed) by correcting the estimated phase with an integrator so that the generation phase of the high-frequency current matches the high-frequency voltage phase. There is a state where this phase error is zero in both the pole and the S pole.

Therefore, the phase of the N pole is first estimated by another method using magnetic saturation, etc., and the N pole and the S pole are distinguished from each other by always following this.
However, when noise is mixed in the current detection, an error occurs between the high-frequency current and the voltage phase difference, and the magnetic pole estimation integrator is mistakenly integrated by about 180 °, it converges to the S pole, and N A 180 ° estimated phase error occurs with respect to the pole. Once an estimation error occurs from the N pole to the S pole, the S pole is always recognized even if the N pole is estimated even if it operates normally.

  As described above, since the high frequency method has no NS determination function of the magnetic pole, if the estimation of the magnetic pole phase is stepped out, the S pole side is erroneously estimated as the d-axis. In this case, a current is generated in a direction opposite to 180 ° with respect to the current command assumed by the speed control system, and as a result, the generated torque of the motor also has a reverse polarity. Even if a forward torque command is given, torque in the reverse direction is generated in the motor and the rotational direction is also accelerated in the reverse direction (acceleration in the reverse direction at zero speed and deceleration in normal rotation).

Furthermore, since the difference between the speed detection and the distance command becomes large, the speed control system increases the torque command in the positive direction. The motor increases the reverse torque. The acceleration in the reverse direction increases and eventually runs away in the reverse direction. As described above, in the high frequency method, when a step-out occurs, the estimated magnetic pole may be switched between the N pole and the S pole, and the runaway may occur on the reverse side. In order to prevent this reverse runaway, the magnetic pole estimation gain must be set appropriately, and the high-frequency current used for measurement must be a large value to suppress error components such as noise.
(2) Problems with the current drawing method Since the current drawing method forcibly draws current, the N pole and the S pole are not erroneously determined. However, if the frequency command or the load fluctuates, vibration is likely to occur, and if the load torque becomes excessive, there is a problem of stepping out.
It is known that the magnetic pole phase becomes oscillating when there is no damper winding in the field core of the synchronous machine. Further, the step-out occurs when the load exceeds the maximum torque that can be generated by the current amplitude. In order to prevent step-out in consideration of this phase vibration, normally, only about 2/3 of the load torque can be applied to the maximum torque that can be generated by the set current command.

Comparing the two methods, since the high frequency method controls the current by estimating the magnetic pole phase, the ratio of the generated torque is large and no vibration is generated. In addition, there is an advantage that the speed decreases but does not step out during overload. However, if the magnetic pole estimation step out by mistake, there is a problem that reverse runaway occurs.
The current drawing method has a problem of stepping out during transient vibration or overloading, but it does not run out of control in reverse.
Accordingly, an object of the present invention is to provide a sensorless control device for a permanent magnet synchronous motor which improves these problems by combining the above two methods.

Claim 1 of the present invention is a control device for an AC synchronous motor,
The control device is provided with a single-phase high-frequency voltage generator, the high-frequency voltage component from the single-phase high-frequency voltage generator is coordinate-converted by a high-frequency voltage coordinate converter, the converted high-frequency voltage and the voltage command from the current controller Combined and output to the reverse rotation coordinate conversion unit as a voltage command value, and the coordinate-converted two-axis current detection component is input to the single-phase high-frequency current phase detection unit after passing through a high-frequency pass filter. A phase signal is detected, this single-phase axis phase signal is output to the magnetic pole phase estimation unit to generate a high-frequency voltage phase signal, and this phase signal is output to the high-frequency voltage coordinate conversion unit to generate a phase signal at the time of coordinate conversion In addition, the phase signal of the high frequency voltage is output to the stabilization frequency correction unit to calculate the correction signal, and the sum of the calculated correction signal and the frequency command is output to the reverse rotation coordinate conversion unit as a reference phase signal. It is obtained by characterized by being configured so.

  According to a second aspect of the present invention, the single-phase high-frequency voltage generation unit generates a high-frequency reference phase signal by integrating a high-frequency frequency command to generate a waveform signal including a high frequency, and the single-phase voltage is generated in the signal. This is characterized in that simple oscillation of voltage is calculated by multiplying the amplitude command.

According to a third aspect of the present invention, in the control device for an AC synchronous motor,
The control device is provided with a single-phase high-frequency voltage generator, the high-frequency voltage component from the single-phase high-frequency voltage generator is coordinate-converted by a high-frequency voltage coordinate converter, the converted high-frequency voltage and the voltage command from the current controller Combined and output to the reverse rotation coordinate conversion unit as a voltage command value, and the coordinate-converted two-axis current detection component is input to the single-phase high-frequency current phase detection unit after passing through a high-frequency pass filter. Detects a phase signal, outputs this signal to the magnetic pole phase estimation unit to generate a high-frequency voltage phase signal, and outputs this phase signal to the high-frequency voltage coordinate conversion unit to generate and generate a phase signal at the time of coordinate conversion The difference between the measured phase signal and the single-phase axis phase signal is output to a stabilizing frequency correction unit to calculate a correction signal, and the reverse rotation coordinate transformation is performed using the sum of the correction signal and the frequency command as a reference phase signal. It is obtained by characterized by being configured to output to.

  According to a fourth aspect of the present invention, the signal input to the high-frequency voltage coordinate conversion unit is an orthogonal two-component signal, one of which is a high-frequency voltage component from the single-phase high-frequency voltage generation unit, and the other one-axis The component is characterized by zero.

  According to a fifth aspect of the present invention, the output signal of the magnetic pole phase estimation unit is obtained by subtracting the voltage axis phase signal from the high frequency current axis phase signal that is the output of the single phase high frequency current phase detection unit, and multiplying this by a proportional gain. It is characterized by being an integrated signal.

As described above, according to the present invention,
(1) Since the current drawing method is used, the S pole is not erroneously estimated as the N pole due to disturbance components or the like unlike the high frequency method. Therefore, the phenomenon of reverse runaway does not occur.
(2) Even when the frequency command changes suddenly, the output frequency is automatically corrected so as not to step out, so the stability at the time of sudden change of the frequency command is improved.
(3) Even if the load changes suddenly and the motor speed changes suddenly, the output frequency is automatically corrected so as not to step out. From (2) and (3), not only vibration suppression but also transient fluctuation The effect of maintaining the stability by automatically correcting the frequency is obtained.
(4) If the magnetic pole estimation gain is not set appropriately in the high frequency method, a step out due to a response delay or a step out occurring due to an excessive gain occurs. Therefore, when adjusting while driving, the initial value must be adjusted appropriately. An inappropriate initial value may cause a step-out during gain adjustment.
On the other hand, in the present invention, the current drawing method is basically used, and it is stable if there is no speed command or sudden load change. The correction gain is initially set to zero, and may be increased little by little in accordance with distance fluctuations and load fluctuations. Therefore, the adjustment is easy, and the present invention is used in applications where there is no transient phenomenon. It can also be set not to apply.

The present invention gives priority to not running out of control when stepping out, and uses a current drawing method as a basic principle. A high-frequency method is also used in combination with this to add a function for suppressing vibration during transition.
In the conventional high frequency method, starting from the initial N-pole phase, and assuming that the previous estimated phase is always the N-pole, the phase correction is relatively performed. Could not be detected. Therefore, the change (corresponding to differentiation) of the magnetic pole position estimation of the high frequency method is calculated. By utilizing the principle that steady component information is lost when differentiation is performed, a method of combining correction with speed estimation by multiplying the differential amount of the magnetic pole thrust phase of the current method and the high-frequency method by gain is provided.

FIG. 1 is a block diagram of an overall control system showing an embodiment of the present invention. In this block diagram, the same or corresponding parts as in FIG. In FIG. 1, reference numeral 21 denotes a single-phase high-frequency voltage generator that calculates a voltage command to be injected in order to generate a high-frequency current according to a high-frequency method. The single-phase high-frequency voltage generating portion 21 includes an integrating means for generating a high frequency reference phase theta _hv ^* by integrating the high-frequency frequency command omega _hv ^*, the function generating means and multiplying means. As a means for generating a waveform including a specific high frequency, an example of a cosin function is shown here. The high-frequency waveform is multiplied by a single phase voltage amplitude command v _hv ^* to calculate a single oscillation of the voltage.

  Reference numeral 22 denotes a high-frequency voltage coordinate conversion unit which applies the single-frequency high-frequency voltage component, which is the output of the single-phase high-frequency voltage generation unit 21, to match the designated magnetic pole phase. In the input of the coordinate conversion unit 22, one axis component of the orthogonal two-axis component is a single vibration voltage component, and the other axis is zero. The phase of the coordinate conversion unit 22 uses the magnetic pole estimation phase φd (= ^ φd).

FIG. 2 shows a vector locus of the voltage component of the single-phase alternating current and a locus of the single-phase alternating current generated thereby. FIG. 2 shows the phase based on the U phase of the armature winding. The magnetic pole axis rotating at the rotational speed ω ₁ ^* is the d-axis, and the axis orthogonal thereto is the q-axis.
The current vector axis of the current command is the γ axis, and its orthogonal axis is the δ axis. Current vector _{i 1 (iγ -0, iδ -0} ) generated by the current command and the voltage _{v 1 (vγ -0, vδ -0} ) at that time with respect to high-frequency voltage (vγ _-h ^*, vδ _-h ^* ) Is superimposed, and the high-frequency current (iγ _-h , iδ _-h ) is also represented as being superimposed on the current.

  FIG. 3 is an enlarged view in which the centers of the high-frequency components are made coincident in order to clarify the phase relationship of the high-frequency components. In FIG. 3, in order to clarify the phase relationship, the high frequency component of the voltage is shown in parallel translation so as to coincide with the center of the current high frequency. The phase angle from the current generation axis γ to the axis for injecting the single-phase AC voltage is φv, and a single-phase current having the same frequency is generated by this high-frequency voltage, and the generation phase of this high-frequency current is φi. The phase from the γ axis to the magnetic pole axis is φd.

Reference numeral 23 denotes a high frequency voltage superimposing unit that adds (superimposes) the high frequency voltage (vγ− _h ^* , vδ− _h ^* ) to the output voltage (vγ ^* , vδ ^* ) of the current control unit 5. Reference numeral 24 denotes an integrator for calculating a current command phase, and the current command is given by a DC value command value | I ₁ ^* | and a frequency command ω ₁ ^* . The frequency command ω ₁ ^* is added to the output signal of the stabilization frequency correction unit 30, the corrected frequency ω ₁ of the addition result is integrated by the integrator 24, and the voltage command is converted into the coordinate system of the armature winding. Phase θ is generated. Based on the phase as a reference, the coordinate conversion unit 6 converts the coordinate and outputs the voltage to the motor 9 by amplifying the voltage by the PWM modulation method or the like via the 2-phase / 3-phase conversion unit 7 and the PWM amplification unit 8. To do.

  Reference numeral 25 denotes a high-frequency rejection filter, and since the current detector 10 obtains a three-phase component armature current (iu, iv, iw), this is determined by the three-phase / two-phase converter 11 and the reference phase θ of the current command. The rotation coordinate conversion unit 12 performs conversion to obtain (iγ, iδ). Since the current detection does not require the high-frequency component of current detection, the high-frequency removal filter 25 removes the high-frequency component from (iγ, iδ) including the fundamental wave component and the high-frequency component, and only the fundamental wave current component is obtained. .

  Reference numeral 26 denotes a high-frequency pass filter. Contrary to the high-frequency rejection filter 25, the fundamental wave current component is not necessary for estimation of the magnetic pole phase, and therefore the high-pass filter 26 is used to remove the fundamental wave component. As a specific configuration example of the high-frequency rejection filter 25 and the high-frequency pass filter 26, a method using a moving average is described in Patent Document 2, and the present invention does not limit the invention portion of the filter portion. Here, detailed description is omitted, and an example of a discrete system is shown in FIGS. 4, 5, and 6.

FIG. 4 shows an example of 25 and 26 filter portions, where 1 / Z indicates a one-sample delay in digital calculation, and Σ indicates the addition of all inputs. Here, only one phase of the γ axis is shown.
When one period of the high frequency component is handled by 8 discrete points, the center of the high frequency can be obtained by a moving average of 8 points. By subtracting this center value from the detected current value, a high frequency component can be extracted. The component from which the high-frequency component has been removed may use the 8-point moving average, but the detection delay when the current changes is large. Therefore, it is assumed that the high frequency component of the previous cycle (8 samples before) and the current high frequency component are the same, and the phase change is small, and the high frequency component of the previous cycle is subtracted from the detected value. As a removal component. In this way, the detection waste time is reduced when the value of the fundamental wave component changes.

  A single-phase high-frequency current generation phase detector 27 detects a single-phase axis phase from the output of the high-frequency pass filter 26, which is a two-axis high-frequency single-phase AC current component. As this detection method, as shown in FIG. 5, a high-frequency component of each axis is obtained by using a sin function delayed by 90 degrees, and converted into a phase by using an arctan function from the result of moving average for one cycle. do it. Further, as shown in FIG. 6, the same calculation can be performed by using a function that outputs 1 if the high-frequency phase is 0 to π, and -1 if π to 2π, instead of the sin function. Also, the moving average can be a half cycle (in the case of FIG. 6, a moving average of 4 samples), but a ripple-like error component may be generated due to positive and negative asymmetry, so that one cycle as shown in FIG. A moving average of minutes is preferred. This moving average can be substituted by an IIR type digital filter. After that, the phase is obtained by the arctan function in the same manner as described above.

  Reference numeral 28 denotes a magnetic pole phase estimation unit. When the relationship between the direct-axis inductance Ld and the horizontal-axis inductance Lq is (Ld <Lq) due to the magnetic saliency of the motor, the generation phase of the current high frequency of the phase detection unit 27 is The high-frequency single-phase current phase exists between the high-frequency single-phase voltage axis and the actual magnetic pole axis. Therefore, the voltage axis phase φv is subtracted from the high-frequency current axis phase φi, multiplied by a proportional gain, and then integrated. This integrated output is set as a new high-frequency voltage output phase φv. Thus, by estimating the magnetic pole phase and feeding back to the generation phase of the high-frequency voltage, the voltage phase and the current phase coincide with each other. At this time, the magnetic pole axis φd and the phase φv of the high-frequency voltage coincide.

  Reference numeral 29 denotes a magnetic pole estimation phase differential calculation unit. As described above, the magnetic pole estimation phase unit 28 sometimes converges over the N and S poles due to current detection noise or the like. Accordingly, the phase change from which the deviation component is removed is obtained by differentiating the magnetic pole phase signal from the magnetic pole estimation phase unit 28.

Reference numeral 30 denotes a stabilization frequency correction unit. Considering the case of forward rotation (ω ₁ ^* > 0), since the amount of change in the estimated magnetic pole phase has been obtained by the differential operation unit 29, frequency correction proportional to this is added. When the load torque increases, the magnetic pole phase of the motor is delayed with respect to the current vector. When this delay increases, the phase change component of the differential calculation unit 29 becomes negative. Therefore, the output frequency is lowered, and the operation is performed so as to suppress the increase in the difference Φd between the current phase θ and the magnetic pole phase. Here, in addition to multiplying the derivative of the magnetic pole estimation phase by a proportional gain, a limiter that suppresses excessive components and a low-pass filter that removes high-frequency component noise are applied to the frequency correction calculation as necessary. Good.
According to the first embodiment configured as described above, a magnetic pole phase is estimated using a high-frequency voltage or current, and a frequency correction is performed using a differential component of the estimated phase, whereby a current is supplied to a synchronous motor without a damper winding. Even if the pull-in method is applied, vibration can be suppressed.

In the first embodiment, the estimation result of the magnetic pole phase is differentiated by the magnetic pole estimation phase differentiation calculation unit 29, but is further differentiated after the (φi 1 φv) component is integrated by the magnetic pole phase estimation unit 28. Since these two integral and differential components can be canceled, equivalent conversion can be made to a block diagram as shown in FIG. Others are the same as Example 1, and the principle is the same.
In the first embodiment, the magnetic pole estimation phase is differentiated, but according to this embodiment, the amount of calculation is reduced by using the internal data of the magnetic pole estimation calculation.

FIG. 8 is a characteristic diagram as a result of simulating the operation by the conventional current drawing method shown in FIG. This simulation is performed for comparison with the present invention, and is a case where K _ωφ in the stabilization frequency correction unit 30 is set as K _ωφ = 0.

<1> Evaluation condition (no current compensation method and compensation function of the present invention)
(A) The amplitude of the current command is set to 100% of the motor rating.
(B) For the frequency of the current command, the speed command is changed to 0% to 5% between 0s and 0.1s, and the speed command is changed to 5% to 10% between 1.0s and 1.1s.
(C) No load torque is applied between 0 s and 2 s of load fluctuation, and 70% of load torque is applied stepwise after 2 s.

<2> Result (a) The motor speed is oscillating, and when a load is applied in 2 s, the motor distance decreases and step-out occurs.
(B) The phase difference φd between the current vector and the d-axis is also oscillating, and stepped out when loaded.
(C) The generated torque is also oscillating like the speed and current described above, and steps out when a load is applied.

FIG. 9 is a characteristic diagram by simulation when the present invention is applied (the effect of improving the stability when the frequency command and the load suddenly change). In the system of the second embodiment, the control function is set by setting _Kωφ to an effective value. This is the case when it is operated.

<1 '> Evaluation conditions (current drawing method: the compensation function of the present invention is valid, otherwise the same conditions as in FIG. 8)
(A ') The amplitude of the current command is set to 100% of the motor rating.
(B ′) The frequency of the current command is changed from 0% to 5% between 0s and 0.1s, and the speed command is changed between 5% and 10% between 1.0s and 1.1s.
(C ′) No load torque between load fluctuations 0 s and 2 s. After 2 s, a load torque of 70% is applied stepwise.

<2 ′> Result (a ′) Even if the frequency command ω ₁ ^* changes, the current output frequency ω ₁ suppresses acceleration if there is a response delay in the motor speed. At the time of load, the current output frequency ω ₁ automatically decreases the frequency following the decrease in the motor distance to prevent the step-out. Moreover, when time stands, it will return to the original speed.
(B ′) The phase difference φd from the current d-axis is stable without vibration. There is no vibration even when loaded.
(C ′) The generated torque is stable similarly to the above speed and current, and torque corresponding to acceleration and load can be generated.

FIG. 10 is a characteristic diagram by simulation for confirming the effect when the present invention is applied (when the magnetic pole of the high frequency method due to detection noise converges NS incorrectly)
The response when a disturbance to the magnetic pole phase estimation is input in the portion indicated by ⇒ of time 1 s is shown.

<3> Evaluation conditions (current draw-in method: compensation function is valid according to the present invention, the same control as FIG. 9)
(A) The amplitude of the current command is set to 100% of the motor rating.
(B) The frequency of the current command is changed from 0% to 5% between 0s and 0.1s, and then set to a constant value.
(C) No load torque is applied between 0 s and 2 s of load fluctuation, and 70% load torque is applied stepwise only during the period of 2 s to 3 s.
(D) Injection of phase estimation disturbance in a period of 1.0 to 1.01 s.
The magnitude of the disturbance is set so that the magnetic pole phase changes by exactly 180 ° (π rad) when integrated.

<4> Result (a) At the time of 1 s, the magnetic pole estimation phase is inverted by 180 ° due to the magnetic pole phase disturbance. That is, although the N pole was estimated, a malfunction has occurred in which the S pole is suddenly estimated due to a disturbance.
(B) However, although the motor speed ω _r and the phase −φd between the current and the d axis have transient fluctuations, they have returned to the original stable state.
(D) The generated torque is also stable in this vicinity.

The magnetic pole estimation phase Φv is suddenly changed by 180 ° due to the disturbance to the magnetic pole estimation. In the case of the conventional high-frequency method, this causes erroneous detection of the N pole and the S pole, and reverse runaway occurs.
However, according to the present invention, as shown in FIG. 10, there is a fluctuation component due to noise (⇒ portion), but no reverse runaway has occurred, and stability can be improved.
Summarizing the results shown above, the following items are common effects in the first and second embodiments.

(1) Since it is a current pull-in method that prevents reverse rotation, there is no possibility that the S pole is erroneously estimated as the N pole due to a disturbance component or the like, unlike the high frequency method. Therefore, the phenomenon of reverse runaway does not occur.

(2) Stability when frequency command suddenly changes Even when the frequency command suddenly changes, the output frequency is automatically corrected so as not to step out. (3) Stability during sudden load changes Even if the load changes suddenly and the motor speed changes suddenly, the output frequency is automatically corrected so as not to step out. From (2) and (3), not only vibration suppression but also the effect of maintaining the stability by automatically correcting the frequency at the time of transient fluctuation is obtained.
If a load torque exceeding the maximum torque that can be generated with the set current command is applied, the step-out will occur. When the high-frequency method and vector control are used as in Patent Documents 1 to 3, the speed is reduced even when overloaded, but no step-out occurs. However, if it is within the maximum torque, it is possible to realize a control system that does not vibrate and is transiently stable, and can be dealt with by setting a larger current.

(4) Simplification effect of gain adjustment In the high frequency method, if the magnetic pole estimation gain is not set appropriately, a step out due to a response delay or a step out due to an excessive gain occurs. Therefore, when adjusting while driving, the initial value must be adjusted appropriately. With an inappropriate initial value. There may be a step-out during gain adjustment.
On the other hand, in the present invention, the current drawing method is basically used, and it is stable if there is no speed command or sudden load change. The correction gain is initially set to zero, and may be increased little by little according to the distance fluctuation or load fluctuation. Therefore, the adjustment is easy, and it is possible to set so that the present invention is not applied in an application without a transient phenomenon.

The block diagram which shows embodiment of this invention. Voltage / current vector diagram of fundamental wave component and high frequency component. The locus enlarged view of a single phase high frequency component. The block diagram of a high frequency removal filter and a low-pass filter. The block diagram of the generation | occurrence | production phase detection part of a high frequency single phase electric current. The block diagram of the generation | occurrence | production phase detection part of another high frequency single phase electric current. The block diagram which shows other embodiment of this invention. FIG. 6 is a characteristic diagram of a conventional current drawing method by simulation, where (a) is a frequency component waveform, (b) is a phase component waveform, and (c) is a motor torque waveform. In the characteristic diagram of the present invention by simulation, (a) is a frequency component waveform, (b) is a phase component waveform, and (c) is a motor torque waveform. In the present invention, (a) is a frequency component waveform, (b) is a phase component waveform, and (c) is a motor torque waveform. The block diagram of the conventional sensorless control apparatus. The block diagram of the other conventional sensorless control apparatus. The block diagram of the sensorless control apparatus by the conventional electric current drawing method.

Explanation of symbols

DESCRIPTION OF SYMBOLS 1 ... γ current command 2 ... δ current command 3 ... Frequency command 4 ... Integrator 5 ... Current control unit 6 ... Reverse rotation coordinate conversion unit 7 ... 2-phase / 3-phase conversion unit 8 ... PWM amplification unit 9 ... Motor 10 ... Current Detector 11 ... Three-phase / two-phase converter 12 ... Rotating coordinate converter 21 ... Single-phase high-frequency voltage generator 22 ... High-frequency voltage coordinate converter 23 ... High-frequency voltage superposition unit 24 ... Current command phase calculation integrator 25 ... High frequency Removal filter 26 ... High-frequency pass filter 27 ... Single-phase high-frequency current generation phase detection unit 28 ... Magnetic pole phase estimation unit 29 ... Magnetic pole estimation phase differential operation unit 30 ... Stabilization frequency correction unit

Claims (5)

In the control device for an AC synchronous motor,
The control device is provided with a single-phase high-frequency voltage generator, the high-frequency voltage component from the single-phase high-frequency voltage generator is coordinate-converted by a high-frequency voltage coordinate converter, the converted high-frequency voltage and the voltage command from the current controller Combined and output as a voltage command value to the reverse rotation coordinate conversion unit,
The coordinate-converted two-axis current detection component is input to a single-phase high-frequency current phase detection unit after passing through a high-frequency pass filter to detect a single-phase axis phase signal, and this single-phase axis phase signal is input to a magnetic pole phase estimation unit. To generate a phase signal of the high-frequency voltage, output this phase signal to the high-frequency voltage coordinate conversion unit to be a phase signal at the time of coordinate conversion, and output the phase signal of the high-frequency voltage to the stabilization frequency correction unit A sensorless control device for a permanent magnet synchronous motor, wherein a correction signal is calculated, and a sum of the calculated correction signal and the frequency command is output to the reverse rotation coordinate conversion unit as a reference phase signal. The single-phase high-frequency voltage generation unit integrates a high-frequency frequency command to generate a high-frequency reference phase signal to generate a waveform signal including the high frequency, and multiplies this signal by a single-phase voltage amplitude command to 3. A sensorless control device for a permanent magnet synchronous motor according to claim 2, wherein simple vibration is calculated. In the control device for an AC synchronous motor,
The control device is provided with a single-phase high-frequency voltage generator, the high-frequency voltage component from the single-phase high-frequency voltage generator is coordinate-converted by a high-frequency voltage coordinate converter, the converted high-frequency voltage and the voltage command from the current controller Combined and output as a voltage command value to the reverse rotation coordinate conversion unit,
The coordinate-converted two-axis current detection component is input to a single-phase high-frequency current phase detection unit after passing through a harmonic pass filter to detect a single-phase axis phase signal, and this signal is output to the magnetic pole phase estimation unit. Generates a phase signal of the high frequency voltage, outputs this phase signal to the high frequency voltage coordinate conversion unit as a phase signal at the time of coordinate conversion, and stabilizes the difference between the generated phase signal and the single phase axis phase signal A sensorless permanent magnet synchronous motor characterized in that a correction signal is calculated by outputting to a frequency correction unit, and the sum of the correction signal and the frequency command is output as a reference phase signal to the reverse rotation coordinate conversion unit. Control device. The signal input to the high-frequency voltage coordinate conversion unit is an orthogonal two-component signal, one of which is a high-frequency voltage component from the single-phase high-frequency voltage generation unit, and the other one-axis component is zero. 4. A sensorless control device for a permanent magnet synchronous motor according to claim 1. The output signal of the magnetic pole phase estimation unit is a signal obtained by subtracting the voltage axis phase signal from the high frequency current axis phase signal which is the output of the single phase high frequency current phase detection unit and multiplying it by a proportional gain and integrating it. The sensorless control device for a permanent magnet synchronous motor according to claim 1.

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