JP2008167557A - Motor control method - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide a motor control method which can control a complex current generated by a rectangular waveform voltage, can make the voltage utilization rate improve, and can increase the output of a motor even by using a normal two-level inverter. <P>SOLUTION: In the control method of the AC motor which comprises at least one rotor, and is driven by superimposing an AC current of at least two frequency components, a plurality of rectangular-wave phase voltage pulses are generated within one electric angular cycle of one of frequencies of the AC current, phases and widths of the plurality of rectangular-wave phase voltage pulses are operated;, and the AC current of the electric angular frequency components and an AC current of frequency components of its plurality of orders are controlled. <P>COPYRIGHT: (C)2008,JPO&INPIT

Description

  The present invention relates to a method for controlling an AC motor that includes at least one rotor and that is driven by superposing AC currents of at least two frequency components.

2. Description of the Related Art Conventionally, there is known an AC motor control method that includes at least one rotor and drives an AC current with at least two frequency components superimposed (see, for example, Patent Document 1). The composite current is controlled by superimposing wave voltage pulses in the amplitude direction.
JP 2003-299393 A

  However, in the conventional motor control method described above, in order to superimpose rectangular wave voltage pulses in the amplitude direction, it is necessary to configure the power converter with a multi-level inverter, which increases the number of inverter switches and increases the cost of the apparatus.・ There was a problem with increasing size. In addition, since the rectangular voltage is obtained by dividing the DC voltage in the amplitude direction, the time ratio in which the DC voltage value is applied between the lines is shortened, and there is a problem that the voltage utilization rate is lowered.

  The object of the present invention is to solve the above-mentioned problems and to enable control of a composite current by a rectangular wave voltage even when a normal two-level inverter is used, to improve the voltage utilization rate and to increase the output of the motor. It is intended to provide a motor control method capable of

  The motor control method of the present invention is an AC motor control method that includes at least one rotor and is driven by superimposing alternating currents of at least two frequency components. In addition, a plurality of rectangular wave phase voltage pulses are generated, the phase and width of the plurality of rectangular wave phase voltage pulses are manipulated, and the alternating current of the electrical angular frequency component and the alternating current of the frequency component of the plurality of orders are obtained. It is characterized by controlling.

  In the motor control method of the present invention, it is possible to control each of the torque based on the fundamental frequency and the torque based on the multi-order component by manipulating the width and phase of the plurality of pulses without using the multi-level inverter. Also, by using multiple pulses, the utilization rate of power supply voltage can be improved compared to multi-level inverters using composite current control by PWM or a voltage lower than the power supply voltage. The output can be increased. Furthermore, for PWM, the switching loss can be reduced and the efficiency can be improved.

  As a preferred example of the motor control method of the present invention, the number of rectangular wave phase voltage pulses to be generated is two, the phase and width of the two rectangular wave phase voltage pulses are manipulated, and the electric angular frequency component And an alternating current having a frequency component twice that of the alternating current can be controlled. With this configuration, it is possible to control each of the torque due to the fundamental frequency and the torque due to the secondary component by manipulating the width and phase of the two pulses without using a multi-level inverter. . Also, by using two-pulse drive, compared to multi-level inverters that use complex current control by PWM or a voltage lower than the power supply voltage, the power supply voltage utilization rate can be improved. The output can be increased. Furthermore, for PWM, the switching loss can be reduced and the efficiency can be improved.

  Furthermore, as a preferred example of the motor control method of the present invention, a means for controlling the width and phase of each of the two rectangular wave phase voltage pulses is provided, and the rectangular wave phase voltage pulse is calculated from the torque command value and the rotation speed. It can be configured to determine the width and phase. With this configuration, when the torque command value and the rotation speed change, the torque command value can be realized by manipulating the pulse width and the phase.

  Furthermore, as a preferred example of the motor control method of the present invention, the two rectangular wave phase voltage pulses can be configured to be pulse voltages in which the times of the high potential and the low potential are equalized. With this configuration, unnecessary offset is not generated in the phase current, and loss and torque ripple are not generated.

  As a preferred example of the motor control method of the present invention, the two rectangular wave phase voltage pulses can be configured to be pulse voltages in which the times of the high potential and the low potential are equalized. With this configuration, even when the pulse voltage is uniform, an unnecessary offset current is not generated when an offset occurs in the phase current due to an inverter circuit or imbalance.

  Furthermore, as a preferred example of the motor control method of the present invention, a phase current offset detecting means and an offset current control means for calculating a correction pulse width are provided. The offset current control means sets the detected offset current value to 0. Thus, the corrected pulse width is calculated so that the corrected pulse voltage command is output from the pulse voltage and the corrected pulse width obtained by equalizing the time of the high potential and the low potential. With this configuration, it is not necessary to generate a phase current offset.

  Furthermore, as a preferred example of the motor control method of the present invention, a PWM voltage pulse and two rectangular wave phase voltage pulses or one rectangular wave phase voltage pulse within one electrical angle period of the rotor are used. Any voltage pulse may be provided to the AC motor. With this configuration, it is possible to improve the voltage utilization rate and improve the motor output by using two-pulse driving rather than PWM driving. In addition, by switching from 2-pulse drive to 1-pulse drive so as not to exceed the DC current upper limit limiter, the voltage utilization rate is improved and the motor output is maximized under the constraints of DC voltage and current. It can be used.

  Further, as a preferred example of the motor control method of the present invention, means for selecting a PWM voltage pulse command and a rectangular wave phase voltage pulse command and outputting the voltage pulse command, and means for adding a dead time to the voltage pulse command The power converter can be configured to be driven based on a voltage pulse command with a dead time added. With this configuration, it is possible to always add a dead time and prevent a short circuit between the upper and lower arms at any rise or fall of a pulse.

  Further, as a preferred example of the motor control method of the present invention, the means for selecting the PWM voltage pulse command and the rectangular wave phase voltage pulse command and outputting the voltage pulse command is based on the rotational speed of the rotor and the torque command value. The voltage pulse command can be selected and output. With this configuration, it is possible to obtain the driving method switching condition and to improve the voltage utilization efficiency without causing a large current to flow through the motor. Further, it is possible to switch from 2-pulse driving to 1-pulse driving so as not to exceed the upper limiter of DC current.

  Furthermore, as a preferred example of the motor control method of the present invention, the motor includes current control means for generating a phase voltage command value, and the PWM voltage pulse command is generated by modulating the phase voltage command value with a PWM carrier frequency. A control device for selecting a PWM voltage pulse command and a rectangular wave phase voltage pulse command and outputting a voltage pulse command exceeds a peak of the phase voltage command value and a preset phase voltage threshold value. In this case, a rectangular wave phase voltage pulse command can be selected. With this configuration, even if the battery voltage fluctuates, the phase voltage command value peak for controlling the two frequencies is compared with the upper limit value determined by the battery voltage, and each order component Can be switched without a large change in voltage value, that is, without a large current fluctuation.

  Further, as a preferred example of the motor control method of the present invention, the AC motor is an AC motor that includes one rotor and generates an induced voltage having an electrical angular frequency of the rotor and a frequency component twice that of the rotor. Can be configured. With this configuration, since the magnet magnetic flux includes a fundamental wave component and a secondary component, the fundamental frequency frequency and the secondary component voltage are divided into two voltages with reference to the direction of each magnetic flux. The drive is generated by voltage pulses, and torque control can be performed with each frequency component.

  Hereinafter, embodiments of the motor control method of the present invention will be described with reference to the drawings.

<Regarding the Motor Targeted by the Motor Control Method of the Present Invention>
FIGS. 1A and 1B are diagrams for explaining an example of a motor that is an object of the motor control method of the present invention. In the example shown in FIGS. 1A and 1B, a motor 1 is formed by coaxially forming a cylindrical stator 2 and a cylindrical rotor 3 provided inside the stator 2 while maintaining a predetermined gap. Configured. The stator 2 is configured by combining stator teeth 2-1 made of a plurality (18 slots in this case) of electromagnetic steel sheets. Further, a coil 2-2 is wound around each stator tooth 2-1, in the axial direction. The rotor 3 is configured by superposing an outer three-pole pair magnet 3-1 and an inner six-pole pair magnet 3-2.

  In the example shown in FIG. 1 (a), a portion of the rotor 3 in which the three pole pairs 3-1 and the six pole pair magnets 3-2 are superposed in the radial direction of the rotor 3 is overlapped with the steel plate 4 (preferably Is replaced with an air gap), and the portion where the three-pole pair magnet 3-1 and the six-pole pair magnet 3-2 overlap each other in the radial direction is replaced with the magnet 5 of one magnetic pole and the other for each polarity. The rotor 3 of the motor 1 is configured by replacing the magnet 6 with the magnetic pole.

  FIG. 2 is a diagram for explaining another example of a motor that is an object of the motor control method of the present invention. In the example shown in FIG. 2, the motor 1 includes one annular stator 2, an inner rotor 3 </ b> I and an outer rotor 3 </ b> O that are coaxially and rotatably arranged inward and outward in the radial direction. A triple structure is formed, and these are housed in a housing 7.

  In both the example shown in FIGS. 1A and 1B and the example shown in FIG. 2, in the example shown in FIGS. 1A and 1B, a composite in which currents corresponding to the outer magnet and the inner magnet are combined is combined. In the example shown in FIG. 2, the rotor can be rotated by supplying a single stator with a combined current obtained by combining the currents corresponding to the magnets in the inner rotor and the outer rotor.

<First embodiment>
FIG. 3 is a circuit / block diagram for explaining the first embodiment of the motor control method of the present invention. In the example shown in FIG. 3, an inverter 52 that supplies power to a three-phase AC motor 51 is connected to a DC power supply 54 via a smoothing capacitor 53. The motor controller 100 generates a gate drive command for each switch of the inverter 52, and receives the torque command value T * as an input, and detects the detected phase currents iu and iw and the rotor obtained from the position sensor (PS) 61. Control calculation is performed based on the phase. Here, the three-phase AC motor 51 is a permanent magnet synchronous motor as shown in FIGS. 1A and 1B, for example, and the induced voltage waveform includes a fundamental frequency and a secondary frequency.

  The phase current is detected by current sensors 60a and 60b. Each of the three-phase alternating currents may be detected, but if the condition that the sum of the three-phase alternating currents is 0 is used, the remaining one phase can be estimated by detecting two phases. Further, using the phase obtained from the position sensor 61, the phase / speed calculator 106 calculates the electrical angle θ and the mechanical rotation angular speed ωm.

  The motor torque controller 102 receives the torque command value T *, the rotational angular velocity ωm, and the DC voltage Vdc of the inverter, and generates current command values id1 *, iq1 *, id2 *, iq2 *. Here, the current command value is a current command in the dq coordinate system that rotates in synchronization with the electrical angular frequency, and id1 * and iq1 * correspond to the fundamental wave current of the motor, and the second harmonic current. These correspond to id2 * and iq2 *. For this current command value, a combination that maximizes the efficiency of the motor / inverter is selected from among current command value combinations that can realize the torque command value T *, and a three-dimensional map of Vdc, ωm, and T * is selected. Record as. When referring to a value that is an intermediate value of the prepared maps, an output value is calculated by performing linear interpolation. Further, the motor torque controller 102 outputs a pulse command in pulse driving at high rotation and a switching signal for selecting switching between PWM driving and pulse driving. Details of the pulse drive will be described later.

In the current conversion 101, the phase current is coordinate-converted into rotational coordinates synchronized with the electrical angles of the fundamental wave and the second harmonic, and id1, iq1, id2, and iq2 are obtained. Using these current values and current command values, the current controller 103 calculates phase voltage command values vu *, vv *, and vw * for each phase. About this fundamental wave and harmonic current control method, a phase voltage command value can be calculated by using harmonic current control such as Japanese Patent No. 3775290. These phase voltage command values are normalized using the DC voltage Vdc, and the modulation factors mu *, mv *, and mw * are calculated as follows.
mu * = 2vu * / Vdc
mv * = 2vv * / Vdc
mw * = 2vw * / Vdc

  These modulation factors are input to the drive pulse generator 105 to generate PWM pulses. The details of the U phase in the drive pulse generator 105 are shown in FIG. In the example shown in FIG. 4, the modulation rate command mu * is compared with a triangular wave having a set carrier frequency by the PWM pulse generator 151 to generate a drive command UP1 for the inverter P-side switch and an N-side command UN1. . This drive command is a pulse waveform with a carrier frequency.

On the other hand, the rectangular wave pulse generator 152 receives a pulse command signal and generates two or one pulse waveform as drive commands UP2 and UN2 within the electrical angular frequency as shown in FIG. In the pulse command, the phases P1 and P2 that are the center positions of the respective pulses and the widths a and b of the pulses are input, and the phases x1, x2, x3, and x4 for comparison with the electrical angle θ are calculated. .
x1 = P1-a / 2
x2 = P1 + a / 2
x3 = P2-b / 2
x4 = P2 + b / 2

  As shown in FIG. 6, these calculation results are compared with θ to generate UP2. The pulse is set to H level at x1 and x3, and the pulse is set to L level at x2 and x4. The sum of the pulse widths a and b is operated to be π. When x1 = x2, only the pulses generated at x3 and x4 are present, so one pulse appears in one cycle of electrical angle. become. UN2 is generated as a signal inverted from UP2. Similarly, V-phase and W-phase drive commands are generated.

  The two pulses shown in FIG. 5 include a fundamental frequency component of the electrical angle θ and its secondary frequency component. If the operation is performed so that the widths of a and b are equal, the secondary frequency component of the electrical angle θ is the largest, and in addition, the even-order frequency component is included. When either one of the pulse widths is set to 0, the fundamental frequency component of the electrical angle θ can be maximized, and in addition, an odd-order frequency component is included. Since the sum of the widths of a and b is π, the ratio of the widths of the fundamental frequency component and the secondary frequency component can be manipulated by these ratios. Further, by manipulating the phases P1 and P2 of these two pulses, the voltage phase of the fundamental frequency / secondary frequency component can be controlled.

  FIG. 7 shows changes in the phase δ1 of the fundamental frequency component when the pulse width b and the P2 phase are manipulated. By changing the width of b, δ1 changes non-linearly with respect to P2. However, in the phase range of FIG. Further, the phase of the secondary frequency component has a relation of ± π / 2 of the phase of P1, and can be controlled by operating P1. In the above relationship, the width is b> a, and thus, the voltage amplitude of the fundamental frequency component is controlled by manipulating the phase of the pulse having a larger pulse width of the two pulses. On the other hand, if the phase of a pulse having a small pulse width is manipulated, the voltage phase of the secondary frequency can be controlled, and the voltage amplitude of the secondary frequency can be controlled by controlling the width.

  In a permanent magnet synchronous motor, it is known that the torque can be controlled by taking the d axis in the direction of the magnetic flux and operating the angle δ (see FIG. 8) formed by the voltage vector with respect to the q axis. If δ is increased in the positive direction in the range of 0 to π / 2, the torque can be increased, and conversely, by operating δ in the negative direction, the negative torque can be increased. . As shown in FIGS. 1A and 1B, the motor of this embodiment is a motor that includes a fundamental wave component and a secondary component in the magnetic flux of the magnet. Therefore, the rotational coordinates that are synchronized based on the direction of each magnetic flux. If a system is provided, the torque can be controlled by the frequency component of the fundamental frequency and the voltage of the secondary component. The motor torque controller 102 generates the width and phase as these pulse command signals based on the torque command value.

  Therefore, according to the present invention, it is possible to control each of the torque due to the fundamental frequency and the torque due to the secondary component by manipulating the width and phase of two pulses without using a multi-level inverter. The two-pulse drive can improve the power supply voltage utilization rate compared to the composite current control by PWM and the multi-level inverter that uses a voltage lower than the power supply voltage. Can be increased. Also, for PWM, the switching loss can be reduced and the efficiency can be improved. FIG. 9 shows a phase current waveform when a two-pulse voltage is applied to the motor according to the present invention. As can be seen from FIG. 9, the current can be controlled to be a composite current of the fundamental wave current and the secondary current.

  Further, the motor torque controller 102 outputs a switching signal for 2-pulse driving, 1-pulse driving, and PWM driving. The motor torque torque controller 102 has map data as shown in FIG. 10 in advance, and switches between the modes with the torque command value and the motor rotation speed as inputs. By performing switching in this way, the voltage utilization factor can be improved and the output of the motor can be improved by using two-pulse driving rather than PWM driving. In addition, during two-pulse driving, DC current pulsation due to the second harmonic of the phase current also occurs, so that switching from two-pulse driving to one-pulse driving is performed so as not to exceed the upper limiter of DC current. Since one-pulse driving is performed by setting b = 0 among the pulse width commands a and b, it is not necessary to have pulse generation means for one-pulse driving. By driving in this way, it becomes possible to improve the voltage utilization factor under the limitation of the DC voltage / current and to use the output of the motor to the maximum.

  Switching of each driving method is performed by gate selection in the gate signal selection 153 as in the example shown in FIG. 4, and after the selection output, the dead time generation 154 sets the rise time of each pulse. By delaying, the short circuit of the upper and lower arms is prevented. When a data time generation circuit is provided for each of PWM driving and rectangular wave driving, it is impossible to prevent a vertical short circuit at the time of switching, but dead time generation is performed after each driving method is selected as in the configuration of the present invention. Therefore, it is possible to always add a dead time and prevent a short circuit between the upper and lower arms at the rise and fall of any pulse.

<Second embodiment>
FIG. 11 is a circuit / block diagram for explaining a second embodiment of the motor control method of the present invention. In the second embodiment, only differences from the first embodiment will be described. In the example shown in FIG. 11, the stator 51 between the outer rotor and the inner rotor is a motor 51b (see the example of FIG. 2) on which six-phase windings are applied. This six-phase motor 51b is driven by an inverter 52b. The motor controller 100b detects the phase θ1 of the outer rotor and the phase θ2 of the inner rotor with the position sensors 61a and 61b. In addition, six phase currents are detected, coordinate conversion is performed, and current control in dq coordinates is performed during PWM driving. In the present embodiment, the six-phase motor is configured in two sets of three-phase windings, and the phase current is estimated by estimating one of the three phases in the same manner as the first embodiment with three phases. Four sensors 60a, 60b, 60c, and 60d may be used.

  When the electrical angular velocity of the inner rotor and the electrical angular velocity of the outer rotor are 2: 1, two pulses of voltage are generated within one electrical angle cycle synchronized with the rotation of the outer rotor, as in the first embodiment. At this time, the pulse having the wider pulse width is used for torque control of the outer rotor, and its phase and width are manipulated. In addition, the pulse with the narrower pulse width is used for torque control of the inner rotor. These pulse widths set the ratio of the voltage supplied to both rotors to the ratio of the pulse width. By performing such two-pulse driving, it is possible to drive the motor with improved voltage utilization and efficiency without making the six-phase inverter multi-level.

<Third embodiment>
FIG. 12 is a circuit / block diagram for explaining a third embodiment of the motor control method of the present invention. In the third embodiment, only differences from the first embodiment will be described. In the example shown in FIG. 12, an LPF (low-pass filter) 107 extracts a DC component included in the phase current detected for each phase. In the correction pulse calculation 108, the DC component current command value is set to 0 based on the extracted DC components iu0 and iw0, PI control is performed, and the U-phase correction pulse vu0 * and the V-phase correction pulse vv0 * as the DC component control pulse are performed. Is calculated.

  The correction pulses vu0 * and vv0 * are values indicating the pulse width. The correction pulse is added as a = a + vu0 * to the wider one of the two pulse width commands a and b output from the motor torque controller. However, as shown in FIG. 13, the limiter 111 is provided so that the corrected pulse width a is not 0 or less. Two pulses are generated using the pulse widths a and b generated in this manner. An example is shown in FIG.

<Fourth embodiment>
FIG. 15 is a circuit / block diagram for explaining a fourth embodiment of the motor control method of the present invention. In the fourth embodiment, only differences from the first embodiment will be described. In the example shown in FIG. 15, in the current controller 103d of the motor controller 100d, the current command values id1 *, iq1 *, id2 *, iq2 * on the dq coordinates of each order component of the motor 51, and the detected current value With id1, iq1, id2, and iq2 as inputs, phase voltage command values vu *, vv *, and vw * for each phase are calculated.

  Details of the current controller 103d are shown in FIG. The difference between each current command value and the detected current value is obtained and input to the PI controllers 201 to 204 to calculate the control voltage. In addition, in the calculation units 205 and 206, the speed electromotive force term is calculated in a feed-forward manner from the current command value and the electrical angular velocity using parameters of the motor inductance value and the induced voltage constant. The outputs vd1f, vq1f, vd2f, and vq2f are added to the outputs of the respective PI controllers, converted into three-phase alternating currents by coordinate converters 207 and 208 for three-phase alternating current, and phase voltages for each phase. The command value is set to k, and vu *, vv *, and vw * are calculated.

  Further, the calculated vd1f, vq1f, vd2f, and vq2f are input to the peak calculator 209 of the phase voltage peak value Vpk. After calculating the voltage phase and amplitude of the fundamental wave from vd1f and vq1f, and calculating the voltage phase and amplitude of the second harmonic from vd2f and vq2f, the peak value Vpk of the phase voltage is calculated from the phase difference and amplitude. Calculate. This Vpk is input to the switching signal generator 210 and compared with the value of half the battery voltage Vdc, Vdc / 2. When Vpk exceeds, switching to two-pulse driving is performed, and when not exceeding, PWM driving is performed. Switch.

  By switching in this way, even when the battery voltage fluctuates, the phase voltage command value peak for controlling the two frequencies is compared with the upper limit value determined by the battery voltage, and each order Switching can be performed without a large change in the voltage value of the component, that is, without a large current fluctuation. Further, the phase voltage command value obtained as a result of feedback control such as PI control is not used for switching determination, but is calculated in a feed-forward manner from the current command value and the rotation speed, so that the steady-state phase voltage command value is obtained. A value close to can be calculated. Thus, even when the peak value of the phase voltage command value becomes transiently large when the torque command value changes transiently, switching between PWM driving and two-pulse driving does not occur unnecessarily.

  According to the motor control method of the present invention, it is possible to control each of the torque based on the fundamental frequency and the torque based on the multi-order component by manipulating the width and phase of the plurality of pulses without using the multi-level inverter. . Therefore, even if a normal two-level inverter is used, it is possible to provide a motor control method that can control a composite current by a rectangular wave voltage, improve a voltage utilization factor, and increase a motor output.

(A), (b) is a figure for demonstrating an example of the motor used as the object of the motor control method of this invention, respectively. It is a figure for demonstrating the other example of the motor used as the object of the motor control method of this invention. It is a circuit and block diagram for demonstrating 1st Example of the motor control method of this invention. FIG. 4 is a block diagram showing details of a U phase in the drive pulse generator shown in FIG. 3 in the first embodiment. It is a figure which shows the state of 2 pulses as an example of the pulse used by 1st Example. In a 1st Example, it is a figure for demonstrating the relationship between calculation result x1, x2, x3, x4 and electrical angle (theta). In the first embodiment, it is a graph for explaining the change of the phase δ1 of the fundamental frequency component when the pulse width b and the P2 phase are manipulated. 5 is a graph for explaining a relationship between a voltage vector Vdq and an angle δ formed by the q axis in the dq coordinate system in the first embodiment. In a 1st Example, it is a graph for demonstrating the phase current waveform at the time of applying 2 pulse voltage to a motor. In a 1st Example, it is a graph for demonstrating the map data of the relationship between a torque command value and a rotor rotational speed. It is a circuit and block diagram for demonstrating 2nd Example of the motor control method of this invention. It is a circuit and block diagram for demonstrating the 3rd Example of the motor control method of this invention. In 3rd Example, it is a figure for demonstrating an example of correction | amendment of the pulse width which used the limiter. It is a figure which shows the state of 2 pulses as an example of the pulse used by 3rd Example. It is a circuit block diagram for demonstrating the 4th Example of the motor control method of this invention. In 3rd Example, it is a block diagram for demonstrating the detail of the current controller shown in FIG.

Explanation of symbols

DESCRIPTION OF SYMBOLS 1 Motor 2 Stator 2-1 Starter teeth 2-2 Coil 3 Rotor 3-1, 3-2, 5, 6 Magnet 3I Inner rotor 3O Outer rotor 4 Steel plate 7 Housing 51 Three-phase AC motor 52 Inverter 53 Smoothing capacitor 54 DC power supply 60a, 60b, 60c, 60d Current sensor 61, 61a, 61b Position sensor 100, 100b, 100c, 100d Motor controller 101, 101b, 101c, 101d Current converter 102, 102b, 102c, 102d Motor torque controller 103, 103b, 103c, 103d Current controller 104, 104b, 104c, 104d Modulation rate calculator 105, 105b, 105c, 105d Drive pulse generator 106, 106b, 106c, 106d Phase / velocity calculator 107 LPF
108 Correction Pulse Calculation 151 PWM Pulse Generator 152 Square Wave Pulse Generator 153 Dead Time Generation 201-204 PI Controller 205, 206 Calculation Unit 207, 208 Coordinate Converter 209 Peak Calculator 210 Switching Signal Generator

Claims (11)

  In a control method of an AC motor including at least one rotor and driving an alternating current of at least two frequency components superimposed, a plurality of rectangular wave phase voltages within one period of an electrical angle of any frequency of the alternating current A motor that generates a pulse, operates a phase and a width of a plurality of rectangular wave phase voltage pulses, and controls an alternating current of the electrical angular frequency component and an alternating current of a frequency component of a plurality of orders. Control method.   The number of rectangular wave phase voltage pulses to be generated is two, the phase and width of the two rectangular wave phase voltage pulses are manipulated, and the alternating current of the electrical angular frequency component and the alternating current of the double frequency component The motor control method according to claim 1, wherein:   A means for controlling the width and phase of each of the two rectangular wave phase voltage pulses is provided, and the width and phase of the rectangular wave phase voltage pulse are obtained from the torque command value and the rotation speed. The motor control method described in 1.   The motor control method according to claim 3, wherein the two rectangular wave phase voltage pulses are pulse voltages obtained by equalizing time of a high potential and a low potential.   The motor control method according to claim 3, wherein the two rectangular wave phase voltage pulses are pulse voltages in which the times of the high potential and the low potential are made uneven.   Means for detecting the offset of the phase current, and offset current control means for calculating the correction pulse width. The offset current control means calculates the correction pulse width so that the detected offset current value becomes zero, and the high potential 6. The motor control method according to claim 5, wherein a corrected pulse voltage command is output from a pulse voltage and a corrected pulse width obtained by equalizing the time between the low potential and the low potential.   Supplying either a PWM voltage pulse and a voltage pulse of two rectangular wave phase voltage pulses or one rectangular wave phase voltage pulse to the AC motor within one electrical angle period of the rotor The motor control method according to claim 1, wherein the motor control method is one of the following.   A means for selecting a PWM voltage pulse command and a rectangular wave phase voltage pulse command and outputting a voltage pulse command and a means for adding a dead time to the voltage pulse command are provided. The motor control method according to claim 7, wherein driving is performed based on a voltage pulse command.   The means for selecting the PWM voltage pulse command and the rectangular wave phase voltage pulse command and outputting the voltage pulse command selects and outputs the voltage pulse command from the rotation speed of the rotor and the torque command value. The motor control method according to claim 7.   Current control means for generating a phase voltage command value is provided, and the PWM voltage pulse command is a motor control device that generates the phase voltage command value by modulating the phase voltage command value with a PWM carrier frequency, the PWM voltage pulse command and the rectangular wave phase voltage A means for selecting a pulse command and outputting a voltage pulse command selects a rectangular wave phase voltage pulse command when a peak of the phase voltage command value and a preset phase voltage threshold value are exceeded. The motor control method according to claim 7.   11. The AC motor according to claim 1, wherein the AC motor includes one rotor, and is an AC motor that generates an induced voltage having an electrical angular frequency of the rotor and a frequency component twice that of the rotor. The motor control method described in 1.

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