JP2008099421A - Dc-dc converter - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To reduce switching loss and surge noise using a simple circuitry. <P>SOLUTION: The DC-DC converter 10 has a coupling inductor 16, a switching function section 18, an auxiliary inductor 22, and a diode 23. Between two input power supply terminals Ti1 and Ti2, the primary inductor of the coupling inductor 16 and a switching element 30 are connected in series. A diode 23 and the auxiliary inductor 22 are provided in between the joint P01 of the primary inductor 16a and the switching element 30 and an output power supply terminal To1. In the switching function section 18, a snubber series circuit of a snubber diode 34 and a snubber capacitor 36 is connected in parallel with the switching element 30, and a regeneration diode 38 is provided between a junction P1 and a secondary inductor 16b. The other end of the secondary inductor 16b is connected to the output terminal To1. <P>COPYRIGHT: (C)2008,JPO&INPIT

Description

  The present invention relates to a non-insulated step-up DC-DC converter, and more particularly to a DC-DC converter capable of reducing switching loss.

  Some DC-DC converters perform a voltage conversion by PWM operation of switching elements such as power transistors, IGBTs, and FETs, and are used in a wide range of fields. DC-DC converters are required to have further low loss, high efficiency and low noise as power saving, miniaturization and high performance of electronic devices. In particular, switching loss and switching surge caused by PWM operation are desired. Reduction is desired.

  One of the technologies for reducing such switching loss and switching surge is soft switching technology. For example, an auxiliary circuit for reducing switching loss in a general buck-boost type DC-DC converter including an inductor, a switching element, and a diode. Japanese Patent Application Laid-Open No. H10-228707 proposes a product with the addition of.

  As shown in FIG. 12, the auxiliary circuit section in Patent Document 1 includes capacitors 901a and 901b connected between the collectors and emitters of the transistors 900a and 900b, and further between the connection point 902 and the output terminal and the reference terminal. In addition, the first and second auxiliary current paths are configured. The first winding 904a of the inductor 903 and the transformer 904 is common to the first and second auxiliary current paths.

  In the first auxiliary current path, a path from the one winding 904a to the output terminal via the transistor 905a is formed, and in the second auxiliary current path, a path from the one winding 904a to the reference terminal via the transistor 905b is formed. The Capacitors 902a and 902b and the first and second auxiliary current paths reduce the terminal voltage during the switching operation of transistors 900a and 900b, thereby reducing the switching loss.

JP 2005-102438 A

  However, although the above technique is effective for reducing the switching loss, the number of parts increases due to the addition of the auxiliary circuit. In particular, since the number of switching elements increases, the number of parts of the control circuit for PWM driving also increases.

  A typical switching loss in the conventional circuit will be described with reference to FIG.

  Here, soft switching is a switching method for realizing ZVS (Zero Voltage Switching) or ZCS (Zero Current Switching), and has low switching loss and stress applied to the power semiconductor device. On the other hand, a switching method in which the voltage / current is directly turned on / off by the switching function of the power semiconductor device is called hard switching. In the following description, a method in which both or one of ZVS / ZCS is realized is called soft switching, and the other is called hard switching.

  FIG. 13 shows a voltage / current waveform at the time of switching of an IGBT (Insulated Gate Bipolar Transistor) as a power semiconductor device, a solid line 920 is a voltage, and a broken line 922 is a current. The IGBT is a power device in which high-speed switching and voltage driving characteristics of a power MOS-FET and low saturation ON voltage characteristics of a bipolar transistor are configured on a single chip. However, this transistor structure is turned on later than the MOS-FET structure during the turn-on operation. Further, the turn-off of the MOS-FET structure blocks a path through which holes that are accumulated minority carriers flow out, so that the turn-off is delayed and a tail current 924 is generated. As can be seen from these characteristics, in the switching characteristics of the IGBT power device, there is a turn-on time and a turn-off time specific to the switch. Therefore, there is a slight voltage / current crossover (see hatching) in the switching time, resulting in switching loss. Is occurring.

  This switching loss is generated as heat at the time of switching, hinders high frequency operation, and the cooling device including the radiating fins becomes larger and becomes a problem that cannot be ignored with higher frequency. In addition, parasitic parameters of floating inductors, capacitor passive circuit elements, and power semiconductor devices exist in the path connecting the power supply, power semiconductor device, and load. A surge voltage 926 and a surge current 928 as shown in FIG. 13 are generated depending on the components, and a voltage / current peak stress is generated in the power semiconductor device.

  In addition, simple voltage / current turn-on / turn-off, so-called hard switching, is often insufficient for high-efficiency control of large power with a large output capacity. In particular, when the surge current di / dt is high, the EMI noise level is high, and the voltage between the noise terminals is generated over a wide frequency band. The size increases. In addition, due to an increase in dv / dt and di / dt stress due to switching and an increase in switching loss, it is expected that the power operating device (SOA) specific to the power semiconductor device will be exceeded depending on the load state. The reliability is not necessarily high. In addition, there is a concern that a ground leakage current due to dv / dt, an increase in voltage between noise terminals due to this, and a dielectric breakdown of a low-pass filter reactor, transformer, and AC motor winding due to di / dt may occur. For this reason, it is necessary to provide a snubber circuit so that the voltage / current surge does not exceed the SOA during high-frequency switching. However, the snubber circuit reduces dv / dt and di / dt stress due to switching loss and surge, but a new problem arises such as loss due to the snubber circuit itself. Thus, the effects of switching loss and voltage / current stress, and the increase in cost and loss caused by the design of the snubber circuit and noise filter provided as countermeasures may negate the advantages of high-frequency switching. From such a background, power converters using hard switching to soft switching technology are being developed.

  The present invention has been made in consideration of the above problems, and in a non-insulated DC-DC converter, a DC- that can realize soft switching with reduced switching loss and surge noise with a simple circuit configuration. An object is to provide a DC converter.

  The DC-DC converter according to the present invention includes a coupled inductor composed of a primary inductor and a secondary inductor, a switching element connected in series with the primary inductor between two input power supply terminals, the primary inductor, and the switching element. And a main diode provided between the connection point and one end of the output power supply terminal, and a snubber series circuit comprising a snubber diode and a snubber capacitor connected in parallel to the switching element A regenerative diode is provided between a connection point between the snubber diode and the snubber capacitor and one end of the secondary inductor, and the other end of the secondary inductor is connected to one end of the output power supply terminal. To do.

  According to such a configuration, surge noise between switching elements can be reduced by the snubber capacitors connected in parallel when the switching elements are turned off. Further, when the switching element is turned on, the snubber capacitor starts to discharge due to a resonance phenomenon between the snubber capacitor and the leakage inductance component of the coupled inductor, and the energy stored in the snubber capacitor can be supplied to the output side. .

  In this case, an auxiliary inductor may be provided between a connection point between the primary inductor and the switching element and one end of the output power supply terminal. Thereby, the current rising at the time of turn-on between the switching elements is suppressed, and the current surge can be reduced.

  A resonant inductor may be provided between a connection point between the snubber diode and the snubber capacitor and one end of the secondary inductor.

  By providing such a resonant inductor, a more reliable resonance phenomenon is generated by the snubber capacitor and the resonant inductor when the switching element is turned on, and the energy of the snubber capacitor can be more reliably regenerated to the output side.

  Further, the rise of current when the switching element is turned on is further suppressed, and after the capacitor is completely discharged, the residual energy accumulated in the resonant inductor is discharged to the output side, thereby further improving the efficiency. Inserting these resonant inductors is particularly effective when the leakage inductance component of the secondary inductor is small.

  According to the DC-DC converter of the present invention, surge noise between switching elements can be reduced by a snubber capacitor connected in parallel when the switching element is turned off. In addition, due to a resonance phenomenon between the snubber capacitor and the leakage inductance component of the coupled inductor when the switching element is turned on, the snubber capacitor starts discharging, and the energy stored in the snubber capacitor can be supplied to the output side. . Furthermore, such switching loss and surge noise can be reduced with a simple circuit configuration.

  Embodiments of the DC-DC converter according to the present invention will be described below with reference to FIGS.

  As shown in FIG. 1, the DC-DC converter 10 according to the present embodiment is a non-insulated boost type, and boosts the voltage of a DC source power supply 11 and supplies it to a load R.

  The DC-DC converter 10 has positive and negative connection Ti1 (one end of the input power supply terminal) and Ti2 (other end of the input power supply terminal) on the input side, and positive and negative connection To1 (output) on the output side. One end of the power supply terminal) and To2 (the other end of the output power supply terminal).

  The DC-DC converter 10 includes an input capacitor 12 and an output capacitor 14 that stabilize the voltage on the input side and the output side, a two-winding type coupled inductor 16, a switching function unit 18, an auxiliary inductor 22, a diode ( Main diode) 23. As the input capacitor 12 and the output capacitor 14, for example, electrolytic capacitors are used.

  The switching function unit 18 collectively represents a plurality of elements for convenience of explanation, and mainly has a boosting action.

  The DC-DC converter 10 includes a resonant inductor 24 that is a leakage inductor that is not generated as a specific element on the circuit, but is generated due to characteristics of the circuit.

  The coupled inductor 16 includes a primary inductor 16a and a secondary-side secondary inductor 16b. One end of the primary inductor 16a is connected to the positive input terminal Ti1, and the other end is connected to the anode of the diode 23 via the connection point P02.

One end of the secondary inductor 16b is connected to a connection point P01 on the same line as the plus-side output terminal To1, and the other end is connected to the switching function unit 18. It is assumed that a resonant inductor 24 exists between the secondary inductor 16b and the switching function unit 18. The turn ratio between the primary inductor 16a and the secondary inductor 16b is R ₁ = n2 / n1.

  The switching function unit 18 includes a switching element 30, a reverse conducting diode (or parasitic diode) 32 provided in parallel with the switching element 30, a snubber diode 34 and a snubber capacitor 36 connected in series, and a regenerative diode 38. Have The snubber capacitor 36 and the snubber diode 34 form a snubber series circuit.

  The switching element 30 is a semiconductor element and includes, for example, a switching element such as a power transistor, IGBT, or FET, and a base terminal is driven by a controller (not shown) to perform a PWM operation.

  The switching element 30 has a collector connected to the connection point P02 and the cathode of the reverse conducting diode 32, and an emitter connected to the ground line G and the anode of the reverse conducting diode 32.

  The anode of the snubber diode 34 is connected to the collector of the switching element 30, and the cathode is connected to one end of the snubber capacitor 36. The other end of the snubber capacitor 36 is connected to the emitter of the switching element 30 (that is, the ground line G).

  The anode of the regenerative diode 38 is connected between the snubber diode 34 and the snubber capacitor 36. Let this connection location be a connection point P1. The cathode of the regenerative diode 38 is connected to the secondary inductor 16 b via the resonant inductor 24.

  The anode of the diode 23 is connected to the connection point P02, and the cathode is connected to one end of the auxiliary inductor 22. The other end of the auxiliary inductor 22 is connected to the connection point P01.

  The name and direction of the current and voltage at each location in the DC-DC converter 10 are defined as follows.

The current in the direction from the primary inductor 16a toward the connection point P02 is i ₁ , and the current in the direction from the connection point P02 to the auxiliary inductor 22 is i ₂ .

The current flowing in the direction from the connection point P02 to the switching element 30 is i _s , the current flowing from the snubber capacitor 36 toward the ground line G is i _cs , and the current flowing from the regenerative diode 38 toward the resonant inductor 24 is i _ls . To do.

The forward current of the reverse conducting diode 32 is i _ds . Let the forward current of the snubber diode 34 be i _da .

  Further, the voltage of the source power supply 11 is Vi, and the voltage supplied to the load R is Vo. Further, a voltage generated at both ends of the switching element 30 (collector voltage with respect to the emitter) is Vs, and a voltage generated at both ends of the snubber capacitor 36 (voltage at the connection point P1 with respect to the ground line G) is Vcs.

  Next, the step-up action using the DC-DC converter 10 configured as described above will be described.

  The step-up action of the DC-DC converter 10 can be divided into six modes, mode 0 to mode 5, as shown in FIG. Mode 0 to mode 5 are repeated in this order. During the step-up action, the switching element 30 performs an on / off operation based on the PWM operation to perform a DC step-up chopper. 3 to 9, the current flow is represented by an arrow I. Further, a portion where no current flows or a portion which is not particularly important in the explanation of each mode is indicated by a broken line.

  As shown in FIG. 3, in mode 0, the energy stored in the source power supply 11 and the coupled inductor 16 is supplied to the output side, and a boosting operation is performed. That is, the power supplied from the source power supply 11 is smoothed by the input capacitor 12, passes through the primary inductor 16 a, the diode 23, and the auxiliary inductor 22, smoothed by the output capacitor 14, and supplied to the load R. In mode 0, it is assumed that the snubber capacitor 36 is charged.

  As shown in FIG. 4, in mode 1, the switching element 30 is first turned on. As a result, a current mainly indicated by an arrow I is generated. That is, the signal is supplied to the connection point P01 through the snubber capacitor 36, the regenerative diode 38, the resonant inductor 24, and the secondary inductor 16b. In this system, resonance is generated by the snubber capacitor 36 and the resonant inductor 24 to form a passive resonant snubber, and the snubber capacitor 36 starts discharging. That is, resonance is generated using the energy stored in the resonant inductor 24, and the electric charge of the snubber capacitor 36 is discharged, thereby obtaining a pulse current regeneration action.

  As indicated by an arrow I, a current flows through the primary inductor 16a, the connection point P02, and the switching element 30.

At this time, a certain amount of energy is stored in the auxiliary inductor 22, and the current i ₂ continues to flow to the connection point P01.

  In mode 1, when the switching element 30 is turned on, the auxiliary inductor 22 suppresses the rising of the current flowing through the switching element 30 and the switching element 30 is turned on (see FIG. 2). That is, since the current I2 flowing through the auxiliary inductor 22 at the time of turn-on decreases linearly, the current flowing through the switching element 30 increases linearly and ZCS turn-on is realized.

The pulse current regeneration is generated as follows by the partial resonance between the resonant inductor 24 and the snubber capacitor 36. First, the voltage R ₁ Vi is generated in the secondary inductor 16b to generate the snubber capacitor voltage Vcs. The circuit state equation in the mode 1 energy regeneration snubber circuit is as shown in equation (1).

  Here, Ls is the inductance of the resonant inductor 24, and Cs is the capacitance of the snubber capacitor 36.

Further, when the switching element 30 is turned on and the initial values of the snubber capacitor voltage and the regenerative current are Vcs = Vco and i _ls = 0, respectively, the snubber capacitor voltage Vcs and the regenerative current i _ls are (2). It becomes like the formula.

  However, ω = 1 / √ (Ls · Cs) is an angular frequency. In order for the energy of the snubber capacitor to be completely discharged, a condition Vcs ≦ 0 is required to make the voltage of the snubber capacitor zero or less at the time of ωt = π.

Here, Vα (= Vco−Vo) is a jump voltage of the snubber capacitor voltage Vcs and depends on the load current. Vα is at least 0, and the turn ratio is determined under this condition. Therefore, if the turn ratio is ½ of the step-up ratio of the coupled inductor 16, the energy of the snubber capacitor 36 is completely discharged. The current that flows through the switching element 30 at the time of turn-on is a combination of the commutation current i ₂ from the diode 23 and the current that the snubber regenerative current i _ls flows through the transformation action of the coupled inductor 16. Therefore, the maximum value of the current increase rate di / dt when the switching element 30 is turned on is expressed by the following equation (4).

Here, L ₂ is the inductance of the auxiliary inductor 22.

Next, as shown in FIG. 5, in the second half of mode 1, the energy stored in the auxiliary inductor 22 is completely released, no current flows through the auxiliary inductor 22 and the diode 23, and the current i ₂ is zero. Become.

  Thereafter, after the snubber capacitor 36 energy is completely discharged, the mode 2 is entered.

As shown in FIG. 6, in mode 2, the snubber diode 34 is turned on after the electric charge stored in the snubber capacitor 36 is completely discharged, and the residual energy stored in the resonant inductor 24 is continuously emitted as the pulse regenerative current i _ls. . That is, as indicated by the arrow I, the current flows through the snubber diode 34, the regenerative diode 38, the resonant inductor 24, and the secondary inductor 16b to reach the connection point P01. Thus, even after the snubber capacitor 36 has been discharged, the regenerative operation can be continued using the energy of the resonant inductor 24.

Thereafter, when the pulse regenerative current i _ls becomes zero, the mode 3 is entered.

  As shown in FIG. 7, in mode 3, as indicated by arrow I, the resonant inductor 24 finishes releasing energy, and the power supplied from the source power supply 11 is smoothed by the input capacitor 12, and the primary inductor 16a is switched. It flows into the ground line G through the element 30. At this time, energy is accumulated in the primary inductor 16a.

  As shown in FIG. 8, in mode 4, the switching element 30 is turned off. As a result, the current supplied from the source power supply 11 flows to the primary inductor 16a, the snubber diode 34, and the snubber capacitor 36, and the snubber capacitor 36 is charged. At this time, since the voltage Vcs across the switching element 30 is 0 (see FIG. 2), the switching element 30 is turned off at ZVS. The timing for turning off the switching element 30 is set by the duty factor of the PWM.

  When the snubber capacitor 36 is charged, Vs increases at a voltage increase rate shown in the equation (5).

  This mode 4 is continued while Vs ≦ Vo. The snubber capacitor 36 charged at this time is subjected to the resonance effect of mode 1 as described above.

In mode 4, the condition that the regenerative current does not flow through the secondary inductor 16b is expressed by the following equation (6) from Vcs ≦ R ₁ (Vo−Vi) + Vo, that is, Vo + Vα ≦ R ₁ (Vo−Vi) + Vo.

  As shown in FIG. 9, in mode 5, when the snubber capacitor 36 is sufficiently charged, no current flows through the snubber capacitor 36 and the snubber diode 34, Vs> Vo, and the diode 23 becomes conductive. As shown, the current flows to the primary inductor 16a, the auxiliary inductor 22, the diode 23, and the connection point P01 as current.

The condition of this mode 5 is Vcs> R ₁ (Vo−Vi) + Vo.

  Thereafter, the mode 0 is returned to and a series of cycles is continued.

  Thus, during the DC boost operation of the DC-DC converter 10, boosting is performed by high-frequency switching of the primary inductor 16 a and the switching element 30 of the two-winding coupled inductor 16, passing through the diode 23, and power from the source power supply 11. Is supplied to the connection point P01. Resonant inductor 24 in which the energy of a snubber series circuit composed of a snubber diode 34 and a snubber capacitor 36 provided in parallel with the switching element 30 is collected on the secondary side of the regenerative diode 38 and the coupled inductor 16 when the switching element 30 is turned on. And the snubber capacitor 36 to resonate, and the snubber energy can be regenerated to the output side.

  Further, by discharging the voltage of the snubber capacitor 36 to zero by this pulse current regeneration operation, the turn-off of the switching element 30 becomes ZVS commutation. When the switching element 30 is turned on, the auxiliary inductor 22 suppresses the rise of the current flowing through the switch, and the ZCS is turned on. As described above, the switching element 30 realizes the soft switching operation.

  Next, the soft switching characteristics used in the DC-DC converter 10 will be described with reference to FIG. FIG. 10 shows an example of the ZVS / ZCS switching waveform of the IGBT. The solid line 100 is a voltage, and the broken line 102 is a current.

  As shown in FIG. 10, generally, at the time of turn-off, a transient crossing of current and voltage slightly occurs from the rising voltage time inherent to the IGBT and the tail current generation period, and switching loss occurs. However, it can be seen that the switching loss causing the transient crossing is greatly reduced as compared with the switching method that directly cuts off the DC voltage or the like as shown in FIG. This is because the LC main resonance or the LC auxiliary resonance is used to increase the voltage between the switch terminals at the time of turn-off, the lossless capacitor incorporated in parallel with the power semiconductor device is charged, and the voltage gradually increases. In addition, suppression of surge voltage is realized at the same time, thus performing zero voltage switching operation. At the time of turn-on, an on signal is sent to the gate of the IGBT while the current flows through the reverse conducting diode 32 connected in parallel with the switching element 30, so that the current starts to flow through the switch when the current naturally commutates to zero. Voltage switching and zero current switching operations are performed.

  As is clear from the comparison between FIG. 10 and FIG. 13, the transient crossing of the current and voltage does not occur except for a slight crossing with the on-voltage of the IGBT, and the switching loss can be reduced as compared with the conventional switching. Surge voltage and current are also suppressed.

  Thus, by performing switching operation using both or one of ZVS / ZCS, switching loss and stress at the time of switching transient are reduced, and EMI noise and RFI noise are suppressed.

  FIG. 11 shows a voltage / current switching locus of the power semiconductor device by a broken line 110 when the conventional hard switching method is used, and a solid line 112 when the soft switching method is used.

  In the case of the hard switching system, the switching loss due to the transient crossing of the current and voltage at the time of switching is large, and the dv / dt stress and di / dt stress both increase and operate near the SOA limit inherent in power semiconductor devices. A voltage surge or current surge has occurred. Therefore, in general, in the hard switching system, a snubber circuit is loaded so that the switching locus of the power semiconductor device is close to both the voltage and current axes.

  On the other hand, in the soft switching method, it is understood that the switching loss is greatly reduced because the switching locus passes through the vicinity of the vertical current axis and the horizontal voltage axis without snubber. From the above, when the soft switching method is applied, switching loss, surge voltage and surge current at the time of switching transient can be reduced, and EMI / RFI noise can be suppressed.

  As described above, the DC-DC converter 10 according to the present embodiment can perform efficient voltage conversion by the passive resonance snubber. Further, only the switching element 30 is sufficient as the switching element, and there are few peripheral elements for soft switching. The control method of the switching element 30 can be easily performed without changing from the conventional hard switching PWM.

  In the snubber circuit of the DC-DC converter 10, surge voltage and surge current can be suppressed and regenerative operation is performed, so that almost no loss is caused by the snubber circuit itself.

  In the DC-DC converter 10, the switching element 30 can of course perform a soft switching operation in both cases of high load and light load.

  The DC-DC converter according to the present invention is not limited to the above-described embodiment, but can of course adopt various configurations without departing from the gist of the present invention.

It is a circuit diagram of the DC-DC converter which concerns on this Embodiment. It is a time chart at the time of pressure | voltage rise. It is a circuit diagram which shows the flow of an electric current in mode 0 at the time of pressure | voltage rise. It is a circuit diagram which shows the flow of an electric current in the first half of the mode 1 at the time of pressure | voltage rise. It is a circuit diagram which shows the flow of an electric current in the second half of the mode 1 at the time of pressure | voltage rise. It is a circuit diagram which shows the flow of an electric current in mode 2 at the time of pressure | voltage rise. It is a circuit diagram which shows the flow of an electric current in the mode 3 at the time of pressure | voltage rise. It is a circuit diagram which shows the flow of an electric current in mode 4 at the time of pressure | voltage rise. It is a circuit diagram which shows the flow of an electric current in mode 5 at the time of pressure | voltage rise. It is a ZVS / ZCS switching waveform example of IGBT. It is a voltage / current switching locus of a power semiconductor device. It is a circuit diagram of the DC-DC converter which concerns on a prior art. It is an example of the switching waveform which concerns on a prior art.

Explanation of symbols

DESCRIPTION OF SYMBOLS 10 ... DC-DC converter 16 ... Coupling inductor 16a ... Primary inductor 16b ... Secondary inductor 18 ... Switching function part 22 ... Auxiliary inductor 23 ... Diode 24 ... Resonance inductor 30 ... Switching element 32 ... Reverse conducting diode 34 ... Snubber diode 36 ... Snubber capacitor 38 ... Regenerative diode Ti1 ... One end of input power supply terminal Ti2 ... Other end of input power supply terminal To1 ... One end of output power supply terminal To2 ...

Claims (3)

A coupled inductor consisting of a primary inductor and a secondary inductor;
A switching element connected in series with the primary inductor between two input power supply terminals;
A main diode provided between a connection point of the primary inductor and the switching element to one end of an output power supply terminal;
A DC-DC converter comprising:
A snubber series circuit composed of a snubber diode and a snubber capacitor is connected in parallel to the switching element,
A regenerative diode is provided between a connection point between the snubber diode and the snubber capacitor and one end of the secondary inductor,
The DC-DC converter characterized in that the other end of the secondary inductor is connected to one end of the output power supply terminal. The DC-DC converter according to claim 1, wherein
A DC-DC converter comprising an auxiliary inductor between a connection point between the primary inductor and the switching element and one end of the output power supply terminal. The DC-DC converter according to claim 1 or 2,
A DC-DC converter comprising a resonant inductor between a connection point between the snubber diode and the snubber capacitor and one end of the secondary inductor.

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