JP2008016884A - Dielectric waveguide device, phase shifter provided with the same, high frequency switch and attenuator, and high frequency transmitter, high frequency receiver, high frequency transmitter-receiver, radar apparatus, and array antenna system - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide a compact dielectric waveguide device capable of attaining excellent phase control and adaptable to surface mount on a plane circuit board, and to provide a high frequency transmitter, a high frequency receiver, a high frequency transmitter-receiver, and a radar apparatus provided with the dielectric waveguide device. <P>SOLUTION: A plane line 3 is connected to both sides of a nonradioactive dielectric line 2 in its extended direction X, a mount dielectric part 41 is provided to the plane lines 3 and the nonradioactive dielectric line 2, and a second strip conductor part 42 is provided on the mount dielectric part 41 while being superposed on the first strip conductor part 11 of the plane line 3. A through-hole 49 is formed to the first ground conductor part 13a of the plane line 3 between the first and second strip conductor parts 13, 42. The second strip conductor part 42 can be mounted oppositely to the plane circuit board to form the dielectric waveguide device adaptable to the surface mount. An electric field is applied to first and second electrodes 4a to change the dielectric constant of a dielectric line 6 of the nonradioactive dielectric line 2 thereby controlling the phase. <P>COPYRIGHT: (C)2008,JPO&INPIT

Description

  The present invention relates to a dielectric waveguide device used in a high frequency circuit such as a microwave, a quasi-millimeter wave band, and a millimeter wave band, a phase shifter including the dielectric waveguide device, a high frequency switch and an attenuator, a high frequency transmitter, a high frequency The present invention relates to a receiver, a high frequency transmitter / receiver, a radar device, and an array antenna device.

  A phase shifter is known as a dielectric waveguide device that controls the phase of a high-frequency signal. This phase shifter is used to adjust the reflection and transmission characteristics of high-frequency signals in high-frequency circuits, and can also be applied to high-function devices such as modulators, high-frequency switches, active phased array antennas, and phased array antennas. Expected basic device.

  The first prior art phase shifter includes a non-radiative dielectric line, and the end of the dielectric line of the non-radiative dielectric line is located close to the conductor strip. A rotating body including a dielectric is rotatably arranged, and the phase is changed by rotating the rotating body (see, for example, Patent Document 1).

  The phase shifter of the second prior art controls the phase by controlling the dielectric constant of a part of the dielectric of the non-radiative dielectric line (for example, see Patent Document 2).

JP 2002-314302 A JP-A-8-102604

  In the phase shifter which is one of the dielectric waveguide devices according to the first prior art, if a sufficient amount of phase change such as ± π (rad) is to be obtained, the device is likely to be large, There is a problem that it is difficult to realize. In addition, since the length of the transmission path is mechanically changed by rotating the rotating body, there is a problem that the device itself becomes large or the mechanism becomes complicated, which is not suitable for mass production. When the device becomes larger, there is a problem that the mountability is remarkably poor as compared with other electronic components whose surface mounting is progressing.

  Further, in the phase shifter which is one of the dielectric waveguide devices of the second prior art, the device can be easily formed into a child shape, but is difficult to be surface-mounted on a planar circuit board such as a printed circuit board. There is a problem that it is difficult to connect to a transmission line formed on the planar circuit board.

  Accordingly, an object of the present invention is to provide a dielectric waveguide device that is small in size, capable of good phase control, and suitable for surface mounting on a planar circuit board, a phase shifter including the dielectric waveguide device, a high-frequency switch, and an attenuator. It is another object of the present invention to provide a high frequency transmitter, a high frequency receiver, a high frequency transmitter / receiver, a radar device, and an array antenna device.

A dielectric waveguide device according to the present invention includes a change part in which at least one of a dielectric constant and a dimension changes according to an applied electric field, a dielectric line through which electromagnetic waves propagate, and a pair of sandwiching the dielectric line A non-radiative dielectric line including a flat conductor part;
Provided along the dielectric line, and in a state perpendicular to the direction in which the dielectric line and the pair of flat plate conductors overlap, with a gap narrower than the gap between the pair of flat plate conductors. A pair of electrodes for sandwiching the dielectric line and applying an electric field to the changing portion;
The first strip conductor part, the dielectric part provided with the first strip conductor part, and the gap between the first strip conductor part in a direction perpendicular to the direction in which the dielectric line and the pair of flat plate conductor parts overlap. A first grounding conductor portion provided on the dielectric portion along the first strip conductor portion, and the non-contact so that the dielectric line and the first strip conductor portion are in contact with each other. A plane line coupled to the non-radiative dielectric line by abutting the end face in the electromagnetic wave propagation direction of the radioactive dielectric line with the end face in the electromagnetic wave propagation direction of the dielectric part,
A second ground conductor portion connected to the first ground conductor portion and provided at an end of the non-radiative dielectric line in a direction perpendicular to a direction in which the dielectric line and the pair of flat plate conductor portions overlap;
A second dielectric portion formed on a surface portion of the first ground conductor portion opposite to the first strip conductor portion and a surface portion of the second ground conductor portion opposite to the non-radiative dielectric line; ,
A second strip conductor portion provided at least partially overlapping the first strip conductor portion on a surface portion of the two dielectric portions opposite to the nonradiative dielectric line and the dielectric line;
Including at least a first electrode connection portion and a second electrode connection portion provided on the surface portion of the second dielectric portion and individually connected to a pair of the electrodes;
The first ground conductor part is formed with a through hole between the first and second strip conductor parts.

  The dielectric waveguide device of the present invention is characterized in that the pair of electrodes are formed thinner than the skin thickness with respect to the electromagnetic wave propagating through the dielectric line.

  Further, the dielectric waveguide device of the present invention is provided between a pair of the plate conductor portions, sandwiching the dielectric line in a direction in which the pair of plate conductor portions overlap, and from the dielectric constant of the dielectric line Includes a third dielectric portion having a low dielectric constant.

The phase shifter of the present invention comprises the dielectric waveguide device,
The phase of the electromagnetic wave propagating through the transmission line is changed by changing at least one of a dielectric constant and a dimension of the changing portion according to an electric field applied to the changing portion.

Moreover, the high frequency switch of the present invention comprises the dielectric waveguide device,
The cutoff frequency in the transmission line becomes lower than the frequency of the electromagnetic wave propagating through the transmission line by changing at least one of the dielectric constant and dimension of the changing part according to the electric field applied to the changing part. It is characterized by being able to switch between a propagation state and a cutoff state that becomes higher.

The attenuator of the present invention comprises a dielectric waveguide device,
According to an electric field applied to the change part, at least one of a dielectric constant and a dimension of the change part is changed to attenuate an electromagnetic wave propagating through the transmission line.

The high frequency transmitter of the present invention includes a high frequency oscillator that generates a high frequency signal,
A transmission line connected to the high-frequency oscillator and transmitting a high-frequency signal from the high-frequency oscillator;
An antenna connected to the transmission line and emitting a high-frequency signal;
The phase shifter inserted into the transmission line such that a high-frequency signal passes through the dielectric line;
And a stub provided on at least one of the upstream side and the downstream side of the phase shifter in the transmission direction of the high-frequency signal.

The high frequency receiver of the present invention includes an antenna that captures a high frequency signal,
A transmission line connected to the antenna and transmitting a high-frequency signal captured by the antenna;
A high-frequency detector connected to the transmission line and detecting a high-frequency signal transmitted to the transmission line;
The phase shifter inserted into the transmission line such that a high-frequency signal passes through the dielectric line;
And a stub provided on at least one of the upstream side and the downstream side of the phase shifter in the transmission direction of the high-frequency signal.

The high frequency transceiver of the present invention includes a high frequency oscillator for generating a high frequency signal,
A first transmission line that is connected to the high-frequency oscillator and transmits a high-frequency signal; and first, second, and third terminals; the first terminal is connected to the first transmission line and applied to the first terminal A branching device that selectively outputs a high-frequency signal to be output to the second terminal or the third terminal;
A second transmission line connected to the second terminal and transmitting a high-frequency signal applied from the second terminal;
A fourth, fifth, and sixth terminal that outputs a high-frequency signal applied to the fourth terminal via the second transmission line to the fifth terminal, and a high-frequency signal applied to the fifth terminal; A duplexer that outputs to the sixth terminal;
A third transmission line connected to the fifth terminal for transmitting a high-frequency signal output from the fifth terminal and transmitting the high-frequency signal to the fifth terminal;
An antenna connected to the third transmission line for radiating and capturing high-frequency signals;
A fourth transmission line connected to the third terminal and transmitting a high-frequency signal output from the third terminal;
A fifth transmission line connected to the sixth terminal for transmitting a high-frequency signal output from the sixth terminal;
A mixer that is connected to the fourth and fifth transmission lines, mixes high-frequency signals given from the fourth and fifth transmission lines, and outputs an intermediate frequency signal;
The phase shifter is inserted into at least one of the first to fifth transmission lines so that a high-frequency signal passes through the dielectric line.

The high frequency transmitter of the present invention includes a high frequency oscillator that generates a high frequency signal,
A high-frequency transmission line connected to the high-frequency oscillator and transmitting a high-frequency signal from the high-frequency oscillator;
An antenna connected to the high-frequency transmission line and emitting a high-frequency signal;
The high-frequency signal inserted into the high-frequency transmission line is transmitted through the high-frequency transmission line by setting the propagation state, and the high-frequency signal transmitted to the high-frequency transmission line is blocked by setting the cutoff state. And a high-frequency switch.

The high frequency transceiver of the present invention includes a high frequency oscillator for generating a high frequency signal,
A first high-frequency transmission line that is connected to the high-frequency oscillator and transmits a high-frequency signal; and first, second, and third terminals; the first terminal is connected to the first high-frequency transmission line; A branching device that selectively outputs a high-frequency signal given to the second terminal or the third terminal;
A second high frequency transmission line connected to the second terminal and transmitting a high frequency signal applied from the second terminal;
A high-frequency signal having fourth, fifth, and sixth terminals, outputting a high-frequency signal supplied to the fourth terminal via the second high-frequency transmission line to the fifth terminal, and supplied to the fifth terminal A duplexer that outputs to the sixth terminal;
A third high-frequency transmission line that is connected to the fifth terminal, transmits a high-frequency signal output from the fifth terminal, and transmits a high-frequency signal to the fifth terminal;
An antenna connected to the third high-frequency transmission line for radiating and capturing high-frequency signals;
A fourth high-frequency transmission line connected to the third terminal and transmitting a high-frequency signal output from the third terminal;
A fifth high-frequency transmission line connected to the sixth terminal and transmitting a high-frequency signal output from the sixth terminal;
A mixer that is connected to the fourth and fifth high-frequency transmission lines, mixes high-frequency signals given from the fourth and fifth high-frequency transmission lines, and outputs an intermediate frequency signal;
The branching device includes two high-frequency switches, and the first high-frequency switch transmits the high-frequency signal between the first terminal and the second terminal and sets the cutoff state by setting the propagation state. The high frequency signal is cut off between the first terminal and the second terminal, and the second high frequency switch transmits the high frequency signal between the first terminal and the third terminal by being in the propagation state, and By setting the cut-off state, a high-frequency signal is blocked between the first terminal and the third terminal.

The high frequency transceiver of the present invention includes a high frequency oscillator for generating a high frequency signal,
A first high-frequency transmission line that is connected to the high-frequency oscillator and transmits a high-frequency signal; and first, second, and third terminals; the first terminal is connected to the first high-frequency transmission line; A branching device that selectively outputs a high-frequency signal given to the second terminal or the third terminal;
A second high frequency transmission line connected to the second terminal and transmitting a high frequency signal applied from the second terminal;
A high-frequency signal having fourth, fifth, and sixth terminals, outputting a high-frequency signal supplied to the fourth terminal via the second high-frequency transmission line to the fifth terminal, and supplied to the fifth terminal A duplexer that outputs to the sixth terminal;
A third high-frequency transmission line that is connected to the fifth terminal, transmits a high-frequency signal output from the fifth terminal, and transmits a high-frequency signal to the fifth terminal;
An antenna connected to the third high-frequency transmission line for radiating and capturing high-frequency signals;
A fourth high-frequency transmission line connected to the third terminal and transmitting a high-frequency signal output from the third terminal;
A fifth high-frequency transmission line connected to the sixth terminal and transmitting a high-frequency signal output from the sixth terminal;
A mixer that is connected to the fourth and fifth high-frequency transmission lines, mixes high-frequency signals given from the fourth and fifth high-frequency transmission lines, and outputs an intermediate frequency signal;
The duplexer includes two high-frequency switches, and the third high-frequency switch transmits the high-frequency signal between the fourth terminal and the fifth terminal by setting the propagation state, and the cutoff state The high frequency signal is cut off between the fourth terminal and the fifth terminal by doing, the fourth high frequency switch transmits the high frequency signal between the fifth terminal and the sixth terminal by being in the propagation state, The high-frequency signal is blocked between the fifth terminal and the sixth terminal by setting the cut-off state.

The high frequency transceiver of the present invention includes a high frequency oscillator for generating a high frequency signal,
A first high-frequency transmission line that is connected to the high-frequency oscillator and transmits a high-frequency signal; and first, second, and third terminals; the first terminal is connected to the first high-frequency transmission line; A branching device that selectively outputs a high-frequency signal given to the second terminal or the third terminal;
A second high frequency transmission line connected to the second terminal and transmitting a high frequency signal applied from the second terminal;
A high-frequency signal having fourth, fifth, and sixth terminals, outputting a high-frequency signal supplied to the fourth terminal via the second high-frequency transmission line to the fifth terminal, and supplied to the fifth terminal A duplexer that outputs to the sixth terminal;
A third high-frequency transmission line that is connected to the fifth terminal, transmits a high-frequency signal output from the fifth terminal, and transmits a high-frequency signal to the fifth terminal;
An antenna connected to the third high-frequency transmission line for radiating and capturing high-frequency signals;
A fourth high-frequency transmission line connected to the third terminal and transmitting a high-frequency signal output from the third terminal;
A fifth high-frequency transmission line connected to the sixth terminal and transmitting a high-frequency signal output from the sixth terminal;
A mixer that is connected to the fourth and fifth high-frequency transmission lines, mixes high-frequency signals given from the fourth and fifth high-frequency transmission lines, and outputs an intermediate frequency signal;
The high-frequency switch is inserted into at least one of the first to third transmission lines so that a high-frequency signal passes through the dielectric line when in the propagation state. .

  In the high-frequency transceiver according to the present invention, the duplexer is formed by a hybrid circuit or a circulator.

The radar apparatus of the present invention includes the high frequency transmitter / receiver,
And a distance detector for detecting a distance from the high-frequency transmitter / receiver to a detection target based on an intermediate frequency signal from the high-frequency transmitter / receiver.

  The array antenna apparatus of the present invention is characterized in that a plurality of antennas with phase shifters each having an antenna element and the phase shifter are arranged side by side.

The radar device of the present invention includes the array antenna device,
A high-frequency transmitter / receiver connected to the array antenna device, for supplying a high-frequency signal to the array antenna device, and for receiving a high-frequency signal captured by the array antenna device.

  According to the present invention, the dielectric line of the non-radiative dielectric line includes the changing portion whose dielectric constant changes according to the applied electric field, and an electric field can be applied to the changing portion by the electrode. The non-radiative dielectric line is an NRD guide. By applying an electric field to the changing portion, at least one of the dielectric constant and dimension of the changing portion changes, the phase of the electromagnetic wave propagating through the dielectric line changes, the cutoff frequency changes, the dielectric The electromagnetic wave propagating through the line can be attenuated. The electrode sandwiches the dielectric line in a direction perpendicular to the direction in which the dielectric line and the pair of flat plate conductors overlap, with a gap narrower than the gap between the pair of flat plate conductors, Since it is provided along the body line, it is possible to apply an electric field with a higher electric field strength than to apply a voltage to a pair of flat conductor portions and apply an electric field to the dielectric changing portion, and the dielectric constant of the changing portion And at least one of the dimensions can be changed stably. Even if the voltage applied to the electrodes to apply an electric field to the changing part is reduced, an electric field having a large electric field strength is applied to the changing part, and even if the transmission line length is short, an electric field having a large electric field strength is applied to the changing part. Therefore, it is possible to realize a dielectric waveguide device such as a phase shifter, a high-frequency switch, and an attenuator that is small and can be operated at a low voltage.

  According to the present invention, the end face in the propagation direction of the electromagnetic wave of the non-radiative dielectric line and the end face in the propagation direction of the electromagnetic wave of the plane line are abutted so that the dielectric line and the first strip conductor portion are in contact with each other. The nonradiative dielectric line and the planar line are coupled. Since the electromagnetic field mode in the vicinity of the first strip conductor portion approximates the LSE mode of the non-radiative dielectric line, the planar line can be coupled with the LSE mode of the non-radiative dielectric line. The electromagnetic field can smoothly transition from one of the non-radiative dielectric line and the planar line to the other at the connection portion between the line and the planar line. Therefore, connection loss can be reduced, that is, transmission loss can be reduced. Further, since the coupling is performed in the LSE mode, a mode suppression circuit is unnecessary, and the size can be reduced as compared with the case of using the LSM mode. The planar line is realized by, for example, a strip line, a microstrip line, a coplanar line, or the like.

  The first and second ground conductor portions, the second dielectric portion, and the second strip conductor portion form a second planar line. A second strip conductor portion is provided so as to at least partially overlap the first strip conductor portion, and a high frequency signal from the planar line is transmitted through a through hole formed in the first ground conductor portion. Propagates to a plane track. Since the second dielectric portion and the second strip conductor portion function as a surface mounting portion, the dielectric waveguide device and the planar circuit can be obtained by surface mounting this portion on a planar circuit substrate such as a printed circuit board. Connection with a transmission line formed on the substrate can be suitably performed.

  Further, each of the first and second electrode connection portions individually connected to the pair of electrodes is provided on the surface portion of the second dielectric portion, and thus formed on the planar circuit board. Connection with wiring for voltage application can be performed satisfactorily.

  According to the invention, the pair of electrodes are formed thinner than the skin thickness with respect to the electromagnetic wave propagating through the dielectric line. The electromagnetic wave propagating through the dielectric line can pass through the electrode, so that the electromagnetic wave can be propagated without being cut off, and the transmission loss caused when the electrode is brought close to the dielectric line is suppressed. Thus, an electric field having a large electric field strength can be applied to the dielectric change portion by the electrode.

  Since the dielectric waveguide device of the present invention has a characteristic that the phase change with respect to the frequency is large in the vicinity of the cutoff of the LSE mode propagating through the non-radiative dielectric line, if this characteristic is used, the dielectric line is sandwiched. The phase of the LSE mode high-frequency signal propagating through the dielectric line can be greatly changed by the pair of electrodes. Therefore, when a dielectric waveguide device is used as a phase shifter, a sufficient amount of phase change can be obtained with a low driving voltage even if the line length is short, so that it is small and good phase control is possible. A phase shifter is realized.

  According to the present invention, since the third dielectric portion functions as a support member that supports the flat conductor portion, the flat conductor portion is manufactured using a thin film forming technique, a thick film printing technique, a sheet-like ceramic technique, or the like. Accordingly, a dielectric waveguide device suitable for miniaturization can be realized in manufacturing. The third dielectric portion has a dielectric constant lower than the dielectric constant of the portion with the lowest dielectric constant among the changed portions. That is, the third dielectric portion has a dielectric constant of a portion having the lowest dielectric constant in the changing portion when an electric field is applied to the changing portion and when no electric field is applied to the changing portion. Has a lower dielectric constant. Thereby, the wavelength of the electromagnetic wave propagating in the third dielectric part can be made smaller than the wavelength when the electromagnetic wave propagates through the air, and thereby the dielectric waveguide device can be miniaturized.

  Further, according to the present invention, since the phase change with respect to the frequency is large in the vicinity of the cutoff of the LSE mode propagating through the non-radiative dielectric line, if this characteristic is used, a pair of sandwiching the dielectric line is used. The phase of the high-frequency signal in the LSE mode propagating through the dielectric line can be greatly changed by the electrode. Therefore, when a dielectric waveguide device is used as a phase shifter, a sufficient amount of phase change can be obtained with a low driving voltage even if the line length is short, so that it can be operated with a small size and a low voltage. And a phase shifter capable of good phase control is realized. Moreover, since there is no mechanical drive part, a highly reliable phase shifter with excellent durability can be realized.

  According to the present invention, the cut-off frequency in the transmission line is lower than the frequency of the electromagnetic wave propagating through the dielectric portion and the cut is higher than the frequency of the electromagnetic wave in accordance with the electric field applied to the changing portion. Since the OFF state can be switched, the propagation state and the cutoff state can be easily switched by changing the voltage applied to the electrode. When the switching mode is in the OFF state, the cut-off state is entered, so that an essentially high ON / OFF ratio can be obtained. In addition, since there is no mechanical drive part, it is possible to realize a highly reliable high-frequency switch having excellent durability. Even if the voltage applied to the electrodes to apply an electric field to the changing part is reduced, an electric field having a large electric field strength is applied to the changing part, and even if the transmission line length is short, an electric field having a large electric field strength is applied to the changing part. Therefore, a high-frequency switch that is small and can be operated at a low voltage can be realized. In addition, since there is no mechanical drive part, it is possible to realize a highly reliable high-frequency switch having excellent durability.

  According to the present invention, even if the voltage applied to the electrode to apply the electric field to the changing portion is reduced, an electric field having a large electric field strength is applied to the changing portion, and even if the transmission line length is short, it is large. Since attenuation is obtained, an attenuator that is small and can be operated at a low voltage can be realized. Moreover, since there is no mechanical drive part, a highly reliable attenuator with excellent durability can be realized.

  Further, according to the present invention, a stub is provided in the middle of the high-frequency oscillator and the antenna, so that mismatching in the connection portion of the high-frequency oscillator to the high-frequency transmission line and the connection portion of the antenna to the high-frequency transmission line can be matched. ing. As a result, reflection at the connection portion can be suppressed to be small, stable oscillation characteristics can be obtained, and insertion loss can be suppressed to be small, so that a high transmission output can be obtained. However, it is not possible to achieve uniform matching due to, for example, variations in the shape of wires and bumps for connecting a high-frequency oscillator and variations in the wiring width of high-frequency transmission lines. In the present invention, the phase shifter is inserted into the high frequency transmission line so that the electromagnetic wave of the high frequency signal transmitted through the high frequency transmission line passes through the dielectric line. The phase shift caused by the high-frequency transmission line can be individually adjusted and matched due to variations in the shape of the wires and bumps and the wiring width of the high-frequency transmission line. Since the insertion loss is kept small, a high-frequency transmitter having a high transmission output can be realized. In addition, since the phase shifter can be operated with a small voltage and a low voltage as described above, a high frequency transmitter can be formed in a small size even if a phase shifter is provided, and a voltage is applied to the phase shifter. Therefore, it is possible to suppress the complicated configuration.

  According to the present invention, the high-frequency signal captured by the antenna is transmitted to the transmission line and detected by the high-frequency detector. A stub is provided in the middle of the antenna and the high-frequency detector so that mismatching can be matched in the connection portion of the high-frequency detector to the high-frequency transmission line and the connection portion of the antenna to the high-frequency transmission line. As a result, reflection at the connection portion can be suppressed to a small level, and stable detection characteristics can be obtained. Further, since the insertion loss is suppressed to a low level, a high detection output can be obtained. However, for example, it cannot be matched uniformly due to variations in the shapes of wires and bumps for connecting a high-frequency detector, variations in the wiring width of the high-frequency transmission line, and the like. In the present invention, the phase shifter is inserted into the high frequency transmission line so that the electromagnetic wave of the high frequency signal transmitted through the high frequency transmission line passes through the dielectric line. The phase shift caused by the high-frequency transmission line can be individually adjusted due to variations in the shape of the wires and bumps and the wiring width of the high-frequency transmission line. At the same time, since the insertion loss is kept small, a high-frequency receiver having a high detection output can be realized. In addition, since the phase shifter can be operated with a small voltage and a low voltage as described above, a high frequency receiver can be formed in a small size even if a phase shifter is provided, and a voltage is applied to the phase shifter. Therefore, it is possible to suppress the complicated configuration.

  According to the invention, the high-frequency signal generated by the high-frequency oscillator is transmitted to the first transmission line and applied to the first terminal of the branching device, and is applied from the second terminal of the branching device to the second transmission line. The signal is given to the fourth terminal of the duplexer, given from the fifth terminal of the duplexer to the third transmission line, and radiated from the antenna. The high-frequency signal received by the antenna is given to the third transmission line, given to the fifth terminal of the duplexer, given from the sixth terminal of the duplexer to the fifth transmission line, and given to the mixer. . In addition, a high frequency signal generated by the high frequency oscillator is supplied as a local signal from the third terminal of the branching unit to the mixer via the fourth transmission line. The mixer mixes the high-frequency signal generated by the high-frequency oscillator and the high-frequency signal received by the antenna and outputs an intermediate frequency signal, thereby obtaining information contained in the received high-frequency signal.

  By inserting the phase shifter into at least one of the first to fifth transmission lines so that a high-frequency signal passes through the dielectric line, transmission is performed due to, for example, variations in wiring width. By adjusting the phase of the high-frequency signal that changes undesirably due to the line, for example, it is possible to realize a high-frequency transceiver with stable oscillation characteristics and high transmission output because the insertion loss is kept small In addition, for example, it has a stable oscillation characteristic, and the insertion loss can be suppressed to be small, so that a high frequency transmitter / receiver having a high detection output can be realized, and the reliability of the received high frequency signal can be improved. For example, the reliability of the intermediate frequency signal generated by the mixer can be improved. In addition, since the phase shifter can be operated with a small voltage and a low voltage as described above, a high frequency transmitter / receiver can be formed in a small size even if a phase shifter is provided, and a voltage is applied to the phase shifter. Therefore, it is possible to suppress the complicated configuration.

  Further, according to the present invention, when the high-frequency switch is in a propagation state, the high-frequency signal generated by the high-frequency oscillator is transmitted through the high-frequency switch, so that it is transmitted through the high-frequency transmission line and applied to the antenna and radiated as a radio wave. Further, when the high frequency switch is in the cut-off state, the high frequency signal generated by the high frequency oscillator does not pass through the high frequency switch, and therefore is cut off and is not radiated from the antenna. By switching the propagation state and cut-off state of the high-frequency switch, a pulse signal wave can be radiated from the antenna. A high-reliability high-frequency transmitter can be realized by using a high-reliability high-frequency switch with excellent durability and high ON / OFF ratio.

  According to the invention, the high-frequency signal generated by the high-frequency oscillator is transmitted to the first high-frequency transmission line and given to the first terminal of the branching device, and given from the second terminal of the branching device to the second high-frequency transmission line. , Given to the fourth terminal of the duplexer, given to the third high-frequency transmission line from the fifth terminal of the duplexer, and radiated from the antenna. The high-frequency signal received by the antenna is given to the third high-frequency transmission line, given to the fifth terminal of the duplexer, given from the sixth terminal of the duplexer to the fifth high-frequency transmission line, and sent to the mixer. Given. In addition, a high-frequency signal generated by the high-frequency oscillator is supplied as a local signal from the third terminal of the branching unit to the mixer via the fourth high-frequency transmission line. The mixer mixes the high-frequency signal generated by the high-frequency oscillator and the high-frequency signal received by the antenna and outputs an intermediate frequency signal, thereby obtaining information contained in the received high-frequency signal.

  The branching device includes two high-frequency switches, and the first high-frequency switch transmits the high-frequency signal between the first terminal and the second terminal by setting the propagation state, and sets the cutoff state. The high-frequency signal is cut off between the first terminal and the second terminal by the second high-frequency switch, and the second high-frequency switch transmits the high-frequency signal between the first terminal and the third terminal by setting the propagation state, and By setting the cut-off state, the high-frequency signal is blocked between the first terminal and the third terminal. When the first high-frequency switch is in the propagation state, the second high-frequency switch is set in the cut-off state, and when the first high-frequency switch is in the cut-off state, the second high-frequency switch is set in the propagation state. The high frequency signal can be selectively output from the second and third terminals. A high ON / OFF ratio can be obtained, and a high-frequency transmitter / receiver with high reliability can be realized by configuring a branching device using a high-reliability high-frequency switch having excellent durability.

  According to the invention, the high-frequency signal generated by the high-frequency oscillator is transmitted to the first high-frequency transmission line and given to the first terminal of the branching device, and given from the second terminal of the branching device to the second high-frequency transmission line. , Given to the fourth terminal of the duplexer, given to the third high-frequency transmission line from the fifth terminal of the duplexer, and radiated from the antenna. The high-frequency signal received by the antenna is given to the third high-frequency transmission line, given to the fifth terminal of the duplexer, given from the sixth terminal of the duplexer to the fifth high-frequency transmission line, and sent to the mixer. Given. In addition, a high-frequency signal generated by the high-frequency oscillator is supplied as a local signal from the third terminal of the branching unit to the mixer via the fourth high-frequency transmission line. The mixer mixes the high-frequency signal generated by the high-frequency oscillator and the high-frequency signal received by the antenna and outputs an intermediate frequency signal, thereby obtaining information contained in the received high-frequency signal. The duplexer includes two high-frequency switches, and the third high-frequency switch transmits the high-frequency signal between the fourth terminal and the fifth terminal by setting the propagation state, and the cutoff state The high frequency signal is cut off between the fourth terminal and the fifth terminal by doing, the fourth high frequency switch transmits the high frequency signal between the fifth terminal and the sixth terminal by being in the propagation state, And by setting it as the said cut-off state, a high frequency signal is interrupted | blocked between the said 5th terminal and the said 6th terminal. When the third high-frequency switch is in the propagation state, the fourth high-frequency switch is set in the cutoff state, and when the third high-frequency switch is in the cutoff state, the fourth high-frequency switch is set in the propagation state, thereby inputting from the fourth terminal. The high frequency signal to be output can be output from the fifth terminal, and the high frequency signal input from the fifth terminal can be output to the sixth terminal. A high ON / OFF ratio can be obtained, and a high-frequency transmitter / receiver with high reliability can be realized by configuring a branching device using a high-reliability high-frequency switch having excellent durability.

  According to the present invention, the high-frequency signal generated by the high-frequency oscillator can be generated by setting all of the high-frequency switches inserted into at least one of the first to third high-frequency transmission lines to the propagation state. Transmitted to the transmission line and supplied to the first terminal of the branching device, supplied from the second terminal of the branching device to the second high-frequency transmission line, and supplied to the fourth terminal of the branching filter. The signal is given from the terminal to the third high-frequency transmission line and radiated from the antenna. In addition, if even one of the high-frequency switches inserted into at least one of the first to third high-frequency transmission lines is cut off, the high-frequency signal generated by the high-frequency oscillator does not pass through the high-frequency switch. It is not radiated from the antenna. By switching the propagation state and cut-off state of the high-frequency switch, a pulse signal wave can be radiated from the antenna. A high-frequency transmitter / receiver with high reliability can be realized by using a high-reliability high-frequency switch excellent in durability and having a large ON / OFF ratio. The high-frequency signal received by the antenna is given to the third high-frequency transmission line, given to the fifth terminal of the duplexer, given from the sixth terminal of the duplexer to the fifth high-frequency transmission line, and sent to the mixer. Given. In addition, a high-frequency signal generated by the high-frequency oscillator is supplied as a local signal from the third terminal of the branching unit to the mixer via the fourth high-frequency transmission line. The mixer mixes the high-frequency signal generated by the high-frequency oscillator and the high-frequency signal received by the antenna and outputs an intermediate frequency signal, thereby obtaining information contained in the received high-frequency signal.

  According to the invention, the duplexer may be formed by a hybrid circuit or a circulator. The hybrid circuit is a directional coupler and is realized by a magic T, a hybrid ring, or a rat race.

  Further, according to the present invention, since the distance detector detects the distance from the high frequency transmitter / receiver to the detection target based on the intermediate frequency signal from the high frequency transmitter / receiver, the distance to the detection target can be accurately detected. A radar device capable of

  According to the present invention, an array antenna apparatus is configured by arranging a plurality of antennas with phase shifters configured by adding the phase shifter to antenna elements. The phase shifter added to each antenna element shifts the phase of the high-frequency signal supplied to the antenna element, thereby adjusting the phase of the radio wave radiated from each antenna element, and the radiation beam from the front of the array antenna. It can be tilted in a predetermined direction. Since the phase shifter is small and can be operated at a low voltage, the array antenna device does not increase in size. In addition, the array antenna apparatus can change the direction of the radiation beam as described above by providing the phase shifter, thereby changing the direction of the radiation beam without mechanically operating the antenna element. It is possible to improve convenience.

  The present invention also includes the array antenna device, and a high-frequency transmitter / receiver connected to the array antenna device, for supplying a high-frequency signal to the array antenna device, and receiving a high-frequency signal captured by the array antenna device. Therefore, the radar apparatus can be easily changed without increasing the size of the radar apparatus, and a highly convenient radar apparatus can be realized.

  1 to 3 are cross-sectional views schematically showing the configuration of a phase shifter 1 according to an embodiment of the present invention. FIGS. 1 and 2 are cross-sections perpendicular to the extending direction X of the dielectric line 6. FIG. 3 is a cross-sectional view in the center in the stacking direction Z, which is parallel to the extending direction X. FIG. 1 is a cross-sectional view taken along section line II shown in FIG. 3, and FIG. 2 is a cross-sectional view taken along section line II-II shown in FIG. FIG. 4 is a cross-sectional view taken along section line IV-IV in FIG. In FIG. 4, for easy understanding, the inter-plate dielectric portion 9 and the second dielectric portion 15 are omitted. FIG. 5 is a plan view of the phase shifter 1.

  The phase shifter 1 includes a non-radiative dielectric line (NRD guide) 2, a planar line 3, first and second electrodes 4 a and 4 b, and a mounting part 5.

  The non-radiative dielectric line 2 is a pair of flat conductor portions provided between the dielectric line 6 through which the electromagnetic wave propagates and the dielectric line 6, that is, in contact with the dielectric line 6 on both sides of the dielectric line 6. 7a and 7b. Hereinafter, the pair of flat plate conductor portions 7a and 7b are referred to as first and second flat plate conductor portions 7a and 7b, respectively.

  The dielectric line 6 extends in a predetermined direction. The dielectric line 6 has a rectangular cross section perpendicular to the extending direction X of the dielectric line 6. The cross-sectional shape of the dielectric line 6 may be slightly deviated from a complete rectangle, but at least the surfaces in contact with the first and second plate conductor portions 7a and 7b are as flat as possible and have high parallelism. Is desirable. In the present embodiment, the dielectric line 6 is formed in a rectangular parallelepiped shape.

The dielectric line 6 is made of a dielectric, and is formed to include a changing portion whose dielectric constant changes according to an applied electric field. In the present embodiment, the dielectric line 6 is composed of a changing portion, for example, Ba _(1-x) Sr _x TiO ₃ (strontium barium titanate, abbreviated as BST), Mg _(1-x) Ca _x TiO ₃ , Zn _{( 1-x)} Sn _x TiO ₃ , BaO—PbO—Nd ₂ O ₃ —TiO ₃ , SrBi ₂ Ta ₂ O ₉ or Bi _1.5 Zn _1.0 Nb _1.5 O ₇ or the like. The dielectric line 6 has a lower dielectric constant as the applied electric field increases, that is, as the applied electric field strength increases.

  The first and second flat plate conductor portions 7a and 7b are formed in a plate shape and are provided with a predetermined interval L1. In the present embodiment, the first and second flat plate conductor portions 7a and 7b are formed to have the same size, and are provided so that their thickness directions coincide with each other and face each other. The surfaces of the first and second flat plate conductor portions 7a and 7b facing each other are formed in a plane. The first and second flat plate conductor portions 7a and 7b are arranged in parallel as much as possible so that the surfaces of the first and second flat plate conductor portions 7a and 7b facing each other are in close contact with the dielectric line 6. Is preferred.

  The predetermined interval L1 is the first and second flat plate in the portion of the electromagnetic wave propagating through the dielectric line 6 between the first and second flat plate conductor portions 7a and 7b, excluding the dielectric line 6. The electromagnetic waves polarized in parallel to the first and second flat plate conductor portions 7a and 7b are cut off from between the conductor portions 7a and 7b. That is, in the region between the first and second flat plate conductor portions 7a and 7b of the electromagnetic wave propagating through the dielectric line 6, in the portion excluding the dielectric line 6, that is, the inter-plate dielectric portion 9 described later. It is selected to be less than or equal to half the wavelength of the propagating electromagnetic wave.

  The width of the dielectric line 6 is selected so that the cutoff frequency of the LSE mode is less than the frequency of the electromagnetic wave to be transmitted, that is, the operating frequency. The dielectric line 6 is formed in a rectangular shape in which a cross section perpendicular to the extending direction X has a short side in contact with the first and second flat plate conductor portions 7a and 7b. The length L2 of the long side in the rectangular cross section of the dielectric line 6 is equal to the predetermined interval L1. The ratio of the length L2 of the long side to the length L3 of the short side in the rectangular cross section of the dielectric line 6 is reduced by decreasing the length L3 of the short side or increasing the length L2 of the long side. Thus, the LSM mode is increased until the LSM mode is cut off and only the LSE mode propagates. The ratio of the long side length L2 to the short side length L3 in the rectangular cross section of the dielectric line 6, ie, L3 / L2, is preferably as large as possible so that the LSE mode is near the cutoff. It is desirable to set to propagate.

The first and second flat conductor portions 7a and 7b are formed of a good conductor having a volume resistivity of 10 ⁻⁴ Ω · m or less, preferably 10 ⁻⁵ Ω · m or less. For example, gold (Au), copper (Cu ), Aluminum (Al), platinum (Pt), titanium (Ti), silver (Ag), palladium (Pd), zinc (Zn) and chromium (Cr), or the group An alloy containing at least two selected from the above or a laminate thereof, or ITO
It is formed of an oxide conductor such as (Indium Tin Oxide), tin oxide, iridium oxide, or SrRuO ₃ .

  The first and second flat plate conductor portions 7a and 7b are perpendicular to the extending direction X and the stacking direction Z in which the first and second flat plate conductor portions 7a and 7b and the dielectric line 6 are stacked. In the width direction Y, the dielectric line 6 extends from the end in the width direction Y to a position of a predetermined distance L4 outward. The predetermined distance L4 is selected to be larger than the length at which the electromagnetic wave attenuation becomes 1/10.

  The first and second electrodes 4 a and 4 b are provided to apply an electric field to the changing portion, that is, to apply an electric field to the dielectric line 6. The first and second electrodes 4a and 4b are provided along the dielectric line 6, and are spaced in the width direction Y smaller than the distance L1 between the first and second plate conductor portions 7a and 7b. In such a state, the dielectric line 6 is interposed therebetween. The first and second electrodes 4 a and 4 b are provided in contact with the dielectric line 6, and are provided by being laminated on both end surfaces of the dielectric line 6 in the width direction Y, respectively. The first and second electrodes 4a and 4b are provided so as to overlap in the width direction Y, and a portion of the first and second electrodes 4a and 4b overlapping in the width direction Y has a predetermined length L6 in the extending direction X. Formed. The predetermined length L6 is selected so as to obtain a necessary phase change.

  The first and second electrodes 4a and 4b are formed apart from the first and second flat plate conductor portions 7a and 7b. The first and second electrodes 4a and 4b have a rectangular parallelepiped shape and are formed in a plate shape.

  In the stacking direction Z, the first and second electrodes 4a and 4b are provided on the first and second flat plate conductor portions 7a and 7b at a predetermined distance L7, respectively. The predetermined distance L7 is such that the first and second electrodes 4a and 4b do not contact the first and second plate conductor portions 7a and 7b, and is selected from 1 μm to 100 μm, for example.

  The first and second electrodes 4a and 4b are formed of the same material as the first and second flat plate conductor portions 7a and 7b described above, or silicon (Si), germanium (Ge) and gallium arsenide (GaAs). ) Or a high resistance material such as tantalum nitride and NiCr alloy.

  The thickness L8 of the first and second electrodes 4a and 4b is selected to be less than the skin thickness with respect to the electromagnetic wave to be propagated through the dielectric line 6. When the skin thickness is “δ”, the magnetic permeability is “μ”, the electrical conductivity is “σ”, and the angular frequency is “ω”, the skin thickness is expressed by Equation (1). Note that ω = 2πf (f is a frequency). When electromagnetic waves enter the conductor, the amplitude becomes 1 / e at the skin thickness.

  Accordingly, even if the first and second electrodes 4a and 4b are provided in contact with the dielectric line 6, the electromagnetic wave propagating through the dielectric line 6 can pass through the first and second electrodes 4a and 4b. Therefore, the electromagnetic waves can be propagated without being cut off, and the first and second electrodes can be transmitted in a state in which transmission loss generated when the first and second electrodes 4a and 4b are brought close to the dielectric line 6 is suppressed. An electric field having a large electric field strength can be applied to the dielectric line 6 by the electrodes 4a and 4b, and the phase of the electromagnetic wave can be changed stably. Since the phase shifter 1 has a characteristic that the phase change with respect to the frequency is large in the vicinity of the cutoff of the LSE mode propagating through the non-radiative dielectric line 2, if this characteristic is used, the phase shifter 1 is provided on both sides of the dielectric line 6. The phase of the LSE mode high-frequency signal propagating through the dielectric line 6 can be greatly changed by the first and second electrodes 4 a and 4 b sandwiching the dielectric line 6. Therefore, since a sufficient amount of phase change can be obtained with a low drive voltage even if the line length is short, a small phase shifter capable of good phase control is realized.

The volume resistivity of the first and second electrodes 4a and 4b in the present embodiment is selected to be 10 ⁻⁵ Ω · m or more, preferably 10 ⁻⁴ Ω · m or more. However, if the first and second electrodes 4a and 4b are made too thin, electric charges are difficult to move in the first and second electrodes 4a and 4b, and an electric field is uniformly distributed over the entire first and second electrodes 4a and 4b. Therefore, the first and second electrodes 4a and 4b do not hinder the movement of charges in the first and second electrodes 4a and 4b, and the entire first and second electrodes 4a and 4b are not disturbed. It is formed to have a predetermined thickness or more so that an electric field can be applied uniformly.

The resistivity of the first and second electrodes 4a and 4b is preferably selected to be 10 ⁻⁵ Ω · m or more and 10 ⁸ Ω · m or less. If the resistivity of the first and second electrodes 4a and 4b is less than 10 ⁻⁵ Ω · m, the electromagnetic wave in the electrode is greatly attenuated and loss is increased, which is not preferable. If the resistivity of the first and second electrodes 4a and 4b is further smaller than 10 ⁻⁵ Ω · m, the desired mode is cut off and does not propagate. Conversely, if the resistivity of the first and second electrodes 4a and 4b exceeds 10 ⁸ Ω · m and becomes too large, the difference in resistivity with the dielectric sandwiched between the first and second electrodes 4a and 4b is small. Therefore, a desired voltage cannot be applied to the dielectric due to the voltage drop.

The thickness of the first and second electrodes 4a and 4b is determined by the resistivity of the material used for the first and second electrodes 4a and 4b. If the thickness is too thick, the loss increases, and if the thickness is further increased, the desired mode is cut off. Will not be transmitted. If it is too thin, a desired voltage cannot be applied to the dielectric due to a voltage drop. For example, the first and second electrodes 4a and 4b have a resistivity of 1 × 10 ⁻⁴ (Ω · m) (assuming TaN is used as the material) and a resistivity of 1 × 10 ⁻³ (Ω · m). Table 1 shows the loss due to the electrode per 1 mm with respect to the electromagnetic wave of 77 GHz when the electromagnetic field analysis is performed.

From the results of electromagnetic field analysis shown in Table 1 and Table 2, it is preferable that the electrode has a resistivity of 1 × 10 ⁻⁴ (Ω · m) and the electrode thickness is practically 30 nm or less. Is practically preferable to be 320 nm or less of the electrode in the case where the resistivity is 1 × 10 ⁻³ (Ω · m). Here, the standard that is practically preferable is 3 dB loss.

The non-radiative dielectric line 2 is provided between the first and second flat plate conductor portions 7a and 7b, and sandwiches the dielectric line 6 and the first and second electrodes 4a and 4b. The inter-plate dielectric portion 9 which is a third dielectric portion provided with the first and second electrodes 4a and 4b interposed therebetween is configured. The inter-plate dielectric portion 9 is formed of a dielectric having a low dielectric constant, and is formed of glass, single crystal, ceramics, or resin. As glass, quartz glass, crystallized glass, or the like is used. As the single crystal, quartz, sapphire, MgO, LaAlO ₃ or the like is used. As the ceramic, alumina, forsterite, cordierite, or the like is used. As the resin, an epoxy resin, a fluorine-containing resin, a liquid crystal polymer, or the like is used.

  The inter-plate dielectric part 9 is provided on both sides of the dielectric line 6 in the width direction Y between the first and second flat conductor parts 7a and 7b. Of the region sandwiched between the flat conductor portions 7a and 7b, the remaining region excluding the region where the dielectric line 6 and the first and second electrodes 4a and 4b are formed is buried.

  The inter-plate dielectric part 9 has a dielectric constant lower than the dielectric constant of the dielectric line 6. The inter-plate dielectric portion 9 has the highest dielectric constant among the dielectric lines 6 when an electric field is applied to the dielectric line 6 and when no electric field is applied to the dielectric line 6. Has a lower dielectric constant than the dielectric constant of the lower portion. The inter-plate dielectric portion 9 is provided in contact with the first and second plate conductor portions 7a and 7b, the dielectric line 6, and the first and second electrodes 4a and 4b. By providing the inter-plate dielectric part 9, the wavelength of the electromagnetic wave propagating in the part excluding the dielectric line 6 among the parts sandwiched between the first and second flat conductor parts 7a and 7b is changed to the wavelength in the air. As a result, the non-radiative dielectric line 2 can be formed in a small size. Further, since the first and second flat plate conductor portions 7a and 7b are mechanically supported by the inter-plate dielectric portion 9, the mechanical strength can be improved. In the width direction Y, the thickness in the Y direction of each part provided on both sides of the dielectric line 6 in the inter-plate dielectric part 9 is selected equally. If one of the portions of the inter-plate dielectric portion 9 sandwiching the dielectric line 6 is thicker than the other and the symmetry is lost, the LSE mode propagation characteristics may be adversely affected. It is preferable to have a structure having symmetry with respect to each other.

  The planar line 3 includes a first strip conductor part 11, a strip dielectric part 12 provided with the first strip conductor part 11, and a first ground conductor part 13. The end surface of the non-radiative dielectric line 3 in the propagation direction of the electromagnetic wave (the extending direction X of the line) and the end surface of the planar line 3 in the propagation direction of the electromagnetic wave are brought into contact with each other. 4 are combined. The planar lines 3 are respectively provided on both sides of the nonradiative dielectric line 2 in the extending direction X.

  The first strip conductor portion 11 and the first ground conductor portion 13 are provided at a predetermined interval L5 in the direction perpendicular to the direction in which the dielectric line 6 and the plate conductor portions 7a and 7b overlap, that is, in the width direction Y. It is done. In the present embodiment, the first ground conductor portions 13 are provided on both sides of the first strip conductor portion 11 in the width direction Y, respectively. One of the first ground conductor portions 13 may be referred to as a first ground conductor portion 13a, and the other may be referred to as a first ground conductor portion 13b.

  The first ground conductor portion 13 is provided along the first strip conductor portion 11 in parallel with the first strip conductor portion 11. The first strip conductor portion 11 and the first ground conductor portion 13 are formed of the same material as the first and second flat plate conductor portions 7a and 7b described above.

  The strip dielectric portion 12 includes a first dielectric portion 14 provided with the first strip conductor portion 11 embedded therein, and a second dielectric provided between the first dielectric portion 14 and the first ground conductor portion 13. And a body part 15. The first dielectric portion 14 is provided continuously with the dielectric line 6 and is formed of the same material as the dielectric line 6. The second dielectric portion 15 is provided continuously with the inter-plate dielectric portion 9 and is formed of the same material as the inter-plate dielectric portion 9. As a result, the portions made of the same substance are in contact with each other at the connection portion between the non-radiative dielectric line 2 and the planar line 3, so that a connection structure with small reflection of electromagnetic waves in the extending direction X can be obtained. The strip dielectric portion 12 has both sides in the width direction Y formed flat, and has a rectangular parallelepiped shape in the embodiment of the present invention.

  The first strip conductor portion 11 is provided at the center in the width direction Y and the stacking direction Z of the first dielectric portion 14. The first strip conductor portion 11 has a rectangular parallelepiped shape and extends along the extending direction X. On the surface portion 16 in the width direction Y of the strip dielectric portion 12, a first ground conductor portion 13 is formed over the entire surface portion 16.

  Of the end faces in the extending direction X of the first strip conductor portion 11, the end face 18 facing the non-radiating dielectric line 3 and the end face in the extending direction X of the dielectric line 6 are abutted to each other, and the non-radiating dielectric line 2 and the planar line 3 are coupled. The planar line 3 is coupled to the LSE mode of the non-radiative dielectric line 2. The center of the end face 18 of the first strip conductor portion 11 continues to the center of the dielectric line 6. The dimension in the stacking direction Z of the strip dielectric portion 12 is selected to be equal to the length between the first and second flat plate conductor portions 8a and 8b.

  The first strip conductor portion 11 includes a first conductor portion 19 on the nonradiative dielectric line 2 side in the extending direction X, a second conductor portion 21 on the opposite side of the nonradiative dielectric line 2, and a first conductor portion. 19 and a third conductor portion 21 that connects the second conductor portion 20 to each other. The length L9 of the first conductor portion 19 in the stacking direction Z is selected to be equal to the length L10 of the first and second electrodes 4a and 4b in the stacking direction Z. The length L11 of the second conductor portion 20 in the stacking direction Z is selected to be smaller than the length L9. The centers in the stacking direction Z of the first and second conductor portions 19 and 20 are provided so as to be aligned with the extending direction X. The third conductor portion 21 is formed in a tapered shape in which the length in the stacking direction Z gradually decreases from the first conductor portion 19 toward the second conductor portion 20. By providing the third conductor portion 21, electromagnetic waves can be smoothly propagated to the planar line 3 including the first strip conductor portions 11 having different lengths in the stacking direction Z. The first strip conductor portion 11 is not formed in a region sandwiched between the first and second electrodes 4a and 4b, but is provided slightly apart from the region in the extending direction X. If the first strip conductor portion 11 is in a region sandwiched between the first and second electrodes 4a and 4b, the relative dielectric is caused by the electric field in contact with the strip conductor portion 11 when a voltage is applied to the first and second electrodes 4a and 4b. The dielectric having a variable rate, that is, the relative dielectric constant of the first dielectric portion 14 is changed to change the characteristic impedance, and the non-radiative dielectric line 2 and the planar line 3 are in a mismatched state. As a result, reflection occurs and transmission loss occurs. By providing the first strip conductor portion 11 slightly spaced from the region in the extending direction X, such transmission loss can be suppressed.

  The length L10 of the first and second electrodes 4a and 4b needs to be equal to or longer than the length L9 of the second strip conductor portion 42, and is selected to be shorter than the length of the strip dielectric portion 12 in the stacking direction Z. Is selected to be less than or equal to the wavelength λ of the high frequency signal to be transmitted.

  The second electrode 4 b is formed to extend between the first and second dielectric portions 14 and 15 of the planar line 3. The second electrode 4 b is formed across the non-radiative dielectric line 2 and the end of the planar line 3 opposite to the non-radiative dielectric line 2. By forming the second electrode portion 4b also on the planar line 3, the portion made of the same material can be continued in the extending direction at the interface at the connection portion between the planar line 3 and the nonradiative dielectric line 2. it can. Thereby, reflection of electromagnetic waves between the non-radiative dielectric line 2 and the planar line 3 can be reduced. Even if the second electrode 4b is formed on the planar line 3, the thickness of the second electrode 4b is selected as described above, so that the characteristic impedance of the planar line 3 due to the provision of the second electrode 4b can be changed. It can be reduced as much as possible.

  The strip conductor portion 13 and the dielectric are arranged such that the longitudinal direction in the cross section perpendicular to the extending direction X of the first strip conductor portion 11 coincides with the longitudinal direction in the cross section perpendicular to the extending direction X of the dielectric line 6. The track 6 is connected. Thereby, the contact between the first strip conductor portion 11 and the second electrode 4b described above can be avoided, and the degree of freedom in designing the first strip conductor portion 11 can be improved. The lengths L5, L9, L11 and the thickness L12 of the first strip conductor portion 11 in the width direction Y are selected so that the characteristic impedance of the planar line 3 matches the characteristic impedance of the nonradiative dielectric line 3.

  A second ground conductor portion 23 connected to the first ground conductor portion 13 is provided on the surface portion in the width direction Y of the non-radiative dielectric line 2. The second ground conductor portion 23 is provided in contact with the inter-plate dielectric portion 9. The second ground conductor portion 23 is formed of the same material as the first and second flat plate conductor portions 7 a and 7 b and has the same thickness as the first ground conductor portion 13. The second ground conductor portion 23 is formed continuously with the first and second flat plate conductor portions 7a and 7b. The first and second ground conductor portions 13 and 23 are continuous in the extending direction X, and the ground conductor portion including the first and second ground conductor portions 13 and 23 is formed in a plate shape.

  On the surface portion of the planar line 3 in the stacking direction Z, a planar line flat plate conductor portion 24 connected to the first and second flat plate conductor portions 7a and 7b is provided in contact with the strip dielectric portion 12. The planar line flat conductor portion 24 is made of the same material as the first and second flat conductor portions 7a and 7b, and has the same thickness as the first and second flat conductor portions 7a and 7b. The planar line flat conductor portion 24 is formed continuously with the first ground conductor portion 14. The first and second flat plate conductor portions 7a and 7b and the flat line flat plate conductor portion 24 are continuous in the extending direction X, and the first and second flat plate conductor portions 7a and 7b and the flat line flat plate conductor portion 24 are connected. The conductor part which consists of is formed in plate shape.

  An end conductor portion 25 is formed in contact with the strip dielectric portion 12 at the end of the planar line 3 opposite to the nonradiative dielectric line 2 in the extending direction. The end conductor portion 25 is formed continuously to the first ground conductor portion 25 and the planar line flat conductor portion 24. Therefore, the first and second flat conductor portions 7a and 7b, the first and second ground conductor portions 13 and 23, and the flat line flat plate conductor portion 24 are integrally formed. It has a cylindrical shape whose cross section perpendicular to the extending direction Z is rectangular. The end conductor portion 25 is provided so as to block the end portion in the extending direction Z of the line conductor portion having the cylindrical shape. The end conductor portion 25 prevents the electromagnetic wave from leaking in the extending direction X from the end portion of the planar line 3 opposite to the nonradiative dielectric line 2.

  The mounting portion 5 includes a mounting dielectric portion 41 that is a second dielectric portion, a second strip conductor portion 42, a mounting ground conductor portion 43, and a mounting electrode connection portion 44.

  The mounting dielectric portion 41 is connected to the surface portion 45 of the first ground conductor portion 13a on the side opposite to the first strip conductor portion 11 and the first ground conductor portion 13a. The two ground conductors 23 are formed on the surface 46 opposite to the nonradiative dielectric line 2. The mounting dielectric portion 41 is provided by being laminated on the surface portions 45 and 46, and is formed over the entire area of the surface portions 45 and 46.

The mounting dielectric part 41 is formed in a rectangular parallelepiped shape. As the material of the mounting dielectric portion 41, a material having high insulation and a low dielectric loss tangent to a high frequency signal is suitable. The mounting dielectric portion 41 is formed of, for example, silicon dioxide (SiO ₂ ), alumina (Al ₂ O ₃ ), aluminum nitride (AlN), magnesium oxide (MgO), or the like. The mounting dielectric portion 41 is formed to a predetermined thickness L13.

  The second strip conductor part 42 is provided on the surface part 50 of the mounting dielectric part 41 opposite to the first flat plate conductor part 7a. The second strip conductor portion 42 is provided so as to be laminated on the surface portion 50, and is formed over the entire region of the surface portion 50.

  The second strip conductor portion 42 is formed in a rectangular parallelepiped shape. The second strip conductor portion 42 is formed of the same material as the first strip conductor portion 11 described above.

  The second strip conductor portion 42 is provided so as to at least partially overlap the first strip conductor portion 11 in the stacking direction Z. The second strip conductor portion 42 extends parallel to the extending direction X. The second strip conductor portion 42 is formed to a predetermined thickness L14. The length L15 of the second strip conductor portion 42 in the stacking direction Z is selected so that the matching state with a high-frequency signal in a through hole 49 described later is good. In the case of a two-port circuit, the matching state means that the reflection is small and the transmission amount is secured. Quantitatively, if reflection is −10 dB or less, about 90% of the transmitted electromagnetic wave does not return, so it can be said that the matching state is good. Further, the length L15 can mount a bump used for surface mounting, that is, the bump does not protrude from the second strip conductor portion 42 when the bump is mounted on the second strip conductor portion 42. Chosen.

  The second strip conductor part 42 includes a second strip conductor part 42a provided on one end part side in the extending direction X and a second strip conductor part 42b provided on the other end part side in the extending direction X. The second strip conductor portions 42a and 42b are provided independently. The second strip conductor portion 42a is provided so as to partially overlap the first strip conductor portion 11a of the planar line 3a provided on one side in the extending direction X of the non-radiative dielectric line 2. The second strip conductor portion The part 42 b is provided so as to partially overlap the first strip conductor part 11 b of the planar line 3 b provided on the other side in the extending direction X of the non-radiative dielectric line 2.

  The second strip conductor portion 42 and the first strip conductor portion 11 are formed so as to overlap each other by a predetermined length L16 or more in the extending direction X, respectively. The predetermined length L16 is selected to be greater than or equal to the dimension in the extending direction X of the through hole 49 described later, and is a length exceeding λ / 4 when the wavelength of the propagating electromagnetic wave is λ, that is, L16 = λ / 4 + α [ mm] is preferred.

  A through hole 49 is formed in the first ground conductor portion 13 at a position where the first strip conductor portion 11 and the second strip conductor portion 42 overlap. The through hole 49 is formed through the first ground conductor portion 13 in the width direction Y. The through hole 49 is formed such that a cross section perpendicular to the width direction Y of the inner peripheral surface 51 of the first ground conductor portion 13 facing the through hole 49 is rectangular. The inner peripheral surface 51 is parallel to the stacking direction Z and the width direction Y or the extending direction X.

  The through hole 49 passes through the center of the through hole 49 in the extending direction X, and an axis parallel to the stacking direction Z in the width direction Y exists between both end portions in the extending direction X of the first strip conductor portion 11. And is formed so as to pass through the center in the stacking direction Z of the phase shifter 1. The length in the extending direction X of the second strip conductor portion 42 is extended from a virtual plane that passes through the center of the extending direction X of the through hole 49 and is perpendicular to the extending direction X in order to achieve alignment. The frequency is selected to be at least one-fourth of the wavelength λ of the high-frequency signal propagating to X.

  The through hole 49 is formed at the end 35 on the opposite side to the nonradiative dielectric line 2 in the extending direction X. A through hole 49 is provided in the end portion 35, and a bump position is provided in the end portion 36 on the nonradiative dielectric line 2 side of the second strip conductor portion 42 in the extending direction X, so that the second direction in the extending direction X is the second. The phase shifter 1 can be made smaller than the configuration in which the bump position is provided at the end portion 35 of the strip conductor portion 42 opposite to the nonradiative dielectric line 2.

  The through hole 49 constitutes a coupling portion that couples a high-frequency signal between the planar line 3 and the microstrip line including the second strip conductor portion 42. The through hole 49 is formed at a position where the electric field of the standing wave in the waveguide mode (TEM mode) in the planar line 3 is maximized. In the connection portion between the planar line 3 and the microstrip line including the second strip conductor part 42, the through-hole 49 to the planar line 3 as viewed from the high-frequency signal input port of the microstrip line having the second strip conductor part 42. It is desirable to have a structure that matches the high-frequency signal in terms of the propagation characteristics, and that the reflection characteristics are −10 dB or less and that the connection loss can be suppressed as low as possible. In the present embodiment, a virtual one plane including the central axis extending in the width direction Y of the through hole 49 and perpendicular to the extending direction X and the nonradiative dielectric line 2 of the first strip conductor portion 11 are The distance L17 to the opposite end face is selected to be (2n-1) / 4 (n is a natural number) of the wavelength of the electromagnetic wave propagating through the planar line 3. Further, the virtual one plane that includes the central axis extending in the width direction Y of the through hole 49 and is perpendicular to the extending direction X, and the nonradiative dielectric line 2 that is the open end of the second strip conductor portion 42 are: The distance L18 between the opposite end face is (2m-1) / 4 (m is a natural number) of the wavelength of the electromagnetic wave propagating between the second strip conductor portion 42 and the first and second ground conductor portions 13 and 23. Chosen. As a result, connection loss can be suppressed.

  The inner dimension L19 in the extending direction X of the inner peripheral surface 21 facing the through hole 49 and the inner dimension L20 in the width direction Y are determined so that the input impedance viewed from the strip conductor can be matched. The

  With the configuration as described above, the high frequency signal to be phase-controlled from the second strip conductor portion 42 on the input side is coupled to the dielectric line 6 of the non-radiative dielectric line 2 and propagated through the dielectric line 6. After controlling the phase of the high-frequency signal, the phase-controlled high-frequency signal is coupled to the output-side second strip conductor portion 42, and the phase-controlled high-frequency signal is taken out from the output-side second strip conductor portion 42. It is.

  The mounting ground conductor portion 43 and the mounting electrode connection portion 44 are provided on the surface portion 50 of the mounting dielectric portion 41. The mounting ground conductor 43 is provided so as to be in non-contact with the second strip conductor 42 and surround the second strip conductor 42, and is provided at a predetermined distance L21 from the second strip conductor 42 in the width direction Y. In addition, the second strip conductor portion 42 is formed at a predetermined distance L22 in the extending direction.

  The mounting electrode connection portion 44 is provided in a non-contact manner on the second strip conductor portion 42 and the mounting ground conductor portion 43. A mounting ground conductor portion 43 is provided between the mounting electrode connection portion 44 and the second strip conductor portion 42. Thus, when a voltage is applied to the mounting electrode connecting portion 44, application of an electric field to the second strip conductor portion 42 is prevented, and a change in impedance in the planar line including the second strip conductor portion 42 is prevented. The second strip conductor portion 44 and the mounting electrode connection portion 44 can be provided on the same surface portion 50.

  The mounting electrode connecting portion 44 is provided at the center in the extending direction X of the mounting dielectric portion 41, and is connected to the bump connecting portion 44a to which the bump is connected during surface mounting, and the bump connecting portion 44a is mounted. The dielectric portion 41 has an extending portion 44 b provided in the peripheral portion and extending in the extending direction X. A pair of mounting electrode connection portions 44 are provided so as to be plane-symmetric with respect to a virtual plane that is perpendicular to the width direction Y of the phase shifter 1 and passes through the center of the width direction Y. The mounting electrode connection portions 44 are provided apart from each other in the width direction Y, and a mounting ground conductor portion 43 is provided between the mounting electrode connection portions 44.

  The mounting portion 5 functions as a high-frequency signal transmission line between the second strip conductor portion 42 and the first and second ground conductor portions 13 and 23. Therefore, the thickness L13 of the mounting dielectric portion 41, the thickness L14 of the second strip conductor portion 42, the predetermined distances L21 and L22, and the dielectric constant of the mounting dielectric portion 41 are matched with the specific impedance of the planar line 3. Further, the characteristic impedance is adjusted in advance.

  An electrode connection line 52 is connected to the first electrode 4a. The electrode connection line 52 is electrically connected to one end of the first electrode 4a in the extending direction X, extends to one side of the extending direction X, and branches in the stacking direction Y at one end of the extending direction X. And it is formed between the both ends of the lamination direction Y, and is formed in T shape. The electrode connection line 52 is formed between the first dielectric portion 14 and the second dielectric portion 15. The electrode connection line 52 can be formed in the same film formation process as the first electrode 4a. The electrode connection line 52 is made of the same material as the first electrode 4a and has the same thickness. The line width of the electrode connection line 52, that is, the length in the stacking direction Z in the portion 52a extending in the extending direction X, and the length L23 in the extending direction X in the portion 52b extending in the stacking direction Z is the first strip conductor. The portion 11 is selected to be equal to or shorter than the length in the stacking direction Z in the portion where the length in the stacking direction Z is the smallest. A portion 52a extending in the extending direction X of the electrode connection line 52a overlaps the first strip conductor portion 11, and is provided with the center in the width direction Y aligned on a virtual plane perpendicular to the width direction. Thereby, it is possible to suppress as much as possible that the electric field is formed between the second electrode 4b and the electrode connection portion 52 and the characteristic impedance of the planar line 3 is changed.

  First penetrating conductor portions 54 are respectively provided at both end portions 53 in the stacking direction Z of the portion 52 b extending in the stacking direction Z of the electrode connection portion 52. The first through conductor portion 54 is provided in a via hole formed in the second dielectric portion 15, the first ground conductor portion 13, and the mounting dielectric portion 41. The first through conductor portion 54 is electrically connected to the electrode connecting portion 52 b and the extending portion 44 b of the mounting electrode connecting portion 44. The first through conductor portion 54 is formed of the same material as the mounting electrode connection portion 44. The first through conductor portion 54 is provided in a non-contact manner with the first ground conductor portion 13.

  The end of the planar line 3 opposite to the nonradiative dielectric line 2 is in contact with and electrically connected to the mounting ground conductor 43, the second ground conductor 23, and the second electrode 4b. A second through conductor portion 55 is provided. The second through conductor portion 55 is provided in a via hole formed in the first dielectric portion 14, the second dielectric portion 15, the first ground conductor portion 13 and the mounting dielectric portion 41. The second through conductor portion 55 is formed of the same material as the mounting ground conductor portion 43. The second through-hole conductor portions 55 are provided at both end portions in the stacking direction Z, respectively. The second through conductor 55 is formed closer to the end opposite to the non-radiative dielectric line 2 than the through hole 49 so that the influence on the characteristic impedance of the planar line 3 is reduced. The first through conductor portion 54 is provided on the planar line 3a on one side of the nonradiative dielectric line 2 in the extending direction X, and the second through conductor portion 55 is on the other side of the nonradiative dielectric line 2 in the extending direction X. Is provided on the planar line 3b.

  The mounting electrode connecting portion 44 and the first electrode 4a can be electrically connected by the first through conductor portion 54, and the mounting ground conductor portion 43 and the second electrode 4b can be electrically connected by the second through conductor portion 54. Therefore, when the phase shifter 1 is mounted on a planar circuit board such as a printed circuit board, a potential can be applied to the first electrode 23a via the mounting electrode connecting portion 44, and mounting grounding can be performed. A potential can be applied to the second electrode 4b through the conductor portion 43, and the mountability can be improved. The mounting electrode connecting portion 44 and the first through conductor portion 54 constitute a first electrode connecting portion, and the mounting ground conductor portion 43 and the second through conductor portion 54 constitute a second electrode connecting portion.

  FIG. 6 is a plan view of the planar circuit board 56 on which the phase shifter 1 is mounted. The phase shifter 1 mounts the second strip conductor portion 42, the mounting ground conductor portion 43, and the mounting electrode connection portion 44 on the planar circuit board 56 through bumps or the like so as to face one surface 56a of the planar circuit board 56. Is done. The mounting ground conductor portion 43 and the second through conductor portion 54 function as ground conductors for the first and second ground conductor portions 13 and 23, and are terminals for applying a voltage to the first and second electrodes 4a and 4b. Also works. By providing the mounting ground conductor portion 43 and the mounting electrode connection portion 44, it is possible to easily connect to the substrate wiring 57 formed on the planar circuit substrate 56 to be surface-mounted through bumps or the like.

  On the planar circuit board 56, a first pad 58a is formed at a position facing the end of the second strip conductor portion 42 on the nonradiative dielectric line 2 side, and a second pad 58b is formed at a position facing the bump connection portion 44a. Is provided. The first and second pads 58a and 58b are individually connected to the substrate wiring 57. The substrate wiring 57 connected to the first pad 58a is provided with a third pad 58c at a position facing the mounting ground conductor 43 on both sides in the extending direction. The third pad 59 is grounded.

  The phase shifter 1 further includes a voltage applying unit 23. The voltage applying means 23 is realized by an electric circuit that applies a voltage in a predetermined range between the pair of first and second electrodes 4a and 4b. The voltage application means 23 is electrically connected to the mounting ground conductor 43 and the mounting electrode connection 44, and applies a predetermined potential to the mounting ground conductor 43 and the mounting electrode connection 44, whereby the first and second voltages are applied. A voltage is applied between the electrodes 4a and 4b. The potential applied to the ground conductor portion 43 is a ground potential. Thus, an electric field is applied to the dielectric line 6 sandwiched between the first and second electrodes 4a and 4b. The voltage applying unit 23 includes, for example, a voltage divider, and applies the voltage divided by the voltage divider to the first and second electrodes 4a and 4b. The voltage application means 23 can apply a plurality of levels of voltages to the mounting ground conductor 43 and the mounting electrode connection 44. The voltage applying means 23 applies an AC voltage having a frequency lower than the frequency of the propagating electromagnetic wave or a DC voltage to the mounting ground conductor 43 and the mounting electrode connection 44. The voltage applying unit 23 applies a voltage corresponding to the phase amount to be shifted to the mounting ground conductor 43 and the mounting electrode connection 44.

  The voltage application means 23 applies a voltage between the first and second electrodes 4a and 4b, and changes the magnitude of the applied voltage within a predetermined range, thereby allowing the phase of the electromagnetic wave guided through the dielectric line 6 to vary. Can be changed according to the magnitude of the applied voltage, that is, the magnitude of the applied electric field. The dielectric forming the dielectric line 6 has a lower dielectric constant when the applied electric field is increased, thereby changing the phase of the electromagnetic wave guided through the dielectric line 6.

  The cutoff frequency fc of the non-radiative dielectric line 2 in the phase shifter 1 is the dielectric constant of the dielectric forming the dielectric line 6 and the size of the dielectric line 6 (the dimension of the cross section perpendicular to the extending direction X). The distance (L3) between the first and second electrodes 4a and 4b, the distance L1 between the first and second flat plate conductor portions 7a and 7b, and the dielectric constant of the dielectric forming the interplate dielectric portion 9 are determined. The size of the dielectric line 6 is selected so that the cut-off frequency is less than the frequency (use frequency) of the electromagnetic wave to be propagated. When a predetermined voltage is applied to the first and second electrodes 4a and 4b and the dielectric constant of the dielectric line 6 decreases, the cut-off frequency is fc, and the use frequency, that is, the electromagnetic wave propagating through the dielectric line 6 is transmitted. Assuming that the frequency is f, 1.03 <f / fc <1.5, preferably 1.03 <f / fc <1.2. The distance (L3) between the second electrodes 4a and 4b, the distance L1 between the first and second flat-plate conductor portions 7a and 7b, and the dielectric that forms the inter-plate dielectric portion 9 are set. When the phase shifter 1 is manufactured, firstly, the dielectric that forms the dielectric line 6 and the inter-plate dielectric part 9 is determined, and then the size of the dielectric line 6 is determined. An interval L4 between the second electrodes 4a and 4b is determined, and finally an interval L1 between the first and second flat plate conductor portions 7a and 7b is determined.

  As described above, the phase shifter 1 can be formed in a small size, phase control can be performed well, and the mounting unit 5 is surface-mounted on the planar circuit board, so that the phase shifter 1 and the phase shifter 1 are planar. Connection with a transmission line formed on the circuit board 56 can be suitably performed. Further, since the inter-plate dielectric portion 9 functions as a support member for supporting the first and second flat plate conductor portions 7a and 7b, the first and second flat plate conductor portions 7a and 7b are formed into a thin film forming technique and a thick film printing technique. Alternatively, the phase shifter 1 can be manufactured by using a sheet-like ceramic technique, and the phase shifter 1 suitable for miniaturization can be realized in the manufacturing.

  In addition, since the electromagnetic field mode in the vicinity of the first strip conductor portion 11 approximates the LSE mode of the non-radiative dielectric line 2, the planar line 3 can be coupled with the LSE mode of the non-radiative dielectric line 2. The electromagnetic field can be smoothly transferred from one of the non-radiative dielectric line 2 and the planar line 3 to the other at the connection portion between the non-radiative dielectric line 2 and the planar line 3. Therefore, connection loss can be reduced, that is, transmission loss can be reduced. Further, since the coupling is performed in the LSE mode, a mode suppression circuit is unnecessary, and the size can be reduced as compared with the case of using the LSM mode.

  A second strip conductor portion 42 is provided so as to at least partially overlap the first strip conductor portion 11, and a high-frequency signal from the planar line 3 passes through a through hole 49 formed in the first ground conductor portion 13. Then, it propagates to a plane line that includes the mounting portion 5. By mounting the mounting portion 5 on the planar circuit board 56, the phase shifter 1 and the transmission line formed on the planar circuit board 56 can be suitably connected.

  Since the non-radiative dielectric line 2 can be configured to have a size in the width direction Y smaller than the size in the stacking direction Z, the phase shifter 1 has the mounting portion 5 in the stacking direction Z of the non-radiative dielectric line 2. Is provided, the mounting height when the phase shifter 1 is mounted on the planar circuit board 56 can be reduced as compared with the configuration in which the mounting portion 5 is provided in the width direction Y of the non-radiative dielectric line 2. In addition, it is possible to reduce the thickness of an electronic device configured using the phase shifter 1.

  Moreover, the phase shifter 1 can be formed by laminating | stacking a dielectric film and a conductor film on a board | substrate using well-known thin film formation methods, such as vacuum evaporation, sputtering, or CVD, and photolithography. The phase shifter 1 is formed by a process of laminating a dielectric film and a conductor film in the width direction Y for each part except for the first and second flat conductor portions 7a and 7b and the planar line flat conductor portion 24. Therefore, the manufacturing process is not complicated and easy to manufacture.

In another embodiment of the present invention, in the phase shifter 1 described above, the first and second electrodes 4 a and 4 b may be embedded in the dielectric line 6. The first and second electrodes 4 a and 4 b are provided so that an electric field can be applied to the dielectric line 6. The thicknesses of the first and second electrodes 4 a and 4 b are selected to be less than the skin thickness with respect to the electromagnetic wave propagating through the dielectric line 6. Thereby, when the first and second electrodes 4a and 4b are embedded in the dielectric line 6, loss due to the first and second electrodes 4a and 4b can be reduced. Even with such a configuration, the same effect as that of the phase shifter 1 can be achieved, and the first and second electrodes 4a and 4b are provided closer to each other than the phase shifter 1 described above. Therefore, the switch can be driven with a lower voltage. The volume resistivity of the first and second electrodes 4a and 4b in the present embodiment is selected to be 10 ⁻⁵ Ω · m or more, preferably 10 ⁻⁴ Ω · m or more.

  When the first and second electrodes 4a and 4b are embedded in the dielectric line 6, a plurality of the first and second electrodes 4a and 4b are separated from the first electrode 4a by a predetermined interval in the width direction Y. The second electrodes 4b may be formed so as to be alternately stacked in the width direction Y. In this case, each first electrode 4a is connected to the first flat plate conductor portion 7a, each second electrode 4b is connected to the second flat plate conductor portion 7b, and the first and second flat plate conductor portions 7a. , 7b are formed so as not to be electrically connected, and a voltage is applied between the first and second electrodes 4a, 4b via the first and second plate conductor portions 7a, 7b.

Increasing the number of the first electrodes 4a and the second electrodes 4b is preferable because the applied electric field strength increases, so that the switch can be operated at a lower voltage, but the first electrodes 4a and the second electrodes 4b are preferred. Increasing the number increases the loss. When the electrodes are stacked, the loss is determined by the total thickness of the electrodes. In the case where the resistivity of the electrode is 1 × 10 ⁻⁴ (Ω · m), it is practically preferable that the total thickness of the electrodes is 30 nm or less, and the resistivity of the electrode is 1 × 10 ⁻³ (Ω · m). In the case of m), it is practically preferable that the total thickness of the electrodes is 320 nm or less.

  In the phase shifter 1 of the above-described embodiment, the dielectric line 6 is made of a material whose dielectric constant changes. In still another embodiment of the present invention, the dielectric line 6 has a dielectric constant that changes. What is necessary is just the structure containing the change part which consists of a substance to do. The changing portion is preferably formed in a portion where the electric field strength is high, for example, in the central portion in the width direction Y and the thickness direction Z. Such a configuration is obtained when the phase shifter is manufactured with the same size according to the ratio of the changed portion in the dielectric line 6 and the region of the dielectric line 6 where the changed portion is formed. The phase change amount is determined, and the phase change amount is smaller than that in the case where the entire dielectric line 6 is made of a material whose dielectric constant changes, but a small phase shifter is provided as in the above-described embodiment. can do.

FIG. 7 is a schematic diagram showing the configuration of the high-frequency transmitter 60 according to the embodiment of the present invention. The high frequency transmitter 60 includes the phase shifter 1 of the embodiment shown in FIG. 1 described above, a high frequency oscillator 61, a transmission line 62, a transmission antenna 63, and a stub 64. The high-frequency oscillator 61 is a Gunn oscillator using a Gunn diode, an Impat oscillator using an Impat diode, or an MMIC (FET) (Field Effect Transistor).
Microwave Monolithic Integrated Circuit) Generates high frequency signals. The transmission line 62 is configured by a microstrip line, a strip line, or a coplanar line. A first end 62 a of the transmission line 62 in the transmission direction of the high frequency signal is connected to the high frequency oscillator 61, and a second end 62 b of the transmission line 62 in the transmission direction of the high frequency signal is connected to the transmitting antenna 63. The transmitting antenna 63 is realized by a patch antenna or a horn antenna. The transmission direction of the high frequency signal is the propagation direction of the electromagnetic wave.

  The phase shifter 1 is inserted into the transmission line 62 so that the high-frequency signal passes through the dielectric line 6. The stub 64 is realized by an open stub, for example, and functions as a characteristic adjustment circuit of the high-frequency oscillator 61. The stub 64 is provided on the transmission line 62 on at least one of the upstream side and the downstream side of the phase shifter 1 in the high-frequency signal transmission direction.

  More specifically, the transmission line 62 includes first and second transmission lines 68 and 69. A first end 68 a of the first transmission line 68 in the high-frequency signal transmission direction is connected to the high-frequency oscillator 61, and a second end 68 b of the first transmission line 68 in the high-frequency signal transmission direction is connected to the phase shifter 1. Connected to the first input / output terminal 2a. The first end 69a of the second transmission line 69 in the high-frequency signal transmission direction is connected to the second input / output terminal 2b of the phase shifter 1, and the second end of the second transmission line 69 in the high-frequency signal transmission direction. 69 b is connected to the transmitting antenna 63.

  The high-frequency signal generated by the high-frequency oscillator 61 passes through the first transmission line 68, the dielectric line 6 of the phase shifter 1, and the second transmission line 69, and is given to the transmission antenna 63. Is emitted as.

  In the high-frequency transmitter 60, a stub 64 is provided in the middle of the high-frequency oscillator 61 and the transmission antenna 63, and there is no connection between the connection portion of the high-frequency oscillator 61 to the transmission line 62 and the connection portion of the transmission antenna 63 to the transmission line 62. Alignment can be matched. As a result, reflection at the connection portion can be suppressed to be small, stable oscillation characteristics can be obtained, and insertion loss can be suppressed to be small, so that a high transmission output can be obtained. However, even if the stub 64 is provided, it cannot be uniformly matched due to, for example, variations in the shape of wires and / or bumps for connecting a high-frequency oscillator and variations in the wiring width of the transmission line 62. In the high frequency transmitter 60, the phase shifter 1 is inserted in the transmission line 62 so that the electromagnetic wave of the high frequency signal transmitted through the transmission line 62 passes through the dielectric line 6. The phase shift caused by the transmission line 62 due to variations in the shape of the wires and / or bumps and the wiring width of the transmission line, etc. can be individually adjusted to achieve matching and stable oscillation A high-frequency transmitter 60 having characteristics and a high transmission output can be realized because the insertion loss is suppressed to be small. Since the phase shifter 1 is small and can be operated at a low voltage as described above, the high-frequency transmitter 60 can be made small even if the phase shifter 1 is provided. It is possible to prevent the configuration for applying a voltage from becoming complicated.

  In the high-frequency transmitter 60, the phase shifter 1 is used, but any one of the phase shifters of the above-described embodiments may be used. Even if comprised in this way, the same effect can be achieved. In the high-frequency transmitter 60, the transmission line 62 may be realized by a coplanar line, a grounded coplanar line, a slot line, a waveguide, a dielectric waveguide, or the like, in addition to the microstrip line and the strip line. .

  FIG. 8 is a schematic diagram showing the configuration of the high-frequency receiver 70 according to the embodiment of the present invention. Components similar to those of the high-frequency transmitter 60 of the above-described embodiment shown in FIG. 7 are denoted by the same reference numerals, and description thereof may be omitted.

  The high frequency receiver 70 includes the phase shifter 1 of the above-described embodiment, the high frequency detector 71, the transmission line 62, the stub 64, and the receiving antenna 73. The high-frequency detector 71 is realized by, for example, a Schottky barrier diode detector, a video detector, a mixer MMIC, or the like.

  A first end 62 a of the transmission line 62 in the transmission direction of the high frequency signal is connected to the high frequency detector 71, and a second end 62 b of the transmission line 62 in the transmission direction of the high frequency signal is connected to the receiving antenna 73. . The receiving antenna 73 is realized by a patch antenna or a horn antenna.

  The phase shifter 1 is inserted into the transmission line 62 so that the high-frequency signal passes through the dielectric line 6. The stub 64 is provided on the transmission line 62 on at least one of the upstream side and the downstream side of the phase shifter 1 in the high-frequency signal transmission direction.

  When a radio wave arriving from the outside is captured by the reception antenna 73, the reception antenna 73 gives a high-frequency signal based on the radio wave to the transmission line 62, passes through the dielectric line 6 of the phase shifter 1, and passes to the high-frequency detector 71. A received high frequency signal is provided. The high frequency detector 71 detects a high frequency signal and detects information included in the high frequency signal.

  In the high frequency receiver 70, the high frequency signal captured by the receiving antenna 73 is transmitted to the transmission line 62 and detected by the high frequency detector 71. A stub 64 is provided in the middle of the receiving antenna 73 and the high-frequency detector 71, and mismatching at the connection portion of the high-frequency detector 71 to the transmission line 62 and the connection portion of the receiving antenna 73 to the transmission line 62 can be matched. It is like that. As a result, reflection at the connection portion can be suppressed to a small level, and stable detection characteristics can be obtained. Further, since the insertion loss is suppressed to a low level, a high detection output can be obtained. Even if the stub is provided, it cannot be uniformly matched due to, for example, variations in the shape of wires and / or bumps for connecting the high-frequency oscillator and variations in the wiring width of the transmission line 62. In the high frequency receiver 70, the phase shifter 1 is inserted into the transmission line 62 so that the electromagnetic wave of the high frequency signal transmitted through the transmission line 62 passes through the dielectric line 6. The phase shift caused by the transmission line 62 due to variations in the shape of wires and / or bumps for connecting 71 and variations in the wiring width of the transmission line can be individually adjusted to achieve matching. The high-frequency receiver 70 having a stable detection characteristic and a high detection output can be realized because the insertion loss is suppressed to a low level. Further, since the phase shifter 1 is small and can be operated at a low voltage as described above, the high-frequency receiver 70 can be made small even if the phase shifter 1 is provided. It is possible to prevent the configuration for applying a voltage from becoming complicated.

  In the high-frequency receiver 70, the phase shifter 1 is used, but any one of the phase shifters of the above-described embodiments may be used instead of the phase shifter. Even if comprised in this way, the same effect can be achieved. In the high-frequency receiver 70, the transmission line 62 may be realized by a coplanar line, a grounded coplanar line, a slot line, a waveguide, a dielectric waveguide, or the like, in addition to the microstrip line and the strip line. .

  FIG. 9 is a schematic diagram illustrating a configuration of a radar apparatus 90 including the high-frequency transmitter / receiver 80 according to the embodiment of this invention. In the radar apparatus 90, the same reference numerals are given to the same configurations as those of the high-frequency transmitter 60 and the high-frequency receiver 70 of the above-described embodiment shown in FIGS. 7 and 8, and the description thereof may be omitted. is there. The radar device 90 includes a high frequency transmitter / receiver 80 and a distance detector 91.

  The high frequency transmitter / receiver 80 includes the phase shifter 1 of the above-described embodiment, the high frequency oscillator 61, the first to fifth transmission lines 81, 82, 83, 84, 85, the branching device 86, and the branching filter 87. A transmission / reception antenna 88, a mixer 89, and a stub 64. The transmission / reception antenna 88 is realized by a patch antenna or a horn antenna. The first to fifth transmission lines 81, 82, 83, 84, 85 have the same configuration as the transmission line 62 described above.

  The first end 81 a of the first transmission line 81 in the high-frequency signal transmission direction is connected to the high-frequency oscillator 61, and the second end 81 b of the first transmission line 81 in the high-frequency signal transmission direction is connected to the branching device 86. Is done. The phase shifter 1 is inserted into the first transmission line 81 so that the high-frequency signal passes through the dielectric line 6. The stub 64 is provided in the first transmission line 81 on at least one of the upstream side and the downstream side of the phase shifter 1 in the high-frequency signal transmission direction.

  The branching device (switching device) 86 has first, second, and third terminals 86a, 86b, 86c, and a high-frequency signal applied to the first terminal 86a is selectively applied to the second terminal 86b and the third terminal 86c. Output to. The branching device 86 is realized by, for example, a high frequency switching element. The branching device 86 is supplied with a control signal from a control unit (not shown), and selectively connects the first terminal 86a and the second terminal 86b or the first terminal 86a and the third terminal 86c based on the control signal. The radar device 90 is realized by a pulse radar. The control unit connects the first terminal 86a and the second terminal 86b, outputs a pulsed high-frequency signal from the second terminal 86b, and then connects the first terminal 86a and the third terminal 86c, A signal is output from the third terminal 86c. A first end 82a of the second transmission line 82 in the transmission direction of the high-frequency signal is connected to the second terminal 86b. A first end 84a of the fourth transmission line 84 in the transmission direction of the high frequency signal is connected to the third terminal 86c.

  The duplexer 87 has fourth, fifth and sixth terminals 87a, 87b and 87c, outputs a high frequency signal applied to the fourth terminal 87a to the fifth terminal 87b, and provides a high frequency signal applied to the fifth terminal 87b. The signal is output to the sixth terminal 87c. A second end 82b of the second transmission line 82 in the transmission direction of the high frequency signal is connected to the fourth terminal 87a. A first end 83a of the third transmission line 83 in the transmission direction of the high frequency signal is connected to the fifth terminal 87b. The second end 83 b of the third transmission line 83 in the transmission direction of the high frequency signal is connected to the transmission / reception antenna 88.

  A first end 85a of the fifth transmission line 85 in the transmission direction of the high frequency signal is connected to the sixth terminal 87c. The second end portion 84 b of the fourth transmission line 84 in the high-frequency signal transmission direction and the second end portion 85 b of the fifth transmission line 85 in the high-frequency signal transmission direction are connected to the mixer 89. The duplexer 87 is realized by a hybrid circuit. The hybrid circuit is a directional coupler and is realized by a magic T, a hybrid ring, or a rat race.

  The high frequency signal generated by the high frequency oscillator 61 passes through the first transmission line 81 and the dielectric line 6 of the phase shifter 1, and the branching device 86, the second transmission line 82, the duplexer 87, and the third transmission line 82. Is transmitted to the transmission / reception antenna 88 and radiated as radio waves from the transmission / reception antenna 88. The high-frequency signal generated by the high-frequency oscillator 61 passes through the first transmission line 81 and the dielectric line 6 of the phase shifter 1 and is supplied as a local signal to the mixer 89 via the branching device 86 and the fourth transmission line 84. Given.

  When a radio wave coming from the outside is received by the transmission / reception antenna 88, the transmission / reception antenna 88 gives a high-frequency signal based on the radio wave to the third transmission line 83, and gives it to the mixer 89 via the duplexer 87 and the fifth transmission line 85. It is done.

  The mixer 89 mixes the high frequency signals given from the fourth and fifth transmission lines 84 and 85 and outputs an intermediate frequency signal. The intermediate frequency signal output from the mixer 89 is given to the distance detector 91.

  The distance detector 91 includes the high frequency detector 71 described above, and is based on the intermediate frequency signal obtained by receiving the radio wave (echo) radiated from the high frequency transmitter / receiver 80 and reflected by the measurement object. The distance to the measurement object is calculated. The distance detector 91 is realized by a microcomputer, for example.

  In the high-frequency transmitter / receiver 80, the phase shifter 1 is inserted into the first transmission line 81 so that a high-frequency signal passes through the dielectric line 6. By adjusting the phase of the high-frequency signal that changes undesirably due to the high-frequency transmitter / receiver, for example, it is possible to realize a high-frequency transmitter / receiver 80 having stable oscillation characteristics and high transmission output because the insertion loss is suppressed to a small level In addition, for example, a high-frequency transmitter / receiver 80 having a stable detection characteristic and a high detection output because the insertion loss is suppressed to a low level can be realized, and the reliability of the intermediate frequency signal generated by, for example, the mixer 89 can be realized. Can be improved. Since the phase shifter 1 is small and can be operated at a low voltage as described above, the high-frequency transmitter / receiver 80 can be formed small even if the phase shifter 1 is provided. It is possible to prevent the configuration for applying a voltage from becoming complicated.

  In the radar device 90, based on the intermediate frequency signal from the high frequency transmitter / receiver 80, the distance detector detects the distance from the high frequency transmitter / receiver 80 to the detection target, for example, the distance between the transmission / reception antenna 88 and the detection target. Therefore, the distance to the detection target can be accurately detected.

  The branching device 86 may be realized by a directional coupler. In this case, the high-frequency signal supplied to the first terminal 87a is branched and output to the second terminal 86b and the third terminal 86c. In this case, the power of the radio wave output from the transmission / reception antenna 88 is lower than that of the above-described configuration, but the control of the apparatus is simplified because it is not necessary to control the branching device 86.

  In the present embodiment, the phase shifter 1 is inserted into the first transmission line 81, but in still another embodiment of the present invention, the phase shifter 1 is at least one of the first to fifth transmission lines 81 to 85. Alternatively, a high-frequency signal may be inserted so as to pass through the dielectric line 6. Even if it is such a structure, the same effect can be achieved.

  Moreover, although the phase shifter 1 is used in the high frequency transmitter / receiver 80, any one of the phase shifters of the respective embodiments described above may be used. Even if comprised in this way, the same effect can be achieved.

  In still another embodiment of the present invention, the duplexer 87 may be realized by a circulator, and the same effect can be achieved even with such a configuration.

  FIG. 10 is a schematic diagram showing a configuration of the radar apparatus 100 including the array antenna apparatus 99 including the phase shifter 1 according to the embodiment of the present invention. In the embodiment of the present invention, the same components as those of the above-described embodiment are denoted by the same reference numerals, and the description thereof is omitted. The radar device 100 includes an array antenna device 99, a high frequency transmitter / receiver 109, and a distance detector 91.

  The array antenna device 99 includes an antenna array body 102 provided with an antenna element 105 with a phase shifter configured by an antenna element 101 and a phase shifter 1 added to the antenna element 101, and each phase shifter. And a transmission line 107 connected to the attached antenna element 105. In the embodiment of the present invention, the plurality of antenna elements 101 are arranged in a line with their radiation directions aligned. The antenna elements 101 are provided at equal intervals along the arrangement direction R.

  The antenna element 101 is realized by, for example, a slot antenna, a microstrip antenna, a horn antenna, a lot antenna, or a reflector antenna. In the embodiment of the present invention, the array antenna device 99 includes eight antenna elements 101 and eight phase shifters 1.

  The transmission line 107 is configured to include a branching device 103, and a high-frequency signal input from the input unit 104 is branched into a plurality by the branching device 103, and is given to each antenna element 105 with phase shifter. The transmission line 107 is realized by a microstrip line, a strip line, a coplanar line, a coplanar line with a ground, a slot line, a waveguide, a dielectric waveguide, or the like.

  The high frequency transmitter / receiver 109 may be configured by the high frequency transmitter / receiver 80 of each of the above-described embodiments, or the high frequency transmitter / receiver 80 may not include a phase shifter. And a conventional high-frequency transceiver that receives a high-frequency signal captured by the array antenna device 99 may be used.

  The phase shifter 1 is provided between the transmission line 107 and the antenna element 101 of each phase shifter-equipped antenna element 105. A high-frequency signal propagating through the transmission line 107 passes through the dielectric line 6 of the phase shifter 1 and is given to the antenna element 101. Each phase shifter 1 adjusts the phase of the radio wave radiated from each antenna element by shifting the phase of the high-frequency signal, and the equiphase surface is moved from the first direction R1 of the arrangement direction R as shown in FIG. The direction of the radiation beam 106 is changed from the front by the first direction R1 or the first direction R1 of the arrangement direction R of the antenna elements 101 by shifting the phase of the radio wave radiated from the adjacent antenna elements 101 by Δφ toward the second direction R2. It can be tilted in the two directions R2 by an angle θ.

  Since the phase shifter 1 is small and can be operated at a low voltage, the array antenna device 99 is not increased in size. By providing the phase shifter 1, the array antenna device 99 can change the direction of the radiation beam, thereby changing the direction of the radiation beam without mechanically operating the antenna element 101. , Can improve convenience.

  Further, since the radar apparatus 100 can be easily changed without increasing the size of the radar apparatus 100, a highly convenient radar apparatus can be realized.

  Although the phase shifter 1 is used in the radar apparatus 100, any one of the phase shifters of the above-described embodiments may be used instead of the phase shifter 1. Even if comprised in this way, the same effect can be achieved.

  In the radar apparatus 100, the phase shifter 1 may be configured in place of any one of the phase shifters of the above-described embodiments.

  The high-frequency switch according to the embodiment of the present invention has the same configuration as any of the phase shifters of the above-described embodiments. Hereinafter, the “high frequency switch” is simply referred to as “switch”. In such a switch, the cutoff frequency in the non-radiative dielectric line 2 can be changed by applying a voltage to the first and second electrodes 4a and 4b.

  The voltage applying means 23 applies an AC voltage or a DC voltage having a frequency lower than the frequency of the propagating electromagnetic wave to the first and second electrodes 4a and 4b. The voltage application means 23 applies a voltage to the first and second electrodes 4a and 4b, so that the dielectric constant of the dielectric line 6 is reduced, thereby increasing the cutoff frequency of the switch. When the voltage applying means 23 does not apply a voltage to the first and second electrodes 4a and 4b, the switch is configured such that the cutoff frequency of the switch is lower than the frequency of the electromagnetic wave to be propagated (use frequency). The voltage applying unit 23 can apply a voltage to the first and second electrodes 4a and 4b so that the cut-off frequency of the switch is equal to or higher than the use frequency. Therefore, in the switch, the voltage application means 23 causes the propagation state where the cutoff frequency is lower than the frequency of the electromagnetic wave propagating through the dielectric line 6, and the cutoff frequency becomes higher than the frequency of the electromagnetic wave propagating through the dielectric line 6. The cut-off state can be switched. In the embodiment of the present invention, the operating frequency is constant, and therefore ON / OFF operation is possible by the switching described above.

  In the switch having the above-described configuration, the propagation state in which the cutoff frequency in the non-radiative dielectric line 2 is lower than the frequency of the electromagnetic wave propagating through the dielectric line 6 according to the electric field applied to the dielectric line 6, and the electromagnetic wave Since it is possible to switch between cutoff states higher than the frequency of the first and second electrodes 24a and 24b, the propagation state and the cutoff state can be easily switched by changing the voltage applied to the first and second electrodes 24a and 24b. it can. When the switching mode is in the OFF state, the cut-off state is entered, so that an essentially high ON / OFF ratio can be obtained. In addition, since there is no mechanical drive part, it is possible to realize a highly reliable high-frequency switch having excellent durability. Moreover, the said structure can implement | achieve the switch which can change a cutoff frequency with a low voltage. In addition, since the high-frequency signal in the LSE mode can be satisfactorily taken out to the planar line by the connection structure, it is possible to realize a high-frequency switch that has good mountability on the planar circuit board.

  Even if the voltage applied to the first and second electrodes 24 a and 24 b is reduced in order to apply an electric field to the dielectric line 6, an electric field having a large electric field strength is applied to the changed portion, and the line length of the dielectric line 6 is increased. Even if is short, an electric field having a large electric field strength is applied to the changing portion, so that a switch that is small and can be operated at a low voltage can be realized. In addition, since there is no mechanical drive part, it is possible to realize a highly reliable high-frequency switch having excellent durability.

  The attenuator according to one embodiment of the present invention has the same configuration as any of the phase shifters according to the respective embodiments described above. In such an attenuator, by applying a voltage to the first and second electrodes 4a and 4b, the cut-off frequency in the non-radiative dielectric line 2 can be changed to change the propagation characteristics. By changing the propagation characteristics in the non-radiative dielectric line 2 according to the electric field applied to the dielectric line 6, the high frequency signal can be attenuated, and the connection structure allows the LSE mode high frequency signal to be attenuated. Since it can be satisfactorily taken out to a planar line, an attenuator with good mountability on a planar circuit board can be realized. As with the phase shifter described above, the attenuator preferably has 1.03 <f / fc <1.5 so that 1.03 <f / fc <1.5 when the cut-off frequency is fc and the use frequency is f. It is formed so that f / fc <1.2. Further, even if the voltage applied to the first and second electrodes 4a and 4b is reduced, an electric field having a large electric field strength is applied to the changed portion, and the attenuation characteristic near the cutoff frequency is used. Even if the length of the transmission line is short, the electromagnetic wave can be sufficiently attenuated even if the transmission line length is short, so that an electric field having a large electric field strength is given to the changing portion, so that it can be operated with a small size and a low voltage. An attenuator can be realized. Moreover, since there is no mechanical drive part, a highly reliable attenuator with excellent durability can be realized.

  FIG. 11 is a schematic diagram showing a configuration of a high-frequency transmitter 160 according to another embodiment of the present invention. The high-frequency transmitter 160 has a configuration in which the high-frequency switch 161 is provided in place of the phase shifter 1 of the high-frequency transmitter 60 in FIG. 7 described above and the stub 64 is omitted. The description is omitted. The switch 161 has the same configuration as any of the phase shifters of the above-described embodiments.

  The switch 161 is inserted into the transmission line 62 and transmits a high-frequency signal transmitted to the transmission line 62 by setting the propagation state, and blocks the high-frequency signal transmitted to the transmission line 62 by setting the cutoff state.

  When the switch 161 is in a propagation state, the high-frequency signal generated by the high-frequency oscillator 61 is transmitted to the transmission line 62, passes through the dielectric line 6 of the switch 161, is given to the transmitting antenna 63, and is radiated as a radio wave. . When the switch 161 is in the cut-off state, the high frequency signal generated by the high frequency oscillator 61 is not transmitted to the transmitting antenna 63 because it does not pass through the switch 161. By switching the propagation state and cut-off state of the switch 161, a pulse signal wave can be radiated from the transmitting antenna 63. A high-frequency transmitter 160 with high reliability can be realized by using a high-reliability high-frequency switch with excellent durability and high ON / OFF ratio. The voltage applying means 23 applies a voltage to the switch 161 on the basis of the predetermined information, and turns on / off the switch 161, whereby a radio wave corresponding to the predetermined information can be radiated from the transmitting antenna 63. .

  In the high-frequency transmitter 160, the transmission line 62 may be realized by a coplanar line, a grounded coplanar line, a slot line, a waveguide, a dielectric waveguide, or the like in addition to the microstrip line and the strip line.

  FIG. 12 is a schematic diagram showing a configuration of a radar apparatus 190 including a high-frequency transceiver 180 according to another embodiment of the present invention. The high frequency transmitter / receiver 180 has a configuration in which a high frequency switch 161 is provided instead of the phase shifter 1 of the high frequency transmitter / receiver 80 of FIG. 9 described above. Description is omitted.

  By setting the switch 161 inserted in the first transmission line 81 in a propagation state, the high frequency signal generated by the high frequency oscillator 61 is transmitted to the first transmission line 81 and given to the first terminal 86a of the branching device 86, The signal is supplied from the second terminal 86 b of the branching device 86 to the second transmission line 82, supplied to the fourth terminal 87 a of the branching filter 87, and supplied from the fifth terminal 87 b of the branching device 87 to the third transmission line 83. And radiated from the transmitting / receiving antenna 88. When the switch 161 inserted into the first transmission line 81 is cut off, the high-frequency signal generated by the high-frequency oscillator 61 is not transmitted through the switch 161 and thus is cut off and is not radiated from the transmission / reception antenna 88. By switching the propagation state and cut-off state of the switch 161, a pulse signal wave can be radiated from the transmission / reception antenna 88. A high-reliability high-frequency transmitter / receiver can be realized by using a highly reliable switch 161 that can obtain a large ON / OFF ratio and has excellent durability. In the present embodiment, the switch 161 is inserted into the first transmission line 81. However, in still another embodiment of the present invention, the switch 161 is connected to at least one of the first to third transmission lines 81 to 83. , May be inserted. Even in such a configuration, all the switches 161 inserted in at least one of the first to third transmission lines 81 to 83 are set in a propagation state, and at least the first to third transmission lines 81 to 83 are set. By setting any one of the switches 161 inserted in any one of the cut-off states to a cut-off state, it is possible to radiate a pulse signal wave from the transmission / reception antenna 88 and achieve the same effect as the above-described radar device. Can do.

  In a radar apparatus according to still another embodiment of the present invention, the branching device 86 constituting the high frequency transmitter / receiver 80 may be configured by two switches 161 in the radar apparatus according to each of the above embodiments.

  FIG. 13 is a schematic diagram illustrating the configuration of the branching device 86 configured by the switch 161. The two switches 161 are referred to as a first switch 161A and a second switch 161B. The first switch 161A transmits the high-frequency signal between the first terminal 86a and the second terminal 86b by setting the propagation state, and the high-frequency signal between the first terminal 86a and the second terminal 86b by setting the cutoff state. Shut off. The second switch 161B transmits the high-frequency signal between the first terminal 86a and the third terminal 86c by setting the propagation state, and the high-frequency signal between the first terminal 86a and the third terminal 86c by setting the cutoff state. Shut off. The first ends of the first and second switches 161A and 161B in the electromagnetic wave propagation direction are connected to serve as the first terminal 86a. The second end of the first switch 161 </ b> A in the waveguide direction X of the electromagnetic wave is a second terminal 86. The second end of the second switch 161 </ b> B in the waveguide direction X of the electromagnetic wave is a third terminal 86.

  A control signal is given to the first and second switches 161A and 161B from a control unit (not shown), and when the first switch 161A is in a propagation state based on the control signal, the second switch 161B is cut off, When the first switch 161A is in the cut-off state, the high frequency signal input from the first terminal 86a is selectively output from the second and third terminals 86b and 86c by setting the second switch 161B to the propagation state. be able to. The radar apparatus is realized by a pulse radar. The control unit controls the first and second switches 161A and 161B, connects the first terminal 86a and the second terminal 86b, and outputs a pulsed high-frequency signal from the second terminal 86b. The first and second switches 161A and 161B are controlled to connect the first terminal 86a and the third terminal 86c, so that a high frequency signal is output from the third terminal 86c. A high-reliability high-frequency transmitter / receiver can be realized by forming the branching device 86 using the highly reliable switch 161 having excellent durability while being able to obtain a large ON / OFF ratio.

  In the radar apparatus according to still another embodiment of the present invention, the duplexer 87 constituting the high frequency transmitter / receiver 80 may be configured by two switches 161 in the radar apparatus according to each of the above embodiments.

  FIG. 14 is a schematic diagram showing the configuration of the duplexer 87 configured by the switch 161. The duplexer 87 includes two switches 161. The two switches 161 are referred to as a third switch 161C and a fourth switch 161D. The third switch 161C transmits a high-frequency signal between the fourth terminal 87a and the fifth terminal 87b by being in a propagation state, and is a high-frequency signal between the fourth terminal 87a and the fifth terminal 87b by being in a cutoff state. Shut off. The fourth switch 161D transmits the high-frequency signal between the fifth terminal 87b and the sixth terminal 87c by setting the propagation state, and the high-frequency signal between the fifth terminal 87b and the sixth terminal 87c by setting the cutoff state. Shut off. The first end of the third switch 161C in the propagation direction of the electromagnetic wave is referred to as a fourth terminal 87a. In addition, the second ends of the third and fourth switches 161C and 161D in the propagation direction of the electromagnetic wave are connected in common to form a fifth terminal 87b. The first end of the fourth switch 161D in the propagation direction of the electromagnetic wave is referred to as a sixth terminal 87c.

  The third and fourth switches 161C and 161D receive a control signal from a control unit (not shown), and when the third switch 161C is in the propagation state based on the control signal, the fourth switch 161D is cut off, When the third switch 161C is in the cutoff state, the high frequency signal input from the fourth terminal 87a is output from the fifth terminal 87b and input from the fifth terminal 87b by setting the fourth switch 161D to the propagation state. A high frequency signal can be output from the sixth terminal 87c. The control unit controls the third and fourth switches 161C and 161D, connects the fourth terminal 87a and the fifth terminal 87b, transmits a pulsed high-frequency signal to the transmitting and receiving antenna, By controlling the fourth switches 161C and 161D, the fifth terminal 87b and the sixth terminal 87c are connected, and the high-frequency signal captured by the transmission / reception antenna is output from the sixth terminal 87c. The control unit sets the first and third switches 161A and 161C to be in a propagation state and the second and fourth switches 161B and 161D to be in a cutoff state, or the first and third switches 161A and 161C are cut. The first to fourth switches 161A to 161D are controlled so that the second and fourth switches 161B and 161D are in the propagation state. A high-frequency transmitter / receiver with high reliability can be realized by configuring the duplexer 87 using the highly reliable switch 161 having excellent durability while being able to obtain a large ON / OFF ratio.

In each of the embodiments described above, the changing portion may be a piezoelectric element whose size changes according to the applied electric field. Depending on the applied voltage, the phase of the electromagnetic wave propagating through the dielectric part including the changing part is changed by changing the dimension of the piezoelectric element in the electromagnetic wave direction, that is, by changing the thickness of the piezoelectric element in the propagation direction. It is possible to achieve the same effects as those of the above-described embodiment. The piezoelectric element is made of, for example, quartz, zinc oxide, aluminum nitride, Pb (Zr, Ti) O ₃ , BaTiO ₃ , LiNbO _3, or SbSI. The phase shifter, switch, and attenuator of each of the above embodiments are dielectric waveguide devices.

  Note that the present invention is not limited to the above-described embodiments and examples, and various modifications may be made without departing from the scope of the present invention.

It is sectional drawing which shows typically the structure of the phase shifter 1 of one Embodiment of this invention. It is sectional drawing which shows typically the structure of the phase shifter 1 of one Embodiment of this invention. It is sectional drawing which shows typically the structure of the phase shifter 1 of one Embodiment of this invention. It is sectional drawing seen from the cut surface line IV-IV of FIG. 3 is a plan view of the phase shifter 1. FIG. It is a top view of the planar circuit board 56 with which the phase shifter 1 is mounted. It is a schematic diagram which shows the structure of the high frequency transmitter 60 of one Embodiment of this invention. It is a schematic diagram which shows the structure of the high frequency receiver 70 of one Embodiment of this invention. It is a schematic diagram which shows the structure of the radar apparatus 90 provided with the high frequency transmitter / receiver 80 of one Embodiment of this invention. It is a schematic diagram which shows the structure of the radar apparatus 100 containing the array antenna apparatus 99 provided with the phase shifter 1 of embodiment of this invention. It is a schematic diagram which shows the structure of the high frequency transmitter 160 of other embodiment of this invention. It is a schematic diagram which shows the structure of the radar apparatus 190 provided with the high frequency transmitter-receiver 180 of other embodiment of this invention. 3 is a schematic diagram showing a configuration of a branching device 86 constituted by a switch 161. FIG. 4 is a schematic diagram showing a configuration of a duplexer 87 configured by a switch 161. FIG.

Explanation of symbols

DESCRIPTION OF SYMBOLS 1 Phase shifter 2 Nonradiative dielectric line 3 Planar line 4a, 4b Electrode 5 Mounting part 6 Dielectric line 7a, 7b Flat plate conductor part 9 Interplate dielectric part (3rd dielectric part)
11 First Strip Conductor 12 Strip Dielectric Part (Dielectric Part)
13 First Ground Conductor Part 41 Mounting Dielectric Part (Second Dielectric Part)
42 Second strip conductor 49 Through-hole 60, 160 High-frequency transmitter 61 High-frequency oscillator 62, 81, 82, 83, 84, 85 Transmission line 63 Transmitting antenna 64 Stub 70 High-frequency receiver 71 High-frequency detector 73 Receiving antenna 80 , 180 High-frequency transmitter / receiver 86 Branching device 87 Demultiplexer 88 Transmitting / receiving antenna 89 Mixer 86a First terminal 86b Second terminal 86c Third terminal 87a Fourth terminal 87b Fifth terminal 87c Sixth terminal 90, 190 Radar device 91 Distance detection Device 100 Array antenna device 101 Antenna element 161 Switch

Claims (17)

A dielectric line in which electromagnetic waves propagate including a change part in which at least one of a dielectric constant and a dimension changes according to an applied electric field, and a non-radiative dielectric line including a pair of flat conductor parts sandwiching the dielectric line When,
Provided along the dielectric line, and in a state perpendicular to the direction in which the dielectric line and the pair of flat plate conductors overlap, with a gap narrower than the gap between the pair of flat plate conductors. A pair of electrodes for sandwiching the dielectric line and applying an electric field to the changing portion;
The first strip conductor part, the dielectric part provided with the first strip conductor part, and the gap between the first strip conductor part in a direction perpendicular to the direction in which the dielectric line and the pair of flat plate conductor parts overlap. A first grounding conductor portion provided on the dielectric portion along the first strip conductor portion, and the non-contact so that the dielectric line and the first strip conductor portion are in contact with each other. A plane line coupled to the non-radiative dielectric line by abutting the end face in the electromagnetic wave propagation direction of the radioactive dielectric line with the end face in the electromagnetic wave propagation direction of the dielectric part,
A second ground conductor portion connected to the first ground conductor portion and provided at an end of the non-radiative dielectric line in a direction perpendicular to a direction in which the dielectric line and the pair of flat plate conductor portions overlap;
A second dielectric portion formed on a surface portion of the first ground conductor portion opposite to the first strip conductor portion and a surface portion of the second ground conductor portion opposite to the non-radiative dielectric line; ,
A second strip conductor portion provided at least partially overlapping the first strip conductor portion on a surface portion of the two dielectric portions opposite to the nonradiative dielectric line and the dielectric line;
Including at least a first electrode connection portion and a second electrode connection portion provided on the surface portion of the second dielectric portion and individually connected to a pair of the electrodes;
The dielectric waveguide device according to claim 1, wherein a through hole is formed in the first ground conductor portion between the first and second strip conductor portions.   The dielectric waveguide device according to claim 1, wherein the pair of electrodes are formed thinner than a skin thickness with respect to an electromagnetic wave propagating through the dielectric line.   A third dielectric portion provided between the pair of plate conductor portions, sandwiching the dielectric line in a direction in which the pair of plate conductor portions overlap, and having a dielectric constant lower than a dielectric constant of the dielectric line; 3. The dielectric waveguide device according to claim 2, further comprising: A dielectric waveguide device according to any one of claims 1 to 3,
A phase shifter that changes a phase of an electromagnetic wave propagating through a transmission line by changing at least one of a dielectric constant and a dimension of the changing portion according to an electric field applied to the changing portion. A dielectric waveguide device according to any one of claims 1 to 3,
The cutoff frequency in the transmission line becomes lower than the frequency of the electromagnetic wave propagating through the transmission line by changing at least one of the dielectric constant and dimension of the changing part according to the electric field applied to the changing part. A high-frequency switch characterized by being able to switch between a propagation state and an increased cutoff state. A dielectric waveguide device according to any one of claims 1 to 3,
An attenuator, wherein an electromagnetic wave propagating through a transmission line is attenuated by changing at least one of a dielectric constant and a dimension of the change part according to an electric field applied to the change part. A high-frequency oscillator that generates a high-frequency signal;
A transmission line connected to the high-frequency oscillator and transmitting a high-frequency signal from the high-frequency oscillator;
An antenna connected to the transmission line and emitting a high-frequency signal;
The phase shifter according to claim 5, wherein the phase shifter is inserted into the transmission line so that a high-frequency signal passes through the dielectric line.
A high frequency transmitter comprising: a stub provided on at least one of the upstream side and the downstream side of the phase shifter in a transmission direction of a high frequency signal. An antenna that captures high-frequency signals;
A transmission line connected to the antenna and transmitting a high-frequency signal captured by the antenna;
A high-frequency detector connected to the transmission line and detecting a high-frequency signal transmitted to the transmission line;
The phase shifter according to claim 5, wherein the phase shifter is inserted into the transmission line so that a high-frequency signal passes through the dielectric line.
A high frequency receiver comprising: a stub provided on at least one of the upstream side and the downstream side of the phase shifter in the transmission direction of the high frequency signal. A high-frequency oscillator that generates a high-frequency signal;
A first transmission line that is connected to the high-frequency oscillator and transmits a high-frequency signal; and first, second, and third terminals; the first terminal is connected to the first transmission line and applied to the first terminal A branching device that selectively outputs a high-frequency signal to be output to the second terminal or the third terminal;
A second transmission line connected to the second terminal and transmitting a high-frequency signal applied from the second terminal;
A fourth, fifth, and sixth terminal that outputs a high-frequency signal applied to the fourth terminal via the second transmission line to the fifth terminal, and a high-frequency signal applied to the fifth terminal; A duplexer that outputs to the sixth terminal;
A third transmission line connected to the fifth terminal for transmitting a high-frequency signal output from the fifth terminal and transmitting the high-frequency signal to the fifth terminal;
An antenna connected to the third transmission line for radiating and capturing high-frequency signals;
A fourth transmission line connected to the third terminal and transmitting a high-frequency signal output from the third terminal;
A fifth transmission line connected to the sixth terminal for transmitting a high-frequency signal output from the sixth terminal;
A mixer that is connected to the fourth and fifth transmission lines, mixes high-frequency signals given from the fourth and fifth transmission lines, and outputs an intermediate frequency signal;
The phase shifter according to claim 4, further comprising: a phase shifter inserted into at least one of the first to fifth transmission lines so that a high-frequency signal passes through the dielectric line. Transceiver. A high-frequency oscillator that generates a high-frequency signal;
A high-frequency transmission line connected to the high-frequency oscillator and transmitting a high-frequency signal from the high-frequency oscillator;
An antenna connected to the high-frequency transmission line and emitting a high-frequency signal;
The high-frequency signal inserted into the high-frequency transmission line is transmitted through the high-frequency transmission line by setting the propagation state, and the high-frequency signal transmitted to the high-frequency transmission line is blocked by setting the cutoff state. A high-frequency transmitter comprising the high-frequency switch according to claim 5. A high-frequency oscillator that generates a high-frequency signal;
A first high-frequency transmission line that is connected to the high-frequency oscillator and transmits a high-frequency signal; and first, second, and third terminals; the first terminal is connected to the first high-frequency transmission line; A branching device that selectively outputs a high-frequency signal given to the second terminal or the third terminal;
A second high frequency transmission line connected to the second terminal and transmitting a high frequency signal applied from the second terminal;
A high-frequency signal having fourth, fifth, and sixth terminals, outputting a high-frequency signal supplied to the fourth terminal via the second high-frequency transmission line to the fifth terminal, and supplied to the fifth terminal A duplexer that outputs to the sixth terminal;
A third high-frequency transmission line that is connected to the fifth terminal, transmits a high-frequency signal output from the fifth terminal, and transmits a high-frequency signal to the fifth terminal;
An antenna connected to the third high-frequency transmission line for radiating and capturing high-frequency signals;
A fourth high-frequency transmission line connected to the third terminal and transmitting a high-frequency signal output from the third terminal;
A fifth high-frequency transmission line connected to the sixth terminal and transmitting a high-frequency signal output from the sixth terminal;
A mixer that is connected to the fourth and fifth high-frequency transmission lines, mixes high-frequency signals given from the fourth and fifth high-frequency transmission lines, and outputs an intermediate frequency signal;
The branching device includes two high-frequency switches according to claim 5, wherein the first high-frequency switch transmits the high-frequency signal between the first terminal and the second terminal by being in the propagation state, and the cut The high-frequency signal is cut off between the first terminal and the second terminal by turning off, and the high-frequency signal is cut off between the first terminal and the third terminal by turning on the propagation state. A high-frequency transmitter / receiver that cuts off a high-frequency signal between the first terminal and the third terminal by being transmitted and in the cut-off state. A high-frequency oscillator that generates a high-frequency signal;
A first high-frequency transmission line that is connected to the high-frequency oscillator and transmits a high-frequency signal; and first, second, and third terminals; the first terminal is connected to the first high-frequency transmission line; A branching device that selectively outputs a high-frequency signal given to the second terminal or the third terminal;
A second high frequency transmission line connected to the second terminal and transmitting a high frequency signal applied from the second terminal;
A high-frequency signal having fourth, fifth, and sixth terminals, outputting a high-frequency signal supplied to the fourth terminal via the second high-frequency transmission line to the fifth terminal, and supplied to the fifth terminal A duplexer that outputs to the sixth terminal;
A third high-frequency transmission line that is connected to the fifth terminal, transmits a high-frequency signal output from the fifth terminal, and transmits a high-frequency signal to the fifth terminal;
An antenna connected to the third high-frequency transmission line for radiating and capturing high-frequency signals;
A fourth high-frequency transmission line connected to the third terminal and transmitting a high-frequency signal output from the third terminal;
A fifth high-frequency transmission line connected to the sixth terminal and transmitting a high-frequency signal output from the sixth terminal;
A mixer that is connected to the fourth and fifth high-frequency transmission lines, mixes high-frequency signals given from the fourth and fifth high-frequency transmission lines, and outputs an intermediate frequency signal;
The duplexer includes two high-frequency switches according to claim 5, wherein the third high-frequency switch transmits a high-frequency signal between the fourth terminal and the fifth terminal by being in the propagation state, and The high-frequency signal is cut off between the fourth terminal and the fifth terminal by setting the cut-off state, and the fourth high-frequency switch is set between the fifth terminal and the sixth terminal by setting the propagation state. And a high-frequency transmitter-receiver that cuts off a high-frequency signal between the fifth terminal and the sixth terminal by setting the cutoff state. A high-frequency oscillator that generates a high-frequency signal;
A first high-frequency transmission line that is connected to the high-frequency oscillator and transmits a high-frequency signal; and first, second, and third terminals; the first terminal is connected to the first high-frequency transmission line; A branching device that selectively outputs a high-frequency signal given to the second terminal or the third terminal;
A second high frequency transmission line connected to the second terminal and transmitting a high frequency signal applied from the second terminal;
A high-frequency signal having fourth, fifth, and sixth terminals, outputting a high-frequency signal supplied to the fourth terminal via the second high-frequency transmission line to the fifth terminal, and supplied to the fifth terminal A duplexer that outputs to the sixth terminal;
A third high-frequency transmission line that is connected to the fifth terminal, transmits a high-frequency signal output from the fifth terminal, and transmits a high-frequency signal to the fifth terminal;
An antenna connected to the third high-frequency transmission line for radiating and capturing high-frequency signals;
A fourth high-frequency transmission line connected to the third terminal and transmitting a high-frequency signal output from the third terminal;
A fifth high-frequency transmission line connected to the sixth terminal and transmitting a high-frequency signal output from the sixth terminal;
A mixer that is connected to the fourth and fifth high-frequency transmission lines, mixes high-frequency signals given from the fourth and fifth high-frequency transmission lines, and outputs an intermediate frequency signal;
The high-frequency switch according to claim 5, wherein the high-frequency switch is inserted into at least one of the first to third transmission lines so that a high-frequency signal passes through the dielectric line when in the propagation state. A high frequency transceiver characterized by.   The high-frequency transceiver according to claim 9, 11 or 13, wherein the duplexer is formed by a hybrid circuit or a circulator. A high-frequency transceiver according to any one of claims 9, 11 to 14, and
A radar apparatus comprising: a distance detector that detects a distance from the high frequency transmitter / receiver to an object to be detected based on an intermediate frequency signal from the high frequency transmitter / receiver.   An array antenna apparatus comprising a plurality of antennas with phase shifters each having an antenna element and the phase shifter according to claim 4. An array antenna apparatus according to claim 16,
A radar apparatus comprising: a high-frequency transmitter / receiver connected to the array antenna apparatus, for applying a high-frequency signal to the array antenna apparatus, and for receiving a high-frequency signal captured by the array antenna apparatus.

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