JP2007318999A - Switching power supply - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To attain a high efficiency, economical and compact device, in a switching power supply provided with a function of step-up chopper operation and a function of half-bridge circuit operation. <P>SOLUTION: The switching power supply has input terminals 2a, 2b for receiving DC voltage; an output transformer T of a core construction, having a first primary winding wire N1; a second primary winding wire N2; and secondary winding wires Na, Nb; a series circuit of the primary winding wire N2, connected between the concerned input terminals and a first switching element Q1; a series circuit of the concerned primary winding wire N1 connected between terminals of the concerned switching element Q1 and a capacitor C1; a series circuit of a second switching element Q2 connected between terminals of the concerned primary winding wire N1 and a capacitor C2; a rectifier circuit, constituted by synchronous rectifying MOSFETQ3, Q4 to be connected to the concerned secondary winding wire of the concerned output transformer; a filter circuit connected to the concerned rectifier circuit, an output terminal; and a control circuit 18 detecting output voltage of the output terminal to alternately on/off control the concerned first switching element and the concerned second switching element. <P>COPYRIGHT: (C)2008,JPO&INPIT

Description

  The present invention relates to high efficiency and small size economy of a switching power supply device.

  In a synchronous rectification type switching power supply using a MOSFET, the problem when trying to achieve high efficiency is that the higher the voltage between the source and drain of the MOSFET, the higher the on-resistance and the greater the power loss, so the efficiency of the power supply There is a problem that decreases. Further, the MOSFET constituting the synchronous rectifier circuit cannot be driven over the entire period of the off period of the primary side switch, and switching loss occurs during this period. Further, since a choke coil constituting an input choke coil and an output filter is required, the mounting area is increased.

  The present applicant has devised a switching power supply device that has improved the above-mentioned problems (Japanese Patent Laid-Open No. 11-262263). FIG. 5 shows an example of this circuit. In FIG. 5, VIN is an input power source, 2a and 2b are input terminals, L1 is a choke coil, Q1 and Q2 are switch elements (MOSFETs), C1 and C2 are capacitors, and T and N1, Na and Nb are output transformers and their Primary winding and secondary winding portions, Q3 and Q4 are synchronous rectification MOSFETs, L and Cout are choke coils and capacitors constituting an output filter, 18a and 18b are output terminals, and 18 is a control circuit.

  In this circuit, the primary side switch elements Q1 and Q2 are alternately turned on and off (the other switch element is turned off when one switch element is turned on) and gated to the synchronous rectification MOSFETs Q3 and Q4 via the output transformer T. A signal is supplied and power is supplied to the load 17 via the choke coil L. Further, the output voltage Vout is controlled at a constant voltage by detecting the voltage of the output terminal and changing the on / off ratio of the primary side switching elements Q1, Q2 through the control circuit 18.

  Incidentally, the circuit as a step-up chopper circuit sends power from the input power source VIN to the series circuit of the capacitors C1 and C2, and simultaneously supplies power to the load from the series circuit of the capacitors C1 and C2 by the operation of the half bridge circuit.

  Since this conventional circuit can always drive the synchronous rectification MOSFETs Q3 and Q4 except for a short off period, a synchronous rectification MOSFET having a low withstand voltage and a low on-resistance can be used, and an output filter can be made small, so that high efficiency switching is possible. A power supply device can be provided.

  The present invention provides a switching power supply device that eliminates the need for a choke coil in an input section and an output section, and that achieves high efficiency and miniaturization.

  In order to achieve the above object, the switching power supply device of the present invention includes a first primary winding and a second primary winding wound around the center legs, and a winding around one of the center legs. The core of the output transformer provided with the rotated secondary winding and the secondary winding wound around the other center leg is used. An input terminal for receiving a DC voltage; an output transformer having a first primary winding, a second primary winding and a secondary winding; and the second primary winding connected between the input terminals and the first A series circuit of the switch element, a series circuit of the first primary winding and the capacitor of the output transformer connected between the terminals of the switch element, and the first primary winding of the output transformer. A series circuit of a switch element and a capacitor connected between the terminals, a rectifier circuit configured by a synchronous rectification MOSFET and connected to the secondary winding of the output transformer, a capacitor connected to the rectifier circuit, and an output terminal And a control circuit that detects the output voltage of the output terminal and alternately turns on and off the first switch element and the second switch element, and eliminates the need for an input choke coil and an output choke coil And features.

  FIG. 1 is a circuit diagram of one embodiment of the present invention. The present embodiment according to the present invention is greatly characterized in that the output transformer T is provided with the second primary winding N2, and the exciting inductances of the primary windings N1 and N2 are used.

  In FIG. 1, VIN is an input power source, 2a and 2b are input terminals, Q1 and Q2 are first and second switch elements, respectively, and C1 and C2 are first and second capacitors, respectively. T, N1, N2, Na and Nb are the transformer and its first primary winding part, the second primary winding part, the first secondary winding part and the second secondary winding part, respectively. , Q3 and Q4 are a first synchronous rectification MOSFET and a second synchronous rectification MOSFET, respectively, Cout is a third capacitor for output smoothing, 18a and 18b are output terminals, and 17 is a load. , 18 is a control circuit.

  Next, the circuit operation of FIG. 1 will be described with reference to FIG. 2 which is a voltage and current waveform of each part. In FIG. 2, T31 is an operation cycle of the switch element, Ton31 is a period in which the first switch element Q1 is on, Ton32 is a period in which the second switch element is on, and Toff31 and Toff32 are both the first and second switch elements. Is a period for preventing the Toff31 and Toff32 switch elements Q1 and Q2 from being simultaneously turned on and the short circuit of the series circuit of the first and second capacitors C1 and C2. Considering the delay time at the time of switching between Q1 and Q2, etc., the minimum necessary value is sufficient. Vgs (Q1) and Vgs (Q2) are the gate drive voltage waveforms of the switch elements Q1 and Q2, respectively. As can be seen from these waveforms, one of the first switch element Q1 and the second switch element Q2 is turned off in the on period and the other in the on period, except for a short period of Toff31 and Toff32. One is controlled to turn off, and the constant voltage control of the output voltage Vout is performed by changing the ratio (duty cycle) of the ON period of one switch element to the operation cycle T31.

  Next, in FIG. 2, I (N2) and V (N2) are the current flowing through the second primary winding N2 of the transformer T and the voltage between the terminals, and I (N1) and V (N1) are the transformer T Current flowing through the first primary winding N1 and the voltage between the terminals, I (Q1) and I (Q2) are currents flowing through the first and second switch elements Q1 and Q2, respectively, and Vds ( Q3) and Vds (Q4) are the drain-source voltages of the first and second synchronous rectification MOSFETs Q3 and Q4, respectively, and V (R) is the voltage at the R point.

  Next, the voltage / current waveform of each part in FIG. 2 will be described. First, during a period when the first switch element Q1 is on (Ton31), a current as indicated by I (N2) is applied to the second primary winding N2 of the transformer T from the input power source VIN to the first switch element Q1. It is flowing toward. The slope of this current has a value of Va / LN2, where Va is the voltage of the input power supply VIN and LN2 is the inductance of the second primary winding N2 of the transformer T. On the other hand, in the first primary winding N1 of the transformer T, a current as indicated by I (N1) flows from the first capacitor C1 toward the first switch element Q1 (current flowing during this period). The direction of is positive.) The total value of the currents flowing through the first primary winding N1 and the second primary winding N2 of the transformer T is a value obtained by converting the current flowing through the load to the primary side of the transformer by the turns ratio of the transformer T. The excitation currents of the first primary winding N1 and the second primary winding N2 of the transformer T are added. Therefore, the sum of the current flowing through the first primary winding N1 of the transformer T and the current flowing through the second primary winding N2 flows through the first switch element Q1. This is a current waveform as indicated by I (Q1) in FIG.

  Next, during a period when the second switch element Q2 is ON (Ton32), a current as indicated by I (N2) is applied to the second primary winding N2 of the transformer T from the input power source VIN to the second switch element Q2. It is flowing toward. The slope of this current is represented by Va as the voltage of the input power source VIN, LN2 as the inductance of the second primary winding N2 of the transformer T, and Vb as the voltage of the series circuit of the first and second capacitors C1 and C2. It has a value of (Va−Vb) / (LN2). At this time, the current flowing through the second primary winding N2 of the transformer T passes through the second switch element Q2, passes through the second capacitor C2, and the first capacitor C1 input power source VIN, and passes through the transformer T. It flows along the path returning to the second primary winding N2. On the other hand, during the period when the second switch element Q2 is on (Ton32), a current as indicated by I (N1) flows through the first primary winding N1 of the transformer T. This flows from the second capacitor C2 through the second switch element Q2, through the first primary winding N1 of the transformer T, and back to the second capacitor C2. The value of the current flowing through the first primary winding N1 of the transformer T is equal to the current flowing through the choke coil L of the output filter, as in the period when the first switch element Q1 is on (Ton31). Excitation currents of the first primary winding N1 and the second primary winding N2 of the transformer T are added to the current converted to the primary side of the transformer by the turn ratio. Therefore, during the period when the second switch element Q2 is on, the current I (N2) flowing through the second primary winding N2 of the transformer T flows from the source of the second switch element Q2 toward the drain terminal, The current I (N1) flowing through the first primary winding N1 of the transformer T flows in the direction opposite to the above I (N2) from the drain terminal of the second switch element Q2 toward the source terminal. The switch element Q2 has a difference current between the current I (N2) flowing through the second primary winding N2 of the transformer T and the current I (N1) flowing through the first primary winding N1 of the transformer T. Flowing. This is a current waveform as indicated by I (Q2) in FIG.

  Next, V (N1) in FIG. 2 indicates the voltage between the terminals of the first primary winding N1 of the transformer T. The voltage in the period of Ton31 of this waveform is the ON state of the first switch element Q1. Therefore, it corresponds to the voltage between the terminals of the first capacitor C1, and the voltage during the period Ton32 corresponds to the voltage between the terminals of the second capacitor C2 because the second switch element Q2 is on. Vds (Q3) and Vds (Q4) are the drain-source voltages of the synchronous rectification MOSFETs Q3 and Q4 shown in FIG. 1, respectively, and these voltages are the gate drive voltages of the other synchronous rectification MOSFET, respectively. Yes. V (R) is a voltage waveform at point R. Further, the waveforms of Vds (Q3) and Vds (Q4) are respectively the voltage V (N1) between the terminals of the first primary winding N1 of the transformer T in the period of Ton31 and Ton32, and the second primary. The inter-terminal voltage V (N2) of the winding N3 is applied to the first primary winding N1 of the transformer T and the second primary winding N2 and the first secondary winding portion Na (or the second secondary winding). The voltage V (R) at point R is a waveform obtained by adding the voltage waveforms of Vds (Q3) and Vds (Q4).

  In the waveform of V (R), Vout indicates the output voltage at the output terminals (18a, 18b). The ripple current value at which the voltage V (R) at this point R and the output voltage Vout flow through this point. And a ripple voltage having a value approximately determined by the product of the equivalent series resistance of the third capacitor Cout of the output filter.

  As is apparent from the above description, the embodiment of FIG. 1 has the synchronous rectification MOSFETs indicated by Vds (Q3) and Vds (Q4) in FIG. Except for the short period of Ton32, it always occurs in either one, so there is no problem that the period during which the synchronous rectification MOSFETs Q3 and Q4 cannot be driven becomes long. Further, as can be seen from the waveforms of Vds (Q3) and Vds (Q4), the voltage waveform applied to the synchronous rectification MOSFETs Q3 and Q4 is a rectangular wave, so that the voltage does not rise abnormally. A synchronous rectification MOSFET having a low breakdown voltage and a low on-resistance can be used.

  Next, FIG. 3 explains the characteristics of the output voltage with respect to the duty cycle of FIG. 1 (the ratio of the on period to the operation cycle of the main switch Q1). In FIG. 1, the voltage of the input power source VIN is Va, the first and second switch elements, the duty cycles of Q1 and Q2 (ratio of the ON period to the operation period of the switch elements) are D and 1-D, respectively. And V (C1) and V (C2), respectively, between the terminals of the first and second capacitors C1 and C2, and the first primary winding N1, the second primary winding N2 and the first secondary winding of the transformer T The turn ratio with the line portion Na (or the second secondary winding Nb) is N: 1, and the drain-source voltages of the first and second synchronous rectification MOSFETs Q3 and Q4 are Vds (Q3 ), Vds (Q4), and the output voltage at the output terminals (18a, 18b) as Vout, the following equation holds.

V (C1) + V (C2) = Va / (1-D) (1)
(However, in the subsequent numerical analysis, the first and second switching elements Q1, Q2 and the first and second synchronous rectification MOSFETs, the voltage drop during conduction in Q3 and Q4, the first and second (The period of Toff31 and Toff32 in which both of the switching elements Q1 and Q2 are off is ignored as being very small.)

  In addition, regarding the operation of the core (magnetic material) of the transformer T, the amount excited when the first switch element Q1 is on is equal to the amount reset when the second switch element Q2 is on. The following equation holds.

V (C1) × D = V (C2) × (1-D) (2)
The following equation is derived from the above equation (1) and the above equation (2).

  V (C1) = Va (3)

  V (C2) = Va × D / (1-D) (4)

  The drain-source voltages of the first and second synchronous rectification MOSFETs Q3 and Q4 when turned off are the terminal voltages of the first and second capacitors C1 and C2, respectively. Since the voltage is converted by the number ratio, the following equation holds.

  Vds (Q3) = V (C1) / N = Va / N (5)

  Vds (Q4) = V (C2) / N = Va * D / {(1-N) * N} (6)

  The output voltage at the output terminals (18a, 18b) is a value obtained by averaging the voltage at the point R with the output capacitor Cout. The voltage at the point R is the first and second synchronous rectification MOSFETs, Since this is a voltage obtained by adding Vds (Q3) and Vds (Q4), which are drain-source voltages of Q3 and Q4, the following equation holds when the switching period is To.

  From Equation 1, it can be seen that in the circuit of FIG. 1, the output voltage Vout is proportional to the duty cycle D (the ratio of the ON period to the operating cycle of the main switch Q1), which is illustrated in FIG.

  Here, since the output characteristic of FIG. 3 is a proportional characteristic, it is possible to set the duty cycle to 0.5 under the condition that the input and output are rated. At this time, the first and second synchronous characteristics can be set. It can be seen that Vds (Q3) and Vds (Q4), which are the drain-source voltages of the rectifying MOSFETs Q3 and Q4, are both Va / N from the above equations (5) and (6). Therefore, even if the input / output conditions change, it remains a square wave and changes around this value. As in the conventional circuit example, it is necessary to use a synchronous rectifier MOSFET that has a particularly high breakdown voltage and a high on-resistance. There is no. Furthermore, the fact that the voltages of Vds (Q3) and Vds (Q4) are the same (in fact, they slightly deviate because the number of turns of the transformer T is an integer) means that the change of the voltage V (R) at the point R is Even if the voltage during the period of Ton31 and the voltage during the period of Ton32 are the same or different, the difference is very small, and as a result, the output filter is reduced. Can do. The validity of the above analysis results has been confirmed by experiments.

  Further, I (N1) in FIG. 2 indicates the current flowing through the first primary winding N1 of the transformer T. Generally, since the transformer T has a leakage inductance, both switch elements are turned off. By properly adjusting Toff31 and Toff32, which are the current periods, the current flowing through the leakage inductance is changed after the one switch element is turned off and before the other switch element is turned on. The parasitic capacitance between the drain and source of the switch element can be discharged, and so-called ZVS operation can be performed. As a result, the energy stored in the parasitic capacitance between the drain and source of the switch element can be recovered, and the efficiency of the switching power supply can be increased.

  7 and 8 are structural diagrams of the transformer T. FIG. In the transformer of this embodiment, four magnetic legs are provided in parallel on the bottom plate, and the first core is arranged so that the distance between one outer magnetic leg and the inner magnetic leg is equal. And a flat plate-like second core that has substantially the same shape as the bottom plate of the first core and is attached to the ends of the four magnetic legs of the first core. The first primary winding N1 and the second primary winding N2 are wound around the inner two magnetic legs, the first secondary winding Na is wound around the inner magnetic leg, and the other The second secondary winding Nb is wound around the inner magnetic leg of the.

  FIG. 9 is an equivalent circuit of the transformer T in the proposed system. Each of the first primary winding N1 and the second primary winding N2 can be represented by equivalently connecting two windings in series. If the circuit diagram of the embodiment of FIG. 1 is redrawn with this transformer equivalent circuit diagram, the equivalent circuit diagram of FIG. 10 is obtained. The operation of the transformer T at this time will be described using the transformer equivalent circuit operation diagram 1 shown in FIG. When the primary current flows in the direction of the arrow, no current flows through the first primary winding N1, and the exciting inductance LN1 is charged. The primary current flows through the second primary winding N2 and supplies power to the load through the secondary winding Nb. At this time, the excitation inductance LN1 plays the same role as the output choke coil, and the output choke coil is not required on the secondary side. When the direction of the primary current is reversed, the transformer equivalent circuit operation shown in FIG. 12 performs the same operation as in the previous stage. At this time, the energy accumulated in the excitation inductance LN1 is discharged in the previous stage. In addition to the proposed method, this effect can be obtained in a method of performing a push-pull operation in which the primary side current of the transformer T flows alternately in the positive direction and the negative direction by turning on and off the primary side switching element.

  As is apparent from the above description, in the circuit of FIG. 1, either one of the synchronous rectification MOSFETs Q3 and Q4 is always driven except for a short period of Toff31 and Toff32 (current flowing through the load flows). Since the voltage between the drain and the source is low, it is possible to use a low withstand voltage and low on-resistance. Further, since the output choke coil can be reduced, the power loss here is also small. As a result, a highly efficient switching power supply device can be made.

  In the circuit diagram of FIG. 1, the first and second switch elements, Q1 and Q2, use N-channel MOSFETs, but either or both of them use P-channel MOSFETs. The circuit operation is exactly the same. The first and second switch elements, Q1 and Q2, are not limited to MOSFETs, and the circuit operation is exactly the same even if, for example, an IGBT is used.

  Further, the embodiment shown in FIG. 13 operates substantially the same as the embodiment shown in FIG. In the embodiment shown in FIG. 13, a transformer as shown in FIG. 14, specifically, four magnetic legs are provided in parallel on the bottom plate, and one of these four magnetic legs and the inner magnetic leg. The first core is erected so that the interval between the first core and the bottom plate of the first core is substantially the same shape, and the flat plate is attached to the ends of the four magnetic legs of the first core. A first winding N1 is wound around the inner two magnetic legs, the first secondary winding Na is wound around one inner magnetic leg, and the other inner A transformer constituted by winding the second secondary winding Nb is provided around the magnetic leg. Further, a choke coil L1 is connected to one input terminal 2a and one end of the primary winding N1 of the transformer T. With the above configuration, an effect of reducing the output choke coil can be obtained.

  Next, a driving method of the first and second synchronous rectification MOSFETs Q3 and Q4 in the circuit diagram of FIG. 1 will be described. The first and second synchronous rectification MOSFETs Q3 and Q4 in the circuit diagram of FIG. 1 drive the gate terminal by the drain-source voltage of the other synchronous rectification MOSFET, respectively. Not only the method shown in FIG. 1, but the same effect can be obtained if the voltage is obtained from the winding of the transformer T. An example of another driving method of the synchronous rectification MOSFET is shown in FIG. Here, regarding the driving method of the synchronous rectification MOSFET, the circuit operation of FIG. 5 is completely equivalent to the circuit operation of FIG.

(The invention's effect)
As apparent from the above description, in the circuit of the present invention, the synchronous rectification MOSFETs Q3 and Q4 can always be driven except for a short period of Toff31 and Toff32, and the synchronous rectification MOSFETs Q3 and Q4 are driven. Since the withstand voltage can be reduced, one having a low on-resistance can be used. Further, the output choke coil L of the output filter can be reduced, and the mounting area can be reduced. As a result, a highly efficient switching power supply device can be made. This has a great effect when a high-efficiency switching power supply with a low output voltage (for example, 3.3 V output or less) and a large output current is made by communication or the like.

Circuit diagram of one embodiment of the present invention Fig. 1 Voltage and current waveform of each part Output voltage characteristic diagram of the embodiment of the present invention Another embodiment of the drive circuit applied to the embodiment of the present invention Conventional circuit diagram 1 Conventional circuit diagram 2 Transformer structure diagram 1 Transformer structure 2 Transformer equivalent circuit diagram Example equivalent circuit diagram Transformer equivalent circuit operation diagram 1 Transformer equivalent circuit operation diagram 2 Circuit diagram of another embodiment FIG. 13 shows an example of a transformer structure.

Explanation of symbols

VIN input power supply Q1, Q2 Switch element (MOSFET)
Q3, Q4 synchronous rectification MOSFET
C1, C2 Capacitor T Transformer N1 First primary winding N2 Second primary winding Na First secondary winding Nb Second secondary winding L Choke coil Cout Capacitors 2a, 2b Input terminal 17 Load 18 Control Circuit 18a, 18b Output terminal

Claims (3)

Two input terminals for receiving a DC voltage, and a transformer functioning as two transformer types by a first transformer section and a second transformer section,
The first transformer part is formed with two coils on the primary side and a coil on the secondary side,
In the second transformer section, two coils on the primary side and a coil on the secondary side are formed,
One coil of the primary side of the first transformer unit and one coil of the primary side of the second transformer unit are connected in series to form a first coil group,
The other coil on the primary side of the first transformer part and the other coil on the primary side of the second transformer part are connected in series to form a second coil group,
One end of the first coil group is connected to one end of the input terminal,
The other end of the first coil group is connected to a first switch element connected to the other end of the input terminal,
One end of the second coil group is connected to the other end of the input terminal via a first capacitor,
The other end of the second coil group is connected to the other end of the first coil group, and the second coil group is connected via a second switch element and a second capacitor connected in series. Connected to one end of the
The secondary coil of the first transformer unit and the secondary coil of the second transformer unit are connected to a synchronous rectifier circuit,
The synchronous rectification is performed by alternately repeating the OFF state of the first switch element and the ON state of the second switch element, and the ON state of the first switch element and the OFF state of the second switch element. A switching power supply that outputs a DC voltage from the output terminal of the circuit.   In the synchronous rectifier circuit, one output terminal is commonly connected to one end of a secondary coil of the first transformer section and one end of a secondary coil of the second transformer section, and the other output terminal 2. The switch is configured by a switch element that conducts a terminal to one of the other end of the secondary side coil of the first transformer unit and the other end of the secondary side coil of the second transformer unit. The switching power supply device described in 1.   2. The switching power supply device according to claim 1, further comprising: a control circuit that detects an output voltage of an output terminal of the synchronous rectifier circuit and controls on and off of the first switch element and the second switch element alternately. .

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