JP2007267510A - Inverter driver and driving method - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide inverter driver and driving method which can drive an inverter circuit such that the rotor of a motor can be brought into preset rotary state correctly through a simple arrangement. <P>SOLUTION: A main controller 4 is provided with first analog rotary information representative of rotary state of the rotor of an induction motor 2 from a resolver 3. The main controller 4 supplies power for driving the induction motor 2 based on an angular speed command value Ω<SB>REF</SB>1. The main controller 4 provides first and second subcontrollers 5 and 6 with second analog sine value data and second analog cosine value data representative of rotary state of the rotor through a first transmission line 8. The first and second subcontrollers 5 and 6 supply power for driving the induction motor 2 in synchronism with the main controller 4 based on the second analog sine value data and second analog cosine value data provided from the main controller 4. <P>COPYRIGHT: (C)2008,JPO&INPIT

Description

  The present invention relates to an inverter driving apparatus and an inverter driving method for driving an inverter circuit that supplies electric power to an electric motor.

There is a resolver / digital conversion device for obtaining a rotation angle and an angular velocity of a rotating device based on analog data given from a resolver (see, for example, Patent Document 1). The number of inverter drive signals for driving the inverter circuit that supplies power to the multiphase induction motor needs to be twice the number of phases of the induction motor. The number of inverter drive signals that a single microcomputer can generate depends on the computing power of the microcomputer and the PWM (Pulse
There is a limit because the number of electrical signals that can be modulated by Width Modulation is limited. Therefore, in order to drive the multiphase induction motor, inverter drive signals for driving a plurality of inverter circuits are generated by a plurality of microcomputers. In this case, since it is necessary to synchronize the amplitude and phase of the voltage applied from the plurality of inverter circuits to the multiphase induction motor, the plurality of microcomputers need to generate a plurality of inverter drive signals in synchronization with each other. is there. In order for a plurality of microcomputers to generate an inverter drive signal in synchronization with each other, a digital value synchronization signal is provided from one microcomputer to another microcomputer, and the other microcomputers receive the digital value synchronization signal. Based on this, signals for driving a plurality of inverter circuits are generated in synchronization with each other.

Japanese Patent No. 3408238

  In some cases, noise is superimposed on a digital synchronization signal supplied from one microcomputer to another microcomputer, and an accurate synchronization signal may not be transmitted to the other microcomputer. The noise includes noise generated when the output of the inverter is switched on and off, noise due to radio waves, and the like. Even if only one bit of the sync signal is inverted due to noise, for example, if the inverted 1 bit is a code bit representing a sign, the sync signal with the inverted sign is given to another microcomputer. It becomes. In this case, there may be a case where the rotor operates to decelerate despite the command to accelerate the rotor. As described above, when the synchronization signal changes due to noise and an erroneous synchronization signal is transmitted to another microcomputer, there is a problem that the control of the motor is seriously affected.

  In order to prevent an erroneous synchronization signal from being transmitted to another microcomputer due to noise, it is detected whether the synchronization signal sent from one microcomputer matches the synchronization signal received by the other microcomputer. It is conceivable to provide an error detection function for communication control. Specifically, one microcomputer and another microcomputer are connected by an RS-232C cable, and a synchronization signal is given from one microcomputer to another microcomputer according to a communication procedure compliant with the RS-232C standard. In this case, since it becomes necessary to follow a communication procedure compliant with the RS-232C standard, there arises a problem that control of communication for providing a synchronization signal from one microcomputer to another microcomputer becomes complicated. Further, when the communication control is complicated, the response of the communication control depends on the communication speed, which causes a problem that the response of the communication control is deteriorated.

  Accordingly, an object of the present invention is to provide an inverter driving device and an inverter driving method capable of driving an inverter circuit so that a rotor of an electric motor can be accurately set in a preset rotational state with a simple configuration. .

In the present invention, the first analog rotation information that represents the rotation state as an analog value is provided from the rotation information acquisition unit that acquires the rotation state of the rotor of the motor, and the first inverter circuit that drives the first inverter circuit that supplies electric power to the motor is provided. 1 inverter driving means, a transmission path for transmitting second analog rotation information that is output from the first inverter driving means and that represents a command for setting the rotor in a preset rotational state as an analog value, and via the transmission path Second inverter driving means for driving a second inverter circuit for supplying electric power to the electric motor based on second analog rotation information given from the first inverter driving means,
The first inverter drive means
A first analog / digital converter that converts the first analog rotation information into first digital rotation information represented by a digital value;
A first inverter drive unit that drives the first inverter circuit so that the rotor is in a preset rotation state;
A digital / analog conversion unit that generates the second analog rotation information based on the first digital rotation information and supplies the second analog rotation information to the second inverter driving means via the transmission line;
The second inverter driving means is
A second analog / digital converter that converts the second analog rotation information given through the transmission path into second digital rotation information represented by a digital value;
And a second inverter driving unit that drives the second inverter circuit in synchronization with the first inverter circuit so that the rotor is in a preset rotation state based on the second digital rotation information. It is an inverter drive device.

  According to the present invention, the second analog rotation information is given from the first inverter driving means to the second inverter driving means through the transmission line. The second inverter driving means drives the second inverter circuit based on the second analog rotation information so that the second inverter circuit operates in synchronization with the first inverter circuit. This second analog rotation information is represented by an analog value.

  When the second analog rotation information is represented by a digital value instead of an analog value, this information may greatly change due to noise while passing through the transmission path. For example, the second inverter drive unit may receive information in which one bit of information represented by a digital value is inverted due to the influence of noise while passing through the transmission path. Even if only 1 bit of the information represented by the digital value is inverted, if the inverted 1 bit is a sign bit representing a sign, the information with the inverted sign is given to the second inverter driving means. It will be. Since the second inverter driving means drives the second inverter circuit based on the information with the sign inverted, there is a possibility that the rotor of the electric motor greatly deviates from a preset rotational state. In the present invention, since the second analog rotation information is represented by an analog value, even if the second analog rotation information changes due to noise while passing through the transmission line, only the change is given to the second inverter driving means as noise. Therefore, the second analog rotation information is less likely to undergo a large change while passing through the transmission line, as compared with the case where the second analog rotation information is represented by a digital value. As described above, the first inverter driving unit can accurately control the communication between the first inverter driving unit and the second inverter driving unit from the first inverter driving unit with a simple configuration that does not include complicated control such as an error detection function. 2 Analog rotation information can be transmitted to the second inverter drive means. Further, since complicated control is not provided for communication control, the response control of communication is good. Based on the second analog rotation information, the second inverter driving means drives the second inverter circuit so as to synchronize with the first inverter circuit. Thus, the inverter drive device can control the rotation state of the rotor to be accurately set to a preset rotation state.

In the present invention, the first inverter drive unit drives the first inverter circuit based on the first digital rotation information and the slip frequency control,
The digital / analog conversion unit generates the second analog rotation information based on first digital rotation information and slip frequency control.

  According to the present invention, the second analog rotation information is generated based on the first digital rotation information and the slip frequency control. Since the second inverter driving unit drives the second inverter circuit based on the second analog rotation information, an inverter driving device that drives the first and second inverter circuits synchronously by slip frequency control is realized. As described above, since the accurate second analog rotation information is given from the first inverter driving means to the second inverter driving means, the second inverter driving unit accurately synchronizes the second inverter circuit with the first inverter circuit. Can be driven.

In the present invention, the second inverter driving means
An estimated angle value calculation unit for obtaining an estimated angle value from the second digital rotation information using a second-order lag element;
An estimated angular velocity value calculation unit for obtaining an estimated angular velocity value from the second digital rotation information using a second-order lag element;
The second inverter driving unit drives the second inverter circuit based on the estimated angle value and the estimated angular velocity value.

  According to the present invention, since the estimated angle calculation unit and the estimated angular velocity calculation unit constitute a low-pass filter, even if high-frequency noise is superimposed on the second analog rotation information, the superimposed high-frequency noise is removed. can do. The estimated angle calculation unit and the estimated angular velocity calculation unit can mainly remove high-frequency noise that occurs when the inverter circuit is switched on and off. Therefore, the estimated angle calculation unit can calculate the estimated angle from which the influence of noise is removed. In addition, the estimated angular velocity calculation unit can calculate an estimated angular velocity value from which the influence of noise has been removed. Thereby, the estimated angle calculation unit can calculate an estimated angle value corresponding to the actual rotation angle of the rotor, and the estimated angular velocity calculation unit calculates an estimated angular velocity value corresponding to the actual rotor angular velocity. be able to.

According to the present invention, the first analog rotation information that represents the rotation state as an analog value is provided from the rotation information acquisition unit that acquires the rotation state of the rotor of the motor, and the first inverter driving unit supplies power to the motor. Based on the second analog rotation information representing an instruction for driving the inverter circuit and setting the rotor given from the first inverter driving means via the transmission path to a preset rotation state by an analog value, the second An inverter driving method for driving a second inverter circuit for supplying electric power to an electric motor by inverter driving means,
The first inverter drive means
Converting the first analog rotation information into first digital rotation information represented by a digital value;
Driving the first inverter circuit so that the rotor is in a preset rotation state;
Based on the first digital rotation information, the second analog rotation information is generated, and the second analog rotation information is provided to the second inverter driving means via the transmission path,
The second inverter driving means is
Converting the second analog rotation information given through the transmission path into second digital rotation information represented by a digital value;
The inverter driving method is characterized in that, based on the second digital rotation information, the second inverter circuit is driven in synchronization with the first inverter circuit so that the rotor is in a preset rotation state.

  According to the present invention, the first inverter driving means provides the second analog rotation information to the second inverter driving means via the transmission line. Based on the second analog rotation information, the second inverter driving means drives the second inverter circuit so that the second inverter circuit operates in synchronization with the first inverter circuit. This second analog rotation information is represented by an analog value.

  When the second analog rotation information is represented by a digital value instead of an analog value, this information may greatly change due to noise while passing through the transmission path. For example, the second inverter drive unit may receive information in which one bit of information represented by a digital value is inverted due to the influence of noise while passing through the transmission path. Even if only 1 bit of the information represented by the digital value is inverted, if the inverted 1 bit is a sign bit representing a sign, the information with the inverted sign is given to the second inverter driving means. It will be. Since the second inverter driving means drives the second inverter circuit based on the information with the sign inverted, there is a possibility that the rotor of the electric motor greatly deviates from a preset rotational state. In the present invention, since the second analog rotation information is represented by an analog value, even if the second analog rotation information changes due to noise while passing through the transmission line, only the change is given to the second inverter driving means as noise. Therefore, the second analog rotation information is less likely to undergo a large change while passing through the transmission line, as compared with the case where the second analog rotation information is represented by a digital value. As described above, the first inverter driving unit can accurately control the communication between the first inverter driving unit and the second inverter driving unit from the first inverter driving unit with a simple configuration that does not include complicated control such as an error detection function. 2 Analog rotation information can be transmitted to the second inverter drive means. Further, since complicated control is not provided for communication control, the response control of communication is good. Based on the second analog rotation information, the second inverter driving means drives the second inverter circuit so as to synchronize with the first inverter circuit. Thus, the inverter drive device can control the rotation state of the rotor to be accurately set to a preset rotation state.

  Accurate second analog rotation information can be obtained from the first inverter driving means by a simple configuration that does not provide complicated control such as an error detection function in the control of communication between the first inverter driving means and the second inverter driving means. Can be transmitted to the second inverter driving means. Further, since complicated control is not provided for communication control, the response control of communication is good. Based on the second analog rotation information, the second inverter driving means drives the second inverter circuit so as to synchronize with the first inverter circuit. Thus, the inverter drive device can control the rotation state of the rotor to be accurately set to a preset rotation state.

  Further, according to the present invention, the estimated angle calculation unit can calculate the estimated angle value from which the influence of noise has been removed. In addition, the estimated angular velocity calculation unit can calculate an estimated angular velocity value from which the influence of noise has been removed. Thereby, the estimated angle calculation unit can calculate an estimated angle value in accordance with the actual rotation angle of the rotor, and the estimated angular velocity calculation unit calculates an estimated angular velocity value according to the actual angular velocity of the rotor. be able to.

FIG. 1 is a block diagram showing a configuration of a control device 1, an induction motor 2, a resolver 3, and a driver 18 according to an embodiment of the present invention. The control device 1 includes first analog rotation information representing the rotation angle φ that is the rotation state of the rotor provided from the resolver 3 connected to the rotor of the induction motor 2, and an angular velocity command value Ω _REF provided to the control device 1. 1, the rotation state of the rotor of the induction motor 2 is controlled to drive the induction motor 2. The control device 1 includes a main control device 4, a first sub control device 5, a second sub control device 6, and a printed board. The main control device 4, the first sub control device 5, and the second sub control device 6 are mounted on a printed circuit board. A conductive printed wiring is formed on the printed circuit board, and the main control device 4 and the first and second sub-control devices 5 and 6 are electrically connected by the printed wiring. A printed wiring that electrically connects the main control device 4 and the first and second sub-control devices 5 and 6 is referred to as a first transmission path 8. The main controller 4 and the resolver 3 are electrically connected by a second transmission path 9 including a printed wiring and a conductive cable.

  The main control device 4, the first sub control device 5, and the second sub control device 6 are each realized by a microcomputer.

  The induction motor 2 is a nine-phase induction motor in the present embodiment, and includes a rotor and a stator. The first to third U-phase windings, the first to third V-phase windings, and the first to third W-phase windings are wound around the stator of the induction motor 2.

  Main controller 4 supplies power to the first U-phase winding, the first V-phase winding, and the first W-phase winding. The first sub-control device 5 supplies power to the second U-phase winding, the second V-phase winding, and the second W-phase winding. Second sub-control device 6 supplies power to the third U-phase winding, the third V-phase winding, and the third W-phase winding. The voltage applied to the first U-phase winding is described as a first U-phase winding voltage U1, the voltage applied to the first V-phase winding is described as a first V-phase winding voltage V1, and the first W-phase winding is described. The voltage applied to the line is referred to as a first W-phase winding voltage W1. The voltage applied to the second U-phase winding is described as a second U-phase winding voltage U2, the voltage applied to the second V-phase winding is described as a second V-phase winding voltage V2, and the second W-phase winding is described. The voltage applied to the winding is referred to as a second W-phase winding voltage W2. The voltage applied to the third U-phase winding is described as a third U-phase winding voltage U3, the voltage applied to the third V-phase winding is described as a third V-phase winding voltage V3, and the third W-phase winding is described. A voltage applied to the winding is referred to as a third W-phase winding voltage W3.

  FIG. 2 is a diagram showing the phasor display of the voltage applied to each winding when the phase of the first U-phase winding voltage U1 is 0 (rad). The main control device 4, the first sub control device 5 and the second sub control device 6 apply a voltage having a predetermined phase relationship to each winding, and generate a rotating magnetic field by causing a current to flow through each winding to rotate. Control the rotation state of the child. Specifically, the phase of the first V-phase winding voltage V1 is advanced by a first angle Θ1 (rad) from the phase of the first U-phase winding voltage U1, and the phase of the first W-phase winding voltage W1 is The second angle Θ2 (rad) is advanced from the phase of the phase winding voltage U1. The first angle Θ1 is 2 · π / 3. The second angle Θ2 is 4 · π / 3. The phase of the second V-phase winding voltage V2 is advanced by a first angle Θ1 (rad) from the phase of the second U-phase winding voltage U2, and the phase of the second W-phase winding voltage W2 is the second U-phase winding voltage U2. Is advanced by a second angle Θ2 (rad) from the phase of. The phase of the third V-phase winding voltage V3 is advanced by a first angle Θ1 (rad) from the phase of the third U-phase winding voltage U3, and the phase of the third W-phase winding voltage W3 is the third U-phase winding voltage U3. Is advanced by a second angle Θ2 (rad) from the phase of. The phase of the second U-phase winding voltage U2 is advanced by a third angle Θ3 (rad) from the first U-phase winding voltage U1, and the phase of the third U-phase winding voltage U3 is the first U-phase winding voltage U1. Is delayed by the third angle Θ3 (rad) from the phase of. The third angle Θ3 is 2 · π / 9. “Π” is a pi, and “•” is a product symbol.

  Thus, in order for main controller 4, first sub controller 5 and second sub controller 6 to apply a voltage to the windings wound around the rotor in synchronization with each other, main controller 4 Second analog rotation information for synchronization is given to the first and second sub-control devices 5 and 6 via the first transmission path 8, and the first and second sub-control devices 6 are given from the main control device 4. A voltage is applied to each winding based on the second analog rotation information.

  FIG. 3 is a diagram schematically showing the configuration of the resolver 3 as rotation information acquisition means. The resolver 3 is connected to the rotor of the induction motor 2, and the first analog rotation information including the sine sinφ and the cosine cosφ of the rotation angle φ of the rotor is rotated via the second transmission path 9 to the main controller 4 to be described later. This is given to the information calculation unit 7. The resolver 3 is a resolver called a one-phase excitation two-phase output method. The resolver 3 includes an R phase winding 11, an A phase winding 12 and a B phase winding 13. The R-phase winding 11 is also called an excitation winding and is wound around a rotor. When the rotor rotates, the magnetic axis of the R-phase winding 11 rotates around a rotation axis of the rotor on a virtual plane perpendicular to the rotation axis of the rotor. The A-phase winding 12 and the B-phase winding 13 are wound around the stator such that virtual planes perpendicular to the magnetic axes are orthogonal to each other so as not to be magnetically coupled to each other.

The R-phase winding 11, with the angular frequency omega, the amplitude is applied excitation voltage f (t) of the rectangular wave of V _R, a magnetic field around the R-phase winding 11 is formed. The excitation voltage f (t) is expressed by the following equation (1).
_{f (t) = V R ·} sgn (sinωt) ... (1)

  In equation (1), t is time and “sin” is a sine function. In equation (1), “sgn” is a sign function, and takes a value of +1 when sin ωt is 0 or more, and takes a value of −1 when sin ωt is less than 0.

When an excitation voltage f (t) is applied to the R-phase winding 11 and a magnetic field is formed around the R-phase winding 11, a voltage V _A is induced in the A-phase winding 12 by the induced electromotive force, and the B-phase A voltage V _B is induced in the winding 13. In this embodiment, voltage _VA induced in phase A winding 12 of resolver 3 is referred to as phase A winding voltage _VA, and voltage V _B induced in phase B winding 13 is referred to as phase B winding. to as voltage _{V B.} The A-phase winding voltage V _A and the B-phase winding voltage V _B are expressed by the following equation (2) and are a function of the angular position φ of the rotor.

  In equation (2), “cos” is a cosine function. In Equation (2), M is a transformation ratio when the magnetic axis of the R-phase winding 11 coincides with any one of the A-phase winding 12 and the B-phase winding 13. The transformation ratio M takes a positive value and is determined by the structure of the resolver 3 including the A-phase winding 12 and the B-phase winding 13.

If M · f (t) is eliminated from the equation (2), the rotation angle φ of the rotor is expressed by the following equation (3).
φ = tan ⁻¹ (V _A / V _B ) (3)

In Expression (3), “tan ⁻¹ ” is an arc tangent function. When the A-phase winding voltage V _A and the B-phase winding voltage V _B are measured, the rotation angle φ of the rotor can be obtained from Equation (3).

As described above, the resolver 3 is configured such that the A-phase winding voltage V _A that represents the rotation angle φ of the rotor as a sine and the rotation angle φ of the rotor from the A-phase winding 12, as represented by the equation (2). The first analog rotation information including the B-phase winding voltage V _B that is expressed by a cosine is output. Of the first analog rotation information, information including the A-phase winding voltage V _A representing the rotation angle φ of the rotor as a sine is referred to as first analog sine value data. Of the first analog rotation information, information including the B-phase winding voltage V _B that represents the rotation angle φ of the rotor with a cosine is referred to as first analog cosine value data.

Main controller 4 includes first inverter driving means realized by executing software, and first inverter circuit 51 </ b> A that supplies power to induction motor 2. The first inverter driving means includes a rotation information calculation unit 7 corresponding to a first analog / digital conversion unit that converts the first analog rotation information into first digital rotation information represented by a digital value, and a rotor is set in advance. Based on the first digital rotation information and the first inverter driving unit that drives the first inverter circuit 51A so as to achieve the rotated state, the second analog rotation is generated via the transmission line. And a digital / analog converter for supplying information to the second inverter driving means. The first inverter drive unit includes an angle command value calculator 19, a first V / F calculator 41A, a first PWM modulation factor generator 42A, first to third sine value generators 43A, 44A, 45A, first to third. Multipliers 46A, 47A, 48A and a first comparator 54A are included. Further, the digital / analog conversion unit that supplies the second analog rotation information to the second inverter driving means via the transmission line includes a fourth sine value generator 52, a first cosine value generator 53, and a first digital / analog converter 55. And a second digital / analog converter 56. The angular velocity command value Ω _REF 1 is given to the rotation information calculation unit 7 of the main controller 4. In this embodiment, a digital / analog converter may be abbreviated as a D / A converter in some cases.

  FIG. 4 is a block diagram illustrating a configuration of the rotation information calculation unit 7. Based on the first analog sine value data and the first analog cosine value data of the analog values given from the resolver 3, the rotation information calculation unit 7 has a first estimated angle value θ1, which is an estimated value of the rotation angle φ of the rotor, And a first estimated angular velocity value Ω1, which is an estimated value of the rotational angular velocity ψ of the rotor, is calculated. The first estimated angle θ1 and the first estimated angular velocity Ω1 calculated by the rotation information calculation unit 7 are digital values.

  The rotation information calculator 7 includes a first analog / digital converter 14A, a second analog / digital converter 15A, a counter timing circuit 17, a first synchronous rectifier 21, a second synchronous rectifier 22, a fourth multiplier 23A, 5 multiplier 24A, first subtractor 25A, first integrator 26A, second integrator 27A, first phase compensator 31A, first adder 32A, fifth sine value generator 33A, and second cosine value generator 34A is included. In this embodiment, an analog / digital converter may be abbreviated as an A / D converter.

  The counter / timing circuit 17 gives the driver 18 an R-phase drive signal representing the waveform of the excitation voltage f (t). The counter / timing circuit 17 gives a sign signal representing a sign function sgn synchronized with the R-phase drive signal to the first synchronous rectifier 21 and the second synchronous rectifier 22. The counter / timing circuit 17 supplies a timing signal synchronized with the R-phase drive signal to the first A / D converter 14A and the second A / D converter 15A.

  The driver 18 is realized by an amplifier circuit such as a buffer integrated circuit. Based on the R-phase drive signal representing the waveform of the excitation voltage f (t) given from the counter / timing circuit 17, the driver 18 generates an excitation voltage that becomes a rectangular wave represented by the equation (1) in the R-phase winding 11. It is amplified to a predetermined voltage so that f (t) can be applied, and applied to the R-phase winding 11. The gain of the driver 18 may be 1, for example. In the present embodiment, the excitation voltage f (t) is not a sine wave but a rectangular wave, so that a linear amplifier and a sine wave oscillation circuit for driving the resolver 3 are not required.

The first A / D converter 14A samples the first analog sine value data supplied from the A-phase winding 12 based on the timing signal supplied from the counter / timing circuit 17, and converts it into the first digital sine value data. Then, the converted first digital sine value data is supplied to the first synchronous rectifier 21. The first digital sine value data and the first analog sine value data include the A-phase winding voltage V _{A of the} resolver 3 expressed by Expression (2).

The second A / D converter 15A samples the first analog cosine value data provided from the B-phase winding 13 based on the timing signal provided from the counter / timing circuit 17, and converts it to the first digital cosine value data. Then, the converted first digital cosine value data is supplied to the second synchronous rectifier 22. The first digital cosine value data and the first analog cosine value data include the B-phase winding voltage V _{B of the} resolver 3 expressed by Expression (2).

  In general, an A / D converter may drift in gain and offset of its transfer function. Due to such a drift, the data output from the A / D converter includes a DC component error ΔV. Accordingly, the first digital sine value data and the first digital cosine value data supplied from the first and second A / D converters 14A and 15A to the first and second synchronous rectifiers 21 and 22 respectively have a DC component error ΔV. included.

  The first and second synchronous rectifiers 21 and 22 have the functions of a high-pass filter and a synchronous rectifier. The first synchronous rectifier 21 removes the carrier wave component and the DC component contained in the first digital sine value data supplied from the first A / D converter 14A based on the sign signal supplied from the counter / timing circuit 17, and this The first synchronous rectifier data subjected to such processing is supplied to the fourth multiplier 23A. The second synchronous rectifier 22 removes the carrier wave component and the DC component contained in the first digital cosine value data supplied from the second A / D converter 15A based on the sign signal supplied from the counter / timing circuit 17, The second synchronous rectifier data subjected to such processing is supplied to the fifth multiplier 24A. The carrier wave component is the excitation voltage f (t) shown in Equation (1). Since the first and second synchronous rectifiers 21 and 22 remove the DC component, the DC component error ΔV included in the first digital sine value data and the first digital cosine value data is equal to the first and second synchronous rectifiers 21 and 22. Is removed by passing through 22.

In order to remove the carrier wave component included in the first digital sine value data, that is, the excitation voltage f (t), the first synchronous rectifier 21 performs a sampling time (k−1) one sampling before the current sampling time k. The first digital sine value data A _{k at} the current sampling time _k is subtracted from the first digital sine value data A _k−1 , and the sign of the current sampling time k is subtracted from this difference (A _k−1 −A _k ). the function sgn calculates the multiplied value _{(a k-1 -A k)} · sgn, the first synchronous rectifier data including the thus calculated value by _{(a k-1 -A k)} · sgn fourth multiplier To the container 23A. Sampling is performed twice in one cycle of the excitation voltage f (t). Once, sampling is performed at a time within a half cycle where the excitation voltage f (t) takes a positive value, and once, sampling is performed at a time within a half cycle where the excitation voltage f (t) takes a negative value. . The sampling time interval is approximately a half cycle of the excitation voltage f (t). The sign function sgn at the sampling time k is given from the counter / timing circuit 17. Thereby, the carrier wave component included in the first digital sine value data, that is, the excitation voltage f (t) can be removed. The value (A _k−1 −A _k ) · sgn calculated by the first synchronous rectifier 21 represents the sine sin φ of the rotation angle φ.

Also, the first digital sine value data A _k of the current sampling time _k, and the sampling time (k-1) first digital sine value data A _k-1 of include, but are DC component error [Delta] V, the difference ( A _k−1 −A _k ) does not include the DC component error ΔV because the DC component errors ΔV included therein cancel each other. As a result, the second synchronous rectifier 22 can remove the DC component error ΔV included in the first digital sine value data. The value calculated by the second synchronous rectifier 22 represents the cosine cos φ of the rotation angle φ of the rotor.

The second synchronous rectifier 22 performs the same processing as the first synchronous rectifier 21 on the first digital cosine value data provided from the second A / D converter 15A. Thus, the DC component of the exception of excitation voltage f (t) from the B phase winding voltage A _B, also occurs when the first 2A / D converter 15A for converting the first analog cosine value data to the first digital cosine value data The influence of the error ΔV can be removed.

  The fourth multiplier 23A is a sine sin φ of the rotation angle φ of the rotor given from the first synchronous rectifier 21, and a cosine cos θ1 of the first estimated angle value θ 1 of the rotor given from the second cosine value generator 34A described later. And the fourth multiplier data including the value sinφ · cos θ1 obtained in this way is given to the first subtractor 25A. The fifth multiplier 24A multiplies the cosine cos φ of the rotation angle φ of the rotor given from the second synchronous rectifier 22 and the sine sin θ1 of the estimated angular position of the rotor given from the fifth sine value generator 33A described later. Then, the fifth multiplier data including the value cosφ · sin θ1 obtained in this way is supplied to the first subtractor 25A.

The first subtractor 25A obtains the value cosφ · sin θ1 included in the fifth multiplier data provided from the fifth multiplier 24A from the value sinφ · cos θ1 included in the fourth multiplier data provided from the fourth multiplier 23A. Subtraction is performed, and subtractor data including the value thus obtained is supplied to the first integrator 26A and the first phase compensator 31A. The value obtained by the first subtracter 25A is expressed by the following equation (4).
sin φ · cos θ1−cos φ · sin θ1 = sin (φ−θ1) (4)

In Formula (4), when the first estimated angle value θ1 of the rotor is sufficiently close to the actual rotation angle φ of the rotor, a value obtained by subtracting the first estimated angle value θ1 from the rotation angle φ (φ−θ1 ) Is sufficiently small, the equation (4) can be approximated as the following equation (5).
sin (φ−θ1) ≈φ−θ1 (5)

  In this embodiment, (φ−θ1) on the right side of Expression (5) may be described as a deviation. In the present embodiment, it is assumed that the first estimated angle value θ1 of the rotor is sufficiently close to the actual rotation angle φ and the approximation of Expression (5) is established, and the first subtractor 25A to the first integrator 26A and the first integrator 26A The subtractor data given to the one phase compensator 31A includes a deviation (φ−θ1).

The first integrator 26A, a value by integrating with respect to the deviation (φ-θ1) included in the subtractor data supplied from the first subtractor 25A time, multiplied by the integral gain value K _I, it was determined in this way The first integrator data including the first estimated angular velocity value Ω1 is supplied to the first adder 32A. In practice the first integrator 26A is a deviation (φ-θ1) to the value K _I · has been multiplied by the integral gain value K _I (φ-θ1), a value obtained by adding the first estimated angular velocity value previously obtained The first estimated angular velocity value Ω1 is calculated. The first phase compensator 31A multiplies the deviation (φ−θ1) included in the subtractor data given from the first subtractor 25A by the proportional gain value K _P and includes compensator data including the value thus obtained. Is supplied to the first adder 32A. Thus, the first integrator 26A and the first phase compensator 31A are configured in parallel.

  The first adder 32A adds the compensator data supplied from the first phase compensator 31A to the first estimated angular velocity value Ω1 included in the first integrator data supplied from the first integrator 26A, and thus The first adder data including the value obtained as described above is supplied to the second integrator 27A. The second integrator 27A integrates the value included in the first adder data given from the first adder 32A with respect to time, and the second integrator including the first estimated angle value θ1 which is the value thus obtained. Is supplied to the fifth sine value generator 33A, the second cosine value generator 34A, the fourth sine value generator 52, and the first cosine value generator 53.

  The fifth sine value generator 33A stores a sine value corresponding to each angle, and calculates a sine sin θ1 corresponding to the first estimated angle value θ1 included in the second integrator data given from the second integrator 27A. The included fifth sine value data is supplied to the fifth multiplier 24A. The second cosine value generator 34A stores a cosine value corresponding to each angle, and calculates a cosine cos θ1 corresponding to the first estimated angle value θ1 included in the second integrator data given from the second integrator 27A. The included second cosine value data is supplied to the fourth multiplier 23A.

  5 includes a first integrator 26A, a first phase compensator 31A, a first adder 32A, and a second integrator 27A in FIG. 4, and an equivalent control system 35 equivalent to a system for obtaining the first estimated angular velocity value Ω1. FIG. The equivalent control system 35 includes an equivalent subtractor 36, a first integrator 26A, a first phase compensator 31A, a first adder 32A, and a second integrator 27A. The first integrator 26A, the first phase compensator 31A, the first adder 32A, and the second integrator 27A are the first integrator 26A, the first phase compensator 31A, the first adder 32A, and the second integrator shown in FIG. The same as the two integrators 27A.

  The equivalent subtractor 36 subtracts the first estimated angle value θ1 given from the second integrator 27A from the rotation angle φ of the rotor given from the outside of the equivalent control system 35, that is, the value thus obtained, The deviation (φ−θ1) is given to the first integrator 26A and the first phase compensator 31A.

  The transfer function Ω (s) / φ (s) of the equivalent control system 35 having the rotation angle φ as an input and the first estimated angular velocity value Ω1 as an output can be expressed by the following equation (6 ).

In Expression (6), s is a Laplace operator, ω _n is a natural angular frequency, and ζ is an attenuation constant. Here, if ω _n ² = K _I and 2 · ζ · ω _n = K _P , the integral gain value K _I of the first integrator 26A is selected by appropriately selecting the natural angular frequency ω _n and the damping constant ζ. And the proportional gain value K _P of the first phase compensator 31A can be determined. And integral gain value K _I, and the proportional gain value K _P, a first estimated angle value .theta.1, the rotor rotational deviation between the angle φ converges to zero faster, and the first estimated angular velocity value Ω1 and rotor The deviation from the rotational angular velocity Ψ is selected so that it quickly converges to zero.

In the present embodiment, the natural angular frequency ω _n may be a value similar to the frequency response of the induction motor 2 to which the rotor is connected, for example. The value similar to this frequency response is a frequency when the gain becomes minus 3 decibels (−3 db) when the frequency response characteristic of the induction motor is drawn on a bode diagram. The attenuation constant ζ is set to a value such that the calculation result of the main controller 4 converges, for example, 1.

  The transfer function of the equivalent control system 35 is a second-order lag element as represented by Expression (6). That is, the transfer function of the rotation information calculation unit 7 is a second-order lag element. The second order lag element does not react excessively to high frequency inputs compared to the first order lag element. Therefore, the rotation information calculation unit 7 constitutes a low-pass filter. As a result, even if high-frequency noise is superimposed while the first analog rotation information is transmitted to the main controller 4 via the second transmission path 9, the rotation information calculation unit 7 does not perform the superimposed high-frequency noise. Can be removed. High-frequency noise superimposed on the first analog rotation information mainly occurs when the transistors constituting the first to third inverter circuits 51A, 51B, 51C are switched from on to off or from off to on. Since the cycle in which the transistor is switched from on to off or from off to on is faster than the cycle in which the rotor of the induction motor 2 rotates, the frequency of the noise mainly superimposed on the first analog rotation information is the first estimated angle. It is higher than the frequency of the value θ1 and the first estimated angular velocity value Ω1. Since the rotation information calculation unit 7 removes high-frequency noise, the first estimated angle value θ1 output from the rotation information calculation unit 7 is a value sufficiently close to the rotation angle φ of the rotor, and the first estimated angular velocity The value Ω1 is sufficiently close to the rotational angular velocity Ψ of the rotor.

  The rotation information calculation unit 7 calculates the first estimated angle value θ1 of the rotor from the first analog rotation information given from the resolver 3 and gives it to the fourth sine value generator 52 and the first cosine value generator 53. .

In the present embodiment, power is supplied to the induction motor 2 so that a value V / F obtained by dividing the amplitude of the primary voltage applied to the induction motor 2 by the frequency of the primary voltage becomes a predetermined value. Controls the rotation state of the rotor. FIG. 6 is a graph showing the relationship between the amplitude of the primary voltage applied to the induction motor 2 and the frequency of the primary voltage. The first V / F calculator 41A calculates the frequency of the primary voltage to be applied to the induction motor 2 from the first inverter circuit 51A based on the angular velocity command value Ω _REF 1 given from the outside of the main controller 4. Next, the first V / F calculator 41 calculates a primary voltage at which a value V / F obtained by dividing the amplitude of the primary voltage by the frequency of the primary voltage from the calculated frequency of the primary voltage becomes a predetermined value. The first V / F data, which is the calculated value, is supplied to the first PWM modulation factor generator 42A. The amplitude of the primary voltage is the amplitude of the primary voltage when the waveform of the primary voltage applied to the induction motor 2 is assumed to be a sine wave. The frequency of the primary voltage is the frequency of the primary voltage when the waveform of the primary voltage is assumed to be a sine wave.

The angle command value calculator 19 calculates an angle command value θ _REF 1 by integrating an angular velocity command value Ω _REF 1 given from the outside of the main control device 4 with respect to time, and calculates the calculated angle command value θ _REF 1 as a first value. 1 to the third sine value generators 43A, 44A and 45A.

The first sine value generator 43A obtains a sine sin θ1 of the angle command value θ _REF 1 given from the angle command value calculator 19, and first multiplies the first sine value data including the value sin θ1 thus obtained. To the vessel 46A. The second sine value generator 44A obtains a sine sin (θ1-2 · π / 3) obtained by subtracting (2 · π / 3) from the angle command value θ _REF 1 given from the angle command value calculator 19. Then, the second sine value data including the value sin (θ1-2 · π / 3) obtained in this way is supplied to the second multiplier 47A. The third sine value generator 45A obtains a sine sin (θ1−4 · π / 3) obtained by subtracting (4 · π / 3) from the angle command value θ _REF 1 given from the angle command value calculator 19. The third sine value data including the value sin (θ1−4 · π / 3) obtained in this way is supplied to the third multiplier 48A.

  The first PWM modulation factor generator 42A obtains the modulation factor η of the voltage applied to the induction motor 2 based on the first V / F data given from the first V / F calculator 41A, and the modulation factor thus obtained. η is supplied to the first to third multipliers 46A, 47A, and 48A.

  The first multiplier 46A multiplies the modulation rate η given from the first PWM modulation rate generator 42A by the value sin θ1 included in the first sine value data given from the first sine value generator 43A, in this way. The first signal wave data including the value η · sin θ1 obtained in this way is applied to the first comparator 54A. The second multiplier 47A includes a modulation rate η given from the first PWM modulation rate generator 42A and a value sin (θ1-2 · π / 3) included in the second sine value data given from the second sine value generator 44A. And the second signal wave data including the value η · sin (θ1-2 · π / 3) obtained in this way is given to the first comparator 54A. The third multiplier 48A includes a value sin (θ1−4 · π / 3) included in the modulation factor η given from the first PWM modulation factor generator 42A and the third sine value data given from the third sine value generator 45A. And the third signal wave data including the value η · sin (θ1−4 · π / 3) obtained in this way is given to the first comparator 54A.

  The first comparator 54A is included in the value η · sin θ1 included in the first signal wave data, the value η · sin (θ1-2 · π / 3) included in the second signal wave data, and the third signal wave data. The value η · sin (θ1−4 · π / 3) is compared with the carrier wave data generated by the timer of the main control device 4 to generate six PWM pulses for driving the first inverter circuit 51A. 1 is given to the inverter circuit 51A.

  FIG. 7 is a diagram schematically showing the configuration of the first inverter circuit 51A. The first inverter circuit 51 </ b> A includes a power supply unit 57 and an inverter unit 58.

The power supply unit 57 is realized by a three-phase diode bridge. The power supply unit 57 rectifies an AC voltage supplied from the outside of the main control device 4 and gives a DC voltage to the inverter unit 58. The inverter unit 58 includes first to sixth insulated gate bipolar transistors (Insulated Gate).
Bipolar Transistor (abbreviated as IGBT) 61, 62, 63, 64, 65, 66, and first to sixth reflux diodes 71, 72, 73, 74, 75, 76.

  Six PWM pulses supplied from the first comparator 54A are input to the gates G of the first to sixth IGBTs 61, 62, 63, 64, 65, 66, respectively.

  The anode of the power supply unit 57, the collector C of the first IGBT 61, the cathode K of the first reflux diode 71, the collector C of the third IGBT 63, the cathode K of the third reflux diode 73, the collector C of the fifth IGBT 65, and the fifth The cathode K of the reflux diode 75 is electrically connected. The cathode of the power supply unit 57, the emitter E of the second IGBT 62, the anode A of the second return diode 72, the emitter E of the fourth IGBT 64, the anode A of the fourth return diode 74, the emitter E of the sixth IGBT 66, and the second The anode A of the reflux diode 72 is electrically connected. The emitter E of the first IGBT 61, the anode A of the first return diode 71, the collector C of the second IGBT 62, and the cathode K of the second return diode 72 are electrically connected and connected to the first U-phase winding. . The emitter E of the third IGBT 63, the anode A of the third return diode 73, the collector C of the fourth IGBT 64, and the cathode K of the fourth return diode 74 are electrically connected and connected to the first V-phase winding. . The emitter E of the fifth IGBT 65, the anode A of the fifth return diode 75, the collector C of the sixth IGBT 66, and the cathode K of the sixth return diode 76 are electrically connected and connected to the first W-phase winding. . The first U-phase winding voltage U1 is applied from the emitter E of the first IGBT 61 to the first U-phase winding, the first V-phase winding voltage V1 is applied from the emitter E of the third IGBT 63 to the first V-phase winding, and the emitter E of the fifth IGBT 65 A first W-phase winding voltage W1 is applied to the first W-phase winding.

  As described above, the first inverter circuit 51A applies the first U-phase winding voltage U1 to the first U-phase winding and the first V-phase winding to the first V-phase based on the PWM pulse supplied from the first comparator 54A. A winding voltage V1 is applied, and a third W-phase winding voltage W1 is applied to the third W-phase winding. As described above, the phase of the first U-phase winding voltage U1, the phase of the first V-phase winding V1, and the phase of the first W-layer winding voltage W1 are different from each other by the first angle Θ1 (rad).

  The fourth sine value generator 52 provides the first D / A converter 55 with the fourth sine value data including the sine sin θ1 of the first estimated angle value θ1 provided from the rotation information calculation unit 7. The first cosine value generator 53 provides the second D / A converter 56 with the first cosine value data including the cosine cos θ1 of the first estimated angle value θ1 given from the rotation information calculation unit 7.

  The first D / A converter 55 is electrically connected to the later-described fourth and sixth A / D converters 14B and 14C via the first transmission path 8. The first D / A converter 55 converts the fourth sine value data including the sine sin θ1 of the first estimated angle value θ1 provided from the fourth sine value generator 52 from a digital value to an analog value, and thus converts it. The second analog sine value data including the calculated value sin θ1 is applied to the fourth and sixth A / D converters 14B and 14C via the first transmission path 8. The second D / A converter 56 is electrically connected to third and fifth A / D converters 15B and 15C, which will be described later, via the first transmission path 8. The second D / A converter 56 converts the first cosine value data including the cosine cos θ1 of the first estimated angle value θ1 supplied from the first cosine value generator 53 from a digital value to an analog value, and thus converts the data. The second analog cosine value data including the measured value cos θ1 is supplied to the third and fifth A / D converters 15B and 15C via the first transmission path 8. In this way, the main control device 4 is given the second analog rotation information representing the rotation state of the rotor to the first sub control device 5 and the second sub control device 6 via the first transmission path 8. The second analog rotation information includes second analog sine value data and second analog cosine value data.

  Since the first sub-control device 5 has the same configuration as that of the main control device 4, the corresponding parts are described by adding the suffix B to the same numerals as the main control device 4. The first sub-control device 5 includes a first θ restoring unit 83B, a second adder 84B, a second V / F calculator 41B, a second PWM modulation factor generator 42B, and a sixth sine value generation realized by executing software. 43B, seventh sine value generator 44B, eighth sine value generator 45B, sixth multiplier 46B, seventh multiplier 47B, eighth multiplier 48B, second comparator 54B, second inverter circuit 51B, A third A / D converter 15B and a fourth A / D converter 14B are included. The first sub-control device 5 receives the second analog sine value data including the sine sin θ1 of the first estimated angle value θ1 and the second analog cosine value including the cosine cos θ1 of the first estimated angle value θ1. Based on the data, the second U-phase winding is given a second U-phase winding voltage U2, the second V-phase winding is given a second V-phase winding voltage V2, and the second W-phase winding is given a second W-phase winding voltage. Give W2.

  The third A / D converter 15B converts the cosine cos θ1 of the first estimated angle value θ1 included in the second analog cosine value data supplied from the second D / A converter 56 from an analog value to a digital value, and thus The second digital cosine value data including the value cos θ1 converted into is supplied to the first θ restoring unit 83B. The fourth A / D converter 14B converts the sine sin θ1 of the first estimated angle value θ1 included in the second analog sine value data supplied from the first D / A converter 55 from an analog value to a digital value. The second digital sine value data including the value sin θ1 converted into is supplied to the first θ restorer 83B.

The first θ reconstructor 83B is based on the second estimated angular velocity based on the sine sin θ1 of the first estimated angle value θ1 and the cosine cos θ1 of the first estimated angle value θ1 provided from the third and fourth A / D converters 14B and 15B. to calculate the value Ω2 as angular velocity command value Omega _REF 2, it calculates a second estimated angle value θ2 as an angle command value theta _REF 2. The first θ restorer 83B gives the calculated second estimated angular velocity value Ω2 to the second V / F calculator 41B, and gives the second estimated angle value θ2 to the second adder 84B.

  FIG. 8 is a block diagram showing the configuration of the first θ restorer 83B. Since the first θ restoring unit 83B has the same configuration as the rotation information calculating unit 7, the corresponding parts are described by adding the suffix B to the same numerals, and redundant descriptions are omitted. The first θ restorer 83B includes a ninth multiplier 23B, a tenth multiplier 24B, a second subtractor 25B, a third integrator 26B, a fourth integrator 27B, and a second phase compensator 31B. A third adder 32B, a ninth sine value generator 33B, and a third cosine value generator 34B are included. The ninth multiplier 23B, the tenth multiplier 24B, the second subtractor 25B, the third integrator 26B, the fourth integrator 27B, the second phase compensator 31B, the third adder 32B, and the ninth sine shown in FIG. The value generator 33B and the third cosine value generator 34B include a fourth multiplier 23A, a fifth multiplier 24A, a first subtractor 25A, a first integrator 26A, a second integrator 27A, a first multiplier shown in FIG. This corresponds to the phase compensator 31A, the first adder 32A, the fifth sine value generator 33A, and the second cosine value generator 34A, respectively. The first θ restoring unit 83B includes the first A / D converter 14A, the second A / D converter 15A, the counter / timing circuit 17, the first synchronous rectifier 21, and the second synchronous rectifier 22 from the configuration of the rotation information calculation unit 7. It is the structure which excluded.

A sine sin θ1 of the first estimated angle value θ1 is given to the ninth multiplier 23B of the first θ restorer 83B, and a cosine cos θ1 of the first estimated angle value θ1 is given to the tenth multiplier 24B of the first θ restorer 83B. third integrator 26B gives to the 2V / F calculator 41B the second estimated angular velocity value Ω2 is an estimate of the rotation speed of the rotor as the angular velocity command value Omega _REF 2, fourth integrator 27B includes a first A second estimated angle value θ2, which is an estimated angle of the rotor, is given to the two adder 84B as an angle command value _θREF2 .

  The second estimated angle value θ2, which is the estimated value of the rotation angle Ψ of the rotor generated by the first θ restorer 83B, is changed to the first estimated angle value θ1 given from the main controller 4 via the first transmission path 8. However, the difference is negligible and hardly affects the control of the induction motor 2.

  Since the first θ restoring unit 83B has the same configuration as the rotation information calculation unit 7 described above, it has a function of a low-pass filter and can remove high-frequency noise. Therefore, the first θ restorer 83B is superimposed when the second analog sine value data and the second analog cosine value data are transmitted from the main control device 4 to the first sub control device 5 via the first transmission path 8. High frequency noise can be removed. The high-frequency noise superimposed on the second analog sine value data and the second analog cosine value data is mainly caused by the transistors constituting the first to third inverter circuits 51A, 51B, and 51C being turned from on to off or from off to on. Occurs when switching to. Since the cycle in which the transistor is switched from on to off or from off to on is faster than the cycle in which the rotor of the induction motor 2 rotates, the second analog sine value data and the second analog cosine value data are mainly used as the main controller. The frequency of noise superimposed when transmitted from 4 to the first sub-control device 5 via the first transmission path 8 is higher than the frequencies of the first estimated angular value θ1 and the first estimated angular velocity value Ω1. Since the first θ restorer 83B can remove high-frequency noise, the second estimated angle value θ2 output from the first θ restorer 83B is a value sufficiently close to the rotation angle φ of the rotor. 2 Estimated angular velocity Ω2 is a value sufficiently close to the rotational angular velocity ψ of the rotor.

Second adder 84B is the second estimated angle value θ2 is an angle command value theta _{REF 2} given from the first 1θ decompressor 83B third angle .THETA.3, i.e. by adding 2 · [pi / 9, thus The second adder data including the added value is supplied to the sixth to eighth sine value generators 43B, 44B, and 45B.

The second V / F calculator 41B performs the same process as the first V / F calculator 41A. The first V / F calculator 41A calculates the first V / F data based on the angular velocity command value Ω _REF 1, whereas the second V / F calculator 41B is the second that has the angular velocity command value Ω _REF 2. Second V / F data is calculated based on the estimated angular velocity value Ω2.

  The second PWM modulation factor generator 42B, the sixth to eighth sine value generators 43B, 44B, 45B, the second comparator 54B, and the sixth to eighth multipliers 46B, 47B, 48B of the first sub-control device 5 are The first PWM modulation factor generator 42A, the first to third sine value generators 43A, 44A, 45A, the first comparator 54A, and the first to third multipliers 46A, 47A, 48A of the main controller 4 described above, The same processing is performed for each.

  The second inverter circuit 51B has the same configuration as the first inverter circuit 51A. The second inverter circuit 51B applies the second U-phase winding voltage U2 to the second U-phase winding and the second V-phase winding to the second V-phase winding based on the six PWM modulation pulses supplied from the second comparator 54B. A line voltage V2 is applied, and a second W-phase winding voltage W2 is applied to the second W-phase winding. The phase of the second U-phase winding voltage U2, the phase of the second V-phase winding V2, and the phase of the second W-layer winding voltage W2 are different from each other by the first angle Θ1 (rad). Further, the values θ2 given from the second adder 84B to the sixth to eighth sine value generators 43B, 44B and 45B, respectively, are given to the first to third sine value generators 43A, 44A and 45A, respectively. Since the third angle Θ3 is larger than the one estimated angle value θ1, the phase of the second U-phase winding voltage U2 advances by the third angle Θ3 (rad) from the phase of the first U-phase winding voltage U1, as described above. .

  Since the second sub-control device 6 has the same configuration as that of the first sub-control device 5, the corresponding parts are described by adding the suffix C to the same numerals. The second sub-control device 6 includes a second θ restoring unit 83C, a fourth adder 84C, a third V / F calculator 41C, a third PWM modulation factor generator 42C, and a tenth sine value generation realized by executing software. 43C, eleventh sine value generator 44C, twelfth sine value generator 45C, eleventh multiplier 46C, twelfth multiplier 47C, thirteenth multiplier 48C, third comparator 54C, third inverter circuit 51C, A fifth A / D converter 15C and a sixth A / D converter 14C are included. The second sub-control device 6 receives the second analog sine value data including the sine sin θ1 of the first estimated angle value θ1 and the second analog cosine value including the cosine cos θ1 of the first estimated angle value θ1. Based on the data, a third U-phase winding voltage U3 is applied to the third U-phase winding, a third V-phase winding voltage V3 is applied to the third V-phase winding, and a third W-phase winding voltage is applied to the third W-phase winding. Give W3. The second sub-control device 6 has the same configuration as the first sub-control device 5 described above, and a duplicate description is omitted.

  The 5th A / D converter 15C, 6th A / D converter 14C, 2nd theta restorer 83C, the 4th adder 84C, the 3rd V / F calculator 41C, the 3rd PWM modulation rate generator of the 2nd sub-control device 6 42C, a tenth sine value generator 43C, an eleventh sine value generator 44C, a twelfth sine value generator 45C, an eleventh multiplier 46C, a twelfth multiplier 47C, a thirteenth multiplier 48C, a third comparator 54C, and The third inverter circuit 51C includes a third A / D converter 15B, a fourth A / D converter 14B, a first θ restorer 83B, a second adder 84B, a second V / F calculator 41B, A second PWM modulation factor generator 42B, a sixth sine value generator 43B, a seventh sine value generator 44B, an eighth sine value generator 45B, a sixth multiplier 46B, a seventh multiplier 47B, an eighth multiplier 48B, Second comparator 54B and second inverter Each corresponds to the circuit 51B.

The fourth adder 84C of the second sub-control device 6 corresponding to the second adder 84B of the first sub-control device 5 is the second estimated angle value that is the angle command value θ _REF 2 given from the second θ restorer 83C. Fourth adder data obtained by adding θ2 and the negative value of the third angle Θ3, that is, (−2 · π / 9), is supplied to the tenth to twelfth sine value generators 43C, 44C, and 45C, respectively.

  The third inverter circuit 51C applies the third U-phase winding voltage U3 to the third U-phase winding and the third V-phase winding voltage to the third V-phase winding based on the PWM modulation pulse supplied from the third comparator 54C. V3 is applied, and the third W-phase winding voltage W3 is applied to the third W-phase winding. The phase of the third U-phase winding voltage U3, the phase of the third V-phase winding V3, and the phase of the third W-layer winding voltage W3 are different from each other by the first angle Θ1 (rad). Further, the values θ2 given from the fourth adder 84C to the tenth to twelfth sine value generators 43C, 44C, and 45C, respectively, are given to the first to third sine value generators 43A, 44A, and 45A, respectively. Since it is 2 · π / 9 smaller than one estimated angle value θ1 and 4 · π / 9 smaller than the value θ2 included in the second adder data given to the sixth to eighth sine value generators 43B, 44B, and 45B. As described above, the phase of the third U-phase winding voltage U3 is delayed by a third angle Θ3 (rad), that is, 2 · π / 9 (rad), from the phase of the first U-phase winding voltage U1, and the second U-phase It is delayed from the phase of the winding voltage U2 by twice the third angle Θ3, that is, 4 · π / 9 (rad).

  The second analog / digital converter is realized by including a third A / D converter 15B, a fourth A / D converter 14B, a fifth A / D converter 15C, and a sixth A / D converter 14C. The second inverter circuit that supplies power to the induction motor 2 in synchronization with the first inverter circuit 51A is realized including the second and third inverter circuits 51B and 51C. The second inverter drive unit that drives the second and third inverter circuits 51B and 51C in synchronization with the first inverter circuit 51A includes first and second θ restorers 83B and 83C, and second and third V / F operations. 41B, 41C, second and third PWM modulation rate generators 42B, 42C, sixth and tenth sine value generators 43B, 43C, seventh and eleventh sine value generators 44B, 44C, and eighth And twelfth sine value generators 45B and 45C, sixth and eleventh multipliers 46B and 46C, seventh and twelfth multipliers 47B and 47C, eighth and thirteenth multipliers 48B and 48C, second and This is realized by including third comparators 54B and 54C and second and fourth adders 84B and 84C. The estimated angle value calculation unit and the estimated angular velocity value calculation unit are realized by the first and second θ reconstructors 83B and 83C.

  As described above, in order for the first and second sub-control devices 5 and 6 to supply electric power for driving the induction motor 2 in synchronization with the main control device 4, the first transmission is performed from the main control device 4. The second analog sine value data and the second analog cosine value data of analog values are given to the first and second sub-control devices 5 and 6 via the path 8. The first and second sub-control devices 5 and 6 supply electric power for driving the induction motor 2 in synchronization with the main control device 4 based on the second analog sine value data and the second analog cosine value data. To do.

When the signals given from the main control device 4 to the first and second sub control devices 5 and 6 are expressed as digital values, the signals are affected by noise while passing through the first transmission path 8. Can change significantly. For example, the first and second sub-control devices 5 and 6 receive information obtained by inverting one bit of a signal represented by a digital value under the influence of noise while passing through the first transmission path 8. There is a case. Even if only one bit of the signal represented by the digital value is inverted, if the inverted 1 bit is a sign bit representing a sign, the signal with the inverted sign is the first and second sub-control devices. 5 and 6 will be given. In this case, since the first and second sub-control devices 5 and 6 supply electric power for driving the induction motor 2 based on the information whose sign is inverted, the rotor of the induction motor 2 is rotated in advance. There is a possibility that it is greatly out of the state. In the present embodiment, since the second analog sine value data and the second analog cosine value data are represented by analog values, even if the second analog sine value data and the second analog cosine value data change due to noise while passing through the transmission line, only the change is regarded as noise. The first and second sub-control devices 5 and 6 are given. Therefore, the second analog sine value data and the second analog cosine value data are less likely to undergo a large change while passing through the first transmission line 8 as compared to the case where the second analog sine value data and the second analog cosine value data are represented by digital values. Based on the second analog sine value data and the second analog cosine value data, the first and second sub-control devices 5 and 6 supply power to the induction motor 2 in synchronization with the main control device 4. As a result, the main control device 4, the first sub control device 5 and the second sub control device 6 cause the rotor of the induction motor 2 to accurately rotate in accordance with the angular velocity command value Ω _REF 1. Can be controlled.

Further, the main controller 4 controls the first and second sub-controllers 5 and 6 with the second analog sine value data and the second analog value by simple communication control without providing complicated control such as an error detection function. Gives analog cosine value data. Therefore, the amount to be processed by the main control device 4, the first sub control device 5, and the second sub control device 6 can be suppressed as compared with the case where complicated control is provided for communication control. Thus, when the angular velocity command value Ω _REF 1 is given to the main control device 4, electric power reflecting the angular velocity command value Ω _REF 1 can be quickly given to the induction motor 2.

Further, as described above, since the rotation information calculation unit 7 constitutes a high-pass filter, the rotation information calculation unit 7 outputs the first estimated angle value θ1 sufficiently close to the rotation angle φ of the rotor, and is sufficient for the rotation angular velocity Ψ of the rotor. A near first estimated angular velocity value Ω1 can be output. In addition, since the first restorer 83B and the second restorer 83B constitute a high-pass filter, the second estimated angle value θ2 that is sufficiently close to the rotation angle of the rotor is output as the angle command value _θREF2 , the second estimated angular velocity value Ω2 close enough to the rotational angular velocity Ψ of the rotor can be output as an angular velocity instruction value Omega _{REF 2.} The first and second sub-control devices 5 and 6 control the induction motor 2 based on the second estimated angular velocity value θ2 sufficiently close to the rotational angle φ of the rotor and the second estimated angular velocity value Ω2 sufficiently close to the angular velocity ψ of the rotor. Supply power to drive. As a result, the main control device 4, the first and second sub control devices 5, 6 control the induction motor 2 so that the rotor of the induction motor 2 accurately rotates in accordance with the angular velocity command value Ω _REF 1. can do.

  FIG. 9 is a block diagram showing configurations of the control device 85, the induction motor 2, the resolver 3, and the driver 18 according to another embodiment of the present invention. Since the control device 85 of the present invention has the same configuration as the control device 1 of the above-described embodiment, the corresponding components are denoted by the same reference numerals and description thereof is omitted.

  The control device 85 of the present embodiment is different in the configuration of the main control device 4 in the control device 1 of the above-described embodiment, and performs slip frequency control.

  The main controller 86 includes a rotation information calculator 87, an angle command value calculator 88, a slip frequency controller 89, a first V / F calculator 90, a first PWM modulation factor generator 42A, which are realized by executing software. First to third sine value generators 43A, 44A, 45A, first to third multipliers 46A, 47A, 48A, a first comparator 54A, a fourth sine value generator 52, and a first cosine value generator 53, A first inverter circuit 51A, a first D / A converter 55, and a second D / A converter 56 are included.

  The rotation information calculation unit 87 functions as a first analog / digital conversion unit that converts the first analog rotation information into first digital rotation information represented by a digital value. Angle command value calculator 88, slip frequency controller 89, first V / F calculator 90, first PWM modulation factor generator 42A, first to third sine value generators 43A, 44A, 45A, first to third multiplications The units 46A, 47A, 48A, and the first comparator 54A function as a first inverter driving unit that drives the first inverter circuit based on the first digital rotation information and the slip frequency control. The angle command value calculator 88, the slip frequency control unit 89, the fourth sine value generator 52, the first cosine value generator 53, the first inverter circuit 51A, the first D / A converter 55, and the second D / A converter. 56 functions as a digital / analog conversion unit that generates second analog rotation information based on the first digital rotation information and slip frequency control, and supplies the second analog rotation information to the second inverter driving means via the transmission line. To do.

  The rotation information calculation unit 87 calculates the first estimated angular velocity value Ω1 and gives the first integrator data including the first estimated angular velocity value Ω1 to the slip frequency control unit 89.

FIG. 10 is a block diagram showing the configuration of the slip frequency control unit 89. The slip frequency control unit 89 performs PI control on the induction motor 2 to be controlled. The slip frequency control unit 89 calculates the angular velocity command value Ω _REF 1 based on the angular velocity target value Ω _CMD and the first estimated angular velocity value Ω 1 given from the outside of the main controller 86, and the first V / F calculator 90 and An angle command value calculator 88 is provided. The angular velocity target value _ΩCMD is a target value of the angular velocity of the rotor to be controlled.

The slip frequency control unit 89 includes a third subtracter 91, a proportional device 92, a fifth integrator 93, a fifth adder 94, and a sixth adder 95. The third subtractor 91 calculates the angular deviation from the angular velocity target value Omega _CMD by subtracting the first estimated angular velocity value .OMEGA.1, this way proportionally 92 a deviation data including the angular velocity deviation value determined and fifth This is supplied to the integrator 93.

Proportional unit 92, the angular velocity deviation values contained in the difference data supplied from the third subtracter 91 is multiplied by a proportional gain value Kf _P, proportional data including a proportional value calculated in this way in the fifth adder 94 give. The fifth integrator 93 integrates the angular velocity deviation value included in the deviation data given from the third subtractor 91 with respect to time, multiplies the integral gain Kf _I, and the integral data including the integral value thus obtained. Is supplied to the fifth adder 94. The fifth adder 94 adds the proportional value and the integral value included in the proportional data and the integral data given from the proportional unit 92 and the fifth integrator 93, and gives the added value to the sixth adder 95 as addition data. . The sixth adder 95 calculates the angular velocity command value Ω _REF 1 by adding the addition value included in the addition data supplied from the fifth adder 94 and the angular velocity target value _ΩCMD, and calculates the calculated angular velocity command value. Angular velocity command data including Ω _REF 1 is _supplied to the angle command value calculator 88 and the first V / F calculator 90.

The angle command value calculator 88 calculates the angle command value θ _REF 1 by integrating the angular velocity command value Ω _REF 1 given from the slip frequency control unit 89 with respect to time. The angle command value calculator 88 supplies the calculated angle command value θ _REF 1 to the first to third sine value generators 43A, 44A, 45A, the fourth sine value generator 52, and the first cosine value generator 53.

The fourth sine value generator 52, the first cosine value generator 53, the first D / A converter 55, and the second D / A converter 56 represent the angle command value θ _REF 1 represented by a digital value as an analog value. The second analog rotation information including the angle command value θ _REF 1 is given to the first and second sub-control devices 5 and 6 by converting into the second analog rotation information.

The first and second sub control devices 5 and 6 are connected to the inverter circuits 51B and 51C in synchronization with the main control device 86 based on the second analog rotation information including the angle command value θ _REF 1 given from the main control device 86. To control. Since the angle command value θ _REF 1 is a value calculated based on the slip frequency control, the main controller 86, the first and second sub-controllers 5, 6 synchronize with each other with respect to the induction motor 2. Take control.

According to the control device 85 described above, the second analog rotation information including the angle command value θ _REF 1 is accurately transmitted from the main control device 86 to the first and second sub control devices 5 and 6 as described above. Can do. As a result, the main control device 86, the first and second sub control devices 5, 6 can perform slip frequency control on the induction motor 2 in precise synchronization, and the control device 85 can control the angular velocity shown in FIG. the angular velocity estimate Ω1 shown in FIG. 10 to the target value Omega _CMD can be accurately follow.

  In the present embodiment, the resolver 3 is a one-phase excitation two-phase output type resolver. However, the present invention is not limited to this. For example, a brushless type resolver in which a slip ring and a brush are replaced with a rotary transformer in order to pass a current through a winding of a resolver rotor may be used. A resolver called a reluctance type resolver having a configuration in which no winding is wound around the rotor may be used. That is, the resolver 3 may be a resolver that has an electrical characteristic that is equivalent to that of the one-phase excitation two-phase output method.

  In the present embodiment, the number of control devices that supply power to the induction motor 2 in synchronization with the main control device 4 is two, that is, the first and second sub-control devices 5 and 6. The number of control devices that supply power to the induction motor 2 in synchronization with the device 4 is not limited to two, and may be one or three or more.

  In the present embodiment, the induction motor 2 is a nine-phase induction motor. However, the number of phases of the induction motor is not limited to nine and may be three or more. The induction motor 2 having three or more phases may be driven in synchronization by the main control device and the sub control device. Further, although the control device 1 drives the induction motor 2, the motor driven by the control device 1 is not limited to the induction motor, and may be a synchronous motor.

It is a block diagram which shows the structure of the control apparatus 1, the induction motor 2, the resolver 3, and the driver 18 of one Embodiment of this invention. It is the figure which displayed the voltage applied to each winding when the phase of the 1st U phase winding voltage U1 is 0 (rad) by phasor display. 2 is a diagram schematically showing a configuration of a resolver 3. FIG. 3 is a block diagram showing a configuration of a rotation information calculation unit 7. FIG. FIG. 4 is a block diagram showing an equivalent control system 35 that is equivalent to a system that obtains the first estimated angular velocity value Ω1, including the first integrator 26A, the first phase compensator 31A, the first adder 32A, and the second integrator 27A in FIG. It is. 4 is a graph showing the relationship between the amplitude of the primary voltage applied to the induction motor 2 and the frequency of the primary voltage. It is a figure which shows typically the structure of 51 A of 1st inverter circuits. It is a block diagram which shows the structure of 1st (theta) restorer 83B. It is a block diagram which shows the structure of the control apparatus 85, the induction motor 2, the resolver 3, and the driver 18 of other embodiment of this invention. 3 is a block diagram showing a configuration of a slip frequency control unit 89. FIG.

Explanation of symbols

DESCRIPTION OF SYMBOLS 1,85 Control apparatus 2 Induction motor 3 Resolver 4,86 Main control apparatus 5 1st sub control apparatus 6 2nd sub control apparatus 7,87 Rotation information calculation part 14A 1st analog / digital converter 14B 4th analog / digital conversion 14C 6th analog / digital converter 15A 2nd analog / digital converter 15B 3rd analog / digital converter 15C 5th analog / digital converter 19 Angle command calculator calculator 51A 1st inverter circuit 51B 2nd inverter circuit 51C Third inverter circuit 54A First comparator 54B Second comparator 54C Third comparator 83B First θ restorer 84B Second θ restorer

Claims (4)

First inverter driving means for driving a first inverter circuit for supplying electric power to the electric motor, provided with first analog rotation information representing the rotational state as an analog value from a rotation information acquiring means for acquiring the rotational state of the rotor of the electric motor. And a transmission path for transmitting second analog rotation information, which is output from the first inverter drive means and represents a command for putting the rotor in a preset rotation state as an analog value, and the first inverter via the transmission path Second inverter driving means for driving a second inverter circuit for supplying electric power to the electric motor based on second analog rotation information given from the driving means,
The first inverter drive means
A first analog / digital converter that converts the first analog rotation information into first digital rotation information represented by a digital value;
A first inverter drive unit that drives the first inverter circuit so that the rotor is in a preset rotation state;
A digital / analog conversion unit that generates the second analog rotation information based on the first digital rotation information and supplies the second analog rotation information to the second inverter driving means via the transmission line;
The second inverter driving means is
A second analog / digital converter that converts the second analog rotation information given through the transmission path into second digital rotation information represented by a digital value;
And a second inverter driving unit that drives the second inverter circuit in synchronization with the first inverter circuit so that the rotor is in a preset rotation state based on the second digital rotation information. Inverter drive device. The first inverter driving unit drives the first inverter circuit based on first digital rotation information and slip frequency control,
2. The inverter driving apparatus according to claim 1, wherein the digital / analog converter generates the second analog rotation information based on first digital rotation information and slip frequency control. The second inverter driving means is
An estimated angle value calculation unit for obtaining an estimated angle value from the second digital rotation information using a second-order lag element;
An estimated angular velocity value calculation unit for obtaining an estimated angular velocity value from the second digital rotation information using a second-order lag element;
The inverter driving device according to claim 1 or 2, wherein the second inverter driving unit drives the second inverter circuit based on the estimated angle value and the estimated angular velocity value. The first analog rotation information representing the rotation state as an analog value is given from the rotation information acquisition means for acquiring the rotation state of the rotor of the electric motor, and the first inverter driving means drives the first inverter circuit that supplies electric power to the motor. Then, based on the second analog rotation information representing an instruction for turning the rotor given from the first inverter driving means through the transmission path to a preset rotation state by an analog value, the electric motor is driven by the second inverter driving means. An inverter driving method for driving a second inverter circuit that supplies electric power to
The first inverter drive means
Converting the first analog rotation information into first digital rotation information represented by a digital value;
Driving the first inverter circuit so that the rotor is in a preset rotation state;
Based on the first digital rotation information, the second analog rotation information is generated, and the second analog rotation information is provided to the second inverter driving means via the transmission path,
The second inverter driving means is
Converting the second analog rotation information given through the transmission path into second digital rotation information represented by a digital value;
An inverter driving method, wherein the second inverter circuit is driven in synchronization with the first inverter circuit so that the rotor is in a preset rotation state based on the second digital rotation information.

Images