JP2007259658A - Control circuit for dc-dc converter, the dc-dc converter and control method for the dc-dc converter - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide a synchronous rectification type DC-DC converter, capable of improving the control voltage value of output voltage, when load is low, and to provide a control circuit for a DC-DC converter capable of improving the power conversion efficiency. <P>SOLUTION: An averaging circuit 12G averages out a coil current IL1. A voltage comparator COMP compares the averaged out average current AI, with a predetermined current value that has been set previously. When a comparison result, with the average current AI of the coil current IL1 being smaller than the predetermined current value, is obtained, a conduction control circuit 17G controls a transistor FET2 to be non-conducting, according to the detection result by a backflow detection circuit 16G, thus preventing generation of backflow current and increasing the oscillation period of an oscillator OSC to be large, according to the output signal of the voltage comparator COMP. Thus, PWM fixed control is switched to PFM control for stretching the cycle period of an operation cycle, according to the amount of load power. <P>COPYRIGHT: (C)2008,JPO&INPIT

Description

  The present invention relates to control of a DC-DC converter, and more particularly, to control at a light load in a synchronous rectification DC-DC converter.

  In portable electronic devices, a battery is mounted as a power source for the device. In general, since the voltage of the battery decreases as the discharge progresses, in order to keep the voltage used inside the electronic device constant. A DC-DC converter is used to make the battery output constant voltage. Since the efficiency of DC-DC used in portable devices affects the operating time of the battery, high efficiency is required in all areas from heavy loads to light loads.

  There has been proposed a DC-DC converter that switches and controls an operation method between a PFM (Pulse Frequency Modulation) method and a PWM (Pulse Width Modulation) method according to a load. The technique disclosed in Patent Document 1 is an example. In a normal operation state including a heavy load, the PWM method is operated, and the PFM method is switched in a light load. Here, the coil current has a sawtooth waveform. The current supplied to the load is the average value of the coil current. When the load is low, the bottom value of the sawtooth waveform of the coil current may be negative, the current direction may be reversed, and the current may flow backward from the load toward the coil. Therefore, when the occurrence of backflow is detected, the backflow is prevented by maintaining the synchronous rectifying element in a non-conductive state. At this time, since the coil current is discontinuous, a so-called discontinuous current mode (DCM) is set.

  In the discontinuous current mode, switching operation is performed at a predetermined cycle by PWM control, and if excessive power is discharged every cycle toward the load side, the output voltage rises in the case of a light load, and control is impossible. There was a problem. Therefore, in the discontinuous current mode, PFM control is performed to limit the increase of the output voltage by limiting the switching of the main switching element. That is, when switching from the continuous current mode (CCM) to the discontinuous current mode (that is, when the bottom value of the coil current becomes negative), it is necessary to always switch from PWM control to PFM control.

  FIG. 7 shows a DC-DC converter 100 according to Prior Literature 2. In the DC-DC converter 100, PWM fixed control is performed in which the switching operation is performed at a fixed period regardless of the load power amount. The DC-DC converter 100 has a configuration in which excess power on the load side is returned every cycle by allowing a reverse flow of the coil current IL1. When the PWM fixed control is performed, the selection signal DSA is set to the high level, and the output terminal of the OR gate circuit OR101 is fixed to the high level regardless of the output voltage of the comparator COMP103. Therefore, regardless of the reverse flow of the coil current IL1, the conduction state of the transistor FET102, which is a synchronous rectifier, is maintained. That is, even if a switching operation is performed in a predetermined cycle by the PWM method and excessive power is discharged every cycle toward the load side, the excess power can be regenerated to the input side every cycle. Therefore, an increase in output voltage can be prevented.

  Therefore, the DC-DC converter 100 can prevent the output voltage from rising even when the bottom value of the sawtooth waveform of the coil current IL1 becomes negative. Therefore, when the bottom value of the coil current IL1 is negative, it is not necessary to switch to the PFM control, and the PWM control state can be maintained. Therefore, even when the power consumption of the load is very low and the bottom value of the coil current IL1 is negative, the DC-DC converter can be operated by PWM control to supply power to the load.

In addition, the prior document 3 is disclosed.
JP-A-6-303766 JP 2006-14482 A JP-A-8-340675

  However, when the load is further reduced during PWM fixed control (such as when the load is stopped), the ratio of the fixed loss (such as switching operation) of the DC-DC converter 100 to the total power consumption becomes significant. . Therefore, in order to reduce the fixed loss and increase the efficiency, it is necessary to switch the DC-DC converter 100 from the PWM fixed control to the PFM control to reduce the operating frequency when the load is stopped.

  Conventionally, when switching from the continuous current mode to the discontinuous current mode (that is, when the bottom value of the coil current IL1 becomes negative), it is necessary to switch from PWM control to PFM control. However, according to the prior art document 2, since PWM control is possible even in a range where the bottom value of the coil current IL1 is negative, it is necessary to switch between PWM fixed control and PFM control in this range. However, there is no specific disclosure about a method for enabling such switching, which is a problem.

  Conventionally, switching from PWM control to PFM control can be performed using generation of a backflow current as a trigger. However, in the prior art document 2, since the PWM control is performed even when the reverse current is generated, the generation of the reverse current cannot be used as a switching trigger. Then, a new trigger for switching from PWM control to PFM control is necessary, but there is no specific disclosure about the trigger, which is a problem.

  The present invention has been made in order to solve at least one of the problems of the background art described above, and in a synchronous rectification DC-DC converter, while improving the control voltage value of the output voltage at light load, It is an object of the present invention to provide a DC-DC converter control circuit, a DC-DC converter, and a control method thereof that can improve power conversion efficiency.

  In order to achieve the above object, a control circuit for a DC-DC converter according to a first aspect of the present invention includes a first switching element that conducts when accumulating electric power in an inductive element, and a load of electric power accumulated in the inductive element. Control of a DC-DC converter including a second switching element that is switched in accordance with the discharge period of the power supply, and a PWM fixed control in which a repetition cycle of an operation cycle is a predetermined cycle regardless of load power amount, and a light load A DC-DC converter control circuit that enables selection between PFM control that expands and contracts the cycle of an operation cycle in accordance with load electric energy, and that measures an inductive element current flowing through the inductive element A comparison circuit for comparing the inductive element current with a predetermined current value, and monitoring a reverse current caused by conduction of the second switching element, When the comparison result of the reverse current detection circuit for detecting inversion and the comparison circuit is input and the inductive element current is less than a predetermined current value, the second switching element is made non-conductive according to the detection result of the reverse current detection circuit, and the inductive element When the current is greater than the predetermined current value, the conduction control circuit that maintains the conduction state of the second switching element until the accumulation of power in the induction element is started regardless of the detection result, and the repetition cycle of the operation cycle And an oscillation circuit that changes an oscillation cycle in accordance with a measurement result by a current measurement circuit or a comparison result by a comparison circuit.

  The DC-DC converter according to the first aspect of the present invention is switched according to a first switching element that conducts when accumulating electric power in the inductive element and a discharge period of the electric power accumulated in the inductive element to the load. The control of the DC-DC converter including the second switching element that conducts is performed by PWM fixed control in which the repetition cycle of the operation cycle is a predetermined cycle regardless of the load power amount, and in the light load, the repetition cycle of the operation cycle is determined by the load power amount. A DC-DC converter that is selectable between PFM control that expands and contracts in accordance with the current measurement circuit that measures the inductive element current flowing through the inductive element, and a predetermined current value that determines the inductive element current in advance. A comparison circuit for comparison, a reverse current detection circuit for monitoring reverse current due to conduction of the second switching element and detecting inversion of the current direction, and a comparison circuit When the comparison result is input and the inductive element current is smaller than the predetermined current value, the second switching element is made nonconductive according to the detection result of the backflow detection circuit, and when the inductive element current is larger than the predetermined current value, the detection result Regardless of whether or not power storage in the inductive element is started, a conduction control circuit that maintains the conduction state of the second switching element and an oscillator that determines a repetition period of the operation cycle, the measurement result by the current measurement circuit Or an oscillation circuit that changes an oscillation cycle in accordance with a comparison result by the comparison circuit.

  The DC-DC converter control method according to the first aspect of the invention is switched according to the first flow path that is conducted when power is stored in the inductive element, and the discharge period of the power stored in the inductive element to the load. The control of the DC-DC converter including the second flow path that is controlled and conducted is PWM fixed control in which the repetition cycle of the operation cycle is a predetermined cycle regardless of the amount of load power, and the repetition cycle of the operation cycle in a light load. A method for controlling a DC-DC converter that can be selected between PFM control that expands and contracts in accordance with load electric energy, the step of measuring an inductive element current flowing through the inductive element, and the inductive element current being predetermined Comparison between the step of comparing with a predetermined current value, the step of monitoring the backflow current due to the conduction of the second flow path and detecting the reversal of the current direction, and the step of comparing When the result is input and the inductive element current is smaller than the predetermined current value, the second flow path is made non-conductive according to the detection result of the backflow detection circuit, and when the inductive element current is larger than the predetermined current value, the detection result is Regardless of whether or not power storage in the inductive element is started, the step of maintaining the conduction state of the second flow path and the oscillator that determines the repetition period of the operation cycle, the measurement result of the measurement step, or And a step of changing an oscillation period according to a comparison result of the comparing step.

  A control circuit for a DC-DC converter or a DC-DC converter according to the present invention includes a first switching element and a second switching element. The DC-DC converter control method according to the present invention includes a first distribution path and a second distribution path. In the DC-DC converter control circuit, the DC-DC converter, and the DC-DC converter control method according to the first aspect of the present invention, the second switching element is selected according to the discharge period of the power accumulated in the inductive element to the load. Alternatively, a so-called synchronous rectification operation is performed by conducting the switching of the second distribution path. In the DC-DC converter control circuit, the DC-DC converter, and the DC-DC converter control method according to the first invention, the control method can be selected between PWM fixed control and PFM control. The PWM fixed control is control that fixes the repetition cycle of the operation cycle to a predetermined cycle regardless of the load power amount. Moreover, PFM control is control which expands / contracts the repetition period of an operation cycle according to load electric energy.

  A sawtooth inductive element current flows through the dielectric element. The current supplied to the load is an average value of the inductive element current. Then, when the load current consumed in the load decreases to reach a light load state, the bottom value of the sawtooth waveform of the inductive element current may become negative. Then, the current direction is reversed, and the current flows backward from the load toward the inductive element. The backflow detection circuit monitors the backflow current and detects reversal of the current direction.

  The step of measuring the current measuring circuit or the inductive element current measures the inductive element current. The comparison circuit or the comparing step compares the inductive element current with a predetermined current value. Here, the predetermined current value is appropriately determined according to the power consumption of the load connected to the DC-DC converter. For example, if the predetermined current value is set to an inductive element current value necessary for supplying the minimum required power consumption of the load, the load is in an operating state when the inductive element current is larger than the predetermined current value. When the current value is smaller than the predetermined current value, it can be detected that the load is stopped. The predetermined current value can be set to a value equal to or less than the average value of the inductive element current when the bottom value of the sawtooth waveform is zero.

  The conduction control circuit is a circuit that controls the conduction / non-conduction state of the second switching element. The step of controlling the conduction state controls the conduction / non-conduction state of the second flow path. The conduction control circuit or the controlling step operates according to the comparison result of the comparison circuit and the detection result of the backflow detection circuit. The oscillation circuit is a circuit that determines the repetition period of the operation cycle of the DC-DC converter. The oscillation circuit is a circuit that performs an operation of variably controlling the oscillation cycle in accordance with the measurement result by the current measurement circuit or the comparison result by the comparison circuit. Further, the oscillation cycle of the oscillation circuit is variably controlled by the step of changing the oscillation cycle.

  When a comparison result that the inductive element current is larger than the predetermined current value is obtained by the comparison circuit or the comparison step, the load is in a state where the power of the DC-DC converter is required. Then, regardless of the detection result of the backflow detection circuit or the detecting step, the second switching element or the second flow is controlled by the step of controlling the conduction control circuit or the conduction state until the accumulation of power in the inductive element is started. The conduction state of the path is maintained. The oscillation circuit outputs an oscillation signal having a predetermined oscillation period. Thereby, PWM fixed control which operates at a fixed frequency determined by the oscillation circuit is performed.

  In the PWM fixed control, even when the current bottom value is negative and the current direction is reversed and the current flows from the load to the inductive element, the load-side power moves and accumulates in the inductive element. It will be. That is, even if a switching operation is performed at a predetermined cycle by the PWM method and excessive power is released every cycle toward the load side, the excess power can be returned to the inductive element every cycle. Therefore, an increase in output voltage is suppressed.

  On the other hand, when the comparison circuit or the comparing step gives a comparison result that the inductive element current is smaller than the predetermined current value, the load is in a stopped state, etc., and the load does not consume the minimum necessary power It is. In this case, the backflow current is prevented from being generated by turning off the second switching element or the second flow path in accordance with the detection result of the backflow detection circuit or the detecting step. Then, the oscillation period of the oscillation circuit is increased according to the measurement result by the current measurement circuit or the comparison result by the comparison circuit. Alternatively, the oscillation period of the oscillation circuit is increased according to the measurement result of the measuring step or the comparison result of the comparing step. As a result, the PWM fixed control is switched to the PFM control that expands and contracts the repetition cycle of the operation cycle according to the load power amount. Since the fixed loss can be reduced by PFM control, the efficiency of the DC-DC converter can be increased.

  As described above, it is possible to switch from PWM control to PFM control even in a region where the bottom value of the sawtooth waveform of the inductive element current reaches a negative value. Thereby, even when the power supplied to the load is very low and the bottom value of the inductive element current becomes negative, the power can be supplied by PWM control. That is, low power can be supplied while keeping the operating frequency of the DC-DC converter constant. Therefore, it is possible to supply power to a load that requires low power supply and is easily affected by fluctuations in the operating frequency of the DC-DC converter.

  When power supply is not required, the PWM fixed control can be switched to the PFM control even in a region where the bottom value of the sawtooth waveform of the inductive element current reaches a negative value. As a result, the fixed loss can be reduced by the PFM control, so that the efficiency of the DC-DC converter can be increased.

  According to the present invention, between the PWM fixed control in which the repetition cycle of the operation cycle is a predetermined cycle regardless of the load electric energy, and the PFM control in which the repetition cycle of the operation cycle is expanded or contracted according to the load electric energy in a light load. Thus, it is possible to provide a control circuit for a DC-DC converter, a DC-DC converter, and a control method therefor that can select a control method. At this time, it is possible to switch between PWM fixed control and PFM control even in a range where the bottom value of the inductive element current is in a negative state.

  A principle diagram of the present invention is shown in FIG. The DC-DC converter 1G includes a control circuit 11G, transistors FET1 and FET2, and a choke coil L1. The control circuit 11G includes a backflow detection circuit 16G, an averaging circuit 12G, a voltage comparator COMP, an oscillator OSC, and a conduction control circuit 17G. The input voltage Vin is input to the drain terminal of the transistor FET1. The source terminal of the transistor FET1 is connected to one terminal of the choke coil L1 and the drain terminal of the transistor FET2. The source terminal of the transistor FET2 is connected to the ground potential. The gate terminals of the transistors FET1 and FET2 are connected to the conduction control circuit 17G. An output voltage Vout is output from the other terminal of the choke coil L1, and the output voltage Vout is supplied to a load (not shown).

  The configuration of the control circuit 11G will be described. The output terminal of the choke coil L1 is connected to the averaging circuit 12G and the backflow detection circuit 16G. The output terminal of the averaging circuit 12G is connected to the voltage comparator COMP. Note that the output terminal of the averaging circuit 12G may be connected to the oscillator OSC. The output terminal of the voltage comparator COMP is connected to the conduction control circuit 17G and the oscillator OSC. The output terminal of the backflow detection circuit 16G is connected to the conduction control circuit 17G.

  The operation of the DC-DC converter 1G will be described. In the DC-DC converter 1G, the transistor FET2 is subjected to switching control in accordance with the discharge period of the electric power accumulated in the choke coil L1 to conduct, whereby the voltage between the terminals of the choke coil L1 flows in an inverted manner. A so-called synchronous rectification operation is performed. In the DC-DC converter 1G according to the present invention, a control method can be selected between the PWM fixed control and the PFM control. The PWM fixed control is control that fixes the repetition cycle of the operation cycle to a predetermined cycle regardless of the load power amount. Moreover, PFM control is control which expands / contracts the repetition period of an operation cycle according to load electric energy.

  A sawtooth coil current IL1 flows through the choke coil L1. The load is supplied with an average current of the coil current IL1. Here, when the load becomes a light load state and the load current consumed in the load decreases, the bottom value of the sawtooth waveform of the coil current IL1 may become negative. Then, in the range where the bottom value is negative, the current direction is reversed, and the current flows backward from the load toward the choke coil L1. The backflow detection circuit 16G monitors the backflow current and detects reversal of the current direction.

  The averaging circuit 12G averages the coil current IL1. The voltage comparator COMP compares the averaged average current AI with a predetermined current value. Here, the predetermined current value is appropriately determined according to the power consumption of the load connected to the DC-DC converter 1G. For example, if the predetermined current value is set to the average current AI necessary for supplying the lower limit value of the power consumption of the load, the operating state and the stopped state of the load can be detected by the voltage comparator COMP. The value of the predetermined current value can be equal to or less than the average value of the coil current IL1 in which the bottom value of the sawtooth waveform is zero. Further, the value of the predetermined current value is set larger than 0 (A).

  The conduction control circuit 17G is a circuit that controls the conduction / non-conduction state of the transistors FET1 and FET2. The conduction control circuit 17G operates in accordance with the output signals of the backflow detection circuit 16G and the voltage comparator COMP. The oscillator OSC is a circuit that determines the repetition period of the operation cycle of the DC-DC converter 1G. The oscillation period of the oscillation circuit is changed according to the output of the averaging circuit 12G or the voltage comparator COMP. Here, when the oscillation cycle is changed in accordance with the output of the voltage comparator COMP, control for switching between the two oscillation cycles is performed. On the other hand, when changing the oscillation cycle according to the output of the averaging circuit 12G, it is possible to control the cycle to be continuously changed according to the output of the averaging circuit 12G.

  When the comparison result that the average current AI of the coil current IL1 is larger than the predetermined current value is obtained by the voltage comparator COMP, the load needs to supply power from the DC-DC converter 1G. At this time, the conduction state of the transistor FET2 is maintained by the conduction control circuit 17G until the accumulation of power in the choke coil L1 is started regardless of the detection result of the backflow detection circuit 16G. The oscillator OSC outputs an oscillation signal having a predetermined period. As a result, PWM fixed control that operates at a fixed frequency determined by the oscillator OSC is performed.

  In the PWM fixed control, even when the bottom value of the coil current IL1 is negative and the current direction is reversed and current flows from the load toward the choke coil L1, the power on the load side moves to the choke coil L1. Will be accumulated. That is, even if a switching operation is performed in a predetermined cycle by the PWM method and excessive power is released every cycle toward the load side, the excess power can be returned to the choke coil L1 every cycle. An increase in Vout can be suppressed. Therefore, even when the power supplied to the load is very low and the bottom value of the coil current IL1 is negative, the power can be supplied by PWM control. In this case, the predetermined current value that determines switching between PWM control and PFM control is an average value of the coil current IL1 having a negative bottom value.

  Thereafter, when the load is stopped and the load current is not consumed, the voltage comparator COMP provides a comparison result that the average current AI of the coil current IL1 is smaller than a predetermined current value. Then, the conduction control circuit 17G performs control to turn off the transistor FET2 in accordance with the detection result of the backflow detection circuit 16G. Therefore, the generation of the reverse current of the choke coil L1 is prevented. The oscillation period of the oscillator OSC is increased according to the output signal of the averaging circuit 12G or the voltage comparator COMP. As a result, the PWM fixed control is switched to the PFM control that expands and contracts the repetition cycle of the operation cycle according to the load power amount.

  As described above, the PWM control can be switched to the PFM control even in the region where the bottom value of the sawtooth waveform of the coil current IL1 reaches a negative value. Thereby, even when the power supplied to the load is very low and the bottom value of the coil current IL1 becomes negative, the power can be supplied to the load by PWM control. That is, it is possible to supply very low power while keeping the operating frequency of the DC-DC converter constant. Therefore, it is possible to supply power to a load that requires low power supply and is easily affected by fluctuations in the operating frequency of the DC-DC converter.

  When power supply is not necessary, the PWM fixed control can be switched to the PFM control even in the region where the bottom value of the sawtooth waveform of the coil current IL1 reaches a negative value. As a result, the fixed loss can be reduced by the PFM control, so that the efficiency of the DC-DC converter can be increased.

  DESCRIPTION OF THE PREFERRED EMBODIMENTS Embodiments of a DC-DC converter control circuit and a control method thereof according to the present invention will be described below in detail with reference to the drawings based on FIGS. FIG. 2 shows a DC-DC converter 1 according to the present invention. An input terminal Tin is connected to the drain terminal of the transistor FET1, and the input voltage Vin is input thereto. The source terminal of the transistor FET1 is connected to one terminal of the choke coil L1 and the drain terminal of the transistor FET2. The source terminal of the transistor FET2 is connected to the ground potential. The gate terminals of the transistors FET1 and FET2 are connected to output terminals DH1 and DL1 of a control circuit 11, which will be described later.

  The other terminal of the choke coil L1 is connected to the output terminal Tout via the sense resistor RS1, and the input voltage Vin is stepped down and output as the output voltage Vout. An output capacitor C1 is connected between the output terminal Tout and the ground potential in order to store electric power supplied via the choke coil L1.

  Here, the back gate terminals of the transistors FET1 and FET2 are generally connected to the source terminals of the transistors FET1 and FET2. The back gate terminal is a well portion in the MOS transistor internal structure, and is a P-type conductive layer in the transistor. Therefore, in the transistors FET1 and FET2, a PN junction structure is generally formed from the source terminal to the drain terminal through the back gate terminal. This is a so-called body diode structure. The body diode has a rectifying action in the direction opposite to the normal current direction.

  The control circuit 11 performs control to maintain the output voltage Vout at a predetermined voltage value when stepping down the input voltage Vin and supplying power to the output terminal Tout by alternately controlling conduction of the transistors FET1 and FET2.

  The configuration of the control circuit 11 will be described. The input terminal CS1 and the input terminal FB1 to which both terminals of the sense resistor RS1 are connected are connected to the averaging circuit 12. The averaging circuit 12 includes a low pass filter LPF and a voltage amplifier AMP2. Input terminals CS1 and FB1 are connected to a non-inverting input terminal and an inverting input terminal of voltage amplifier AMP2 through low-pass filter LPF. The output voltage Vr output from the voltage amplifier AMP2 is input to the non-inverting input terminal of the voltage comparator COMP4, and the reference voltage e3 is input to the inverting input terminal. The output terminal of the voltage comparator COMP4 is connected to the OR gate circuit OR1 and the oscillator OSC.

  The input terminal CS1 and the input terminal FB1 are connected to the non-inverting input terminal and the inverting input terminal of the voltage amplifier AMP1. The input terminal FB1 is further connected to one terminal of the resistance element R1 connected to the ground potential via the resistance element R2, and the connection point of the resistance elements R1 and R2 is connected to the inverting input terminal of the error amplifier ERA1. ing. A reference voltage e1 is applied to the non-inverting input terminal of the error amplifier ERA1.

  The output terminal of the voltage amplifier AMP1 is connected to the non-inverting input terminal of the comparator COMP1. The output terminal of the comparator COMP1 is connected to the reset terminal R of the flip-flop circuit FF. The output terminal of the error amplifier ERA1 is connected to the inverting input terminal of the comparator COMP1 and the non-inverting input terminal of the voltage comparator COMP2. The reference voltage e2 is applied to the inverting input of the voltage comparator COMP2. The output terminal of the oscillator OSC is connected to the first input of the AND gate circuit AND2. The output terminal of the voltage comparator COMP2 is connected to the second input of the AND gate circuit AND2. The output terminal of the AND gate circuit AND2 is connected to the set terminal S of the flip-flop circuit FF.

  The set terminal S of the flip-flop circuit FF is triggered by a pulse signal PS of an oscillator OSC that oscillates at a predetermined cycle. The output terminal Q of the flip-flop circuit FF is connected to the output terminal DH1 of the control circuit 11, and the output terminal * Q is connected to the output terminal DL1 via the AND gate circuit AND1. Control signals VQ and * VQ are output from output terminals Q and * Q, respectively.

  The inverting input terminal of the voltage comparator COMP3 is connected to the input terminal LX1 of the control unit, and the non-inverting input terminal is grounded. The output terminal of the voltage comparator COMP4 is connected to the first input terminal of the OR gate circuit OR1, and the output terminal of the voltage comparator COMP3 is connected to the second input terminal. The output terminal of the OR gate circuit OR1 is connected to the first input terminal of the AND gate circuit AND1, and the output terminal * Q of the flip-flop circuit FF is connected to the second input terminal. The output terminal of the AND gate circuit AND1 is connected to the output terminal DL1 of the control circuit 11. The voltage comparator COMP3 and the AND gate circuit AND1 are a backflow prevention circuit that makes the transistor FET2 non-conductive in order to prevent the coil current IL1 of the choke coil L1 from backflowing. The OR gate circuit OR1 is a circuit for stopping the function of the backflow prevention circuit.

  FIG. 3 shows a circuit configuration of the oscillator OSC. The oscillator OSC includes a constant current circuit 20, transistors Q2 to Q4, a capacitor CT, a voltage comparator COMP11, and a reference voltage setting unit e13R. The constant current circuit 20 includes a transistor Q1, a current measurement resistor RT, a voltage amplifier AMP11, reference voltage setting units e11R and e12R, and a switch circuit SW11. The output voltage Vx output from the voltage comparator COMP4 is input to the switch circuit SW11. The switch circuit SW11 alternatively connects one of the reference voltage setting units e11R and e12R to the non-inverting input terminal of the voltage amplifier AMP11.

  The emitter terminal of the transistor Q1 is connected to the input terminal of the current measuring resistor RT and also to the inverting input terminal of the voltage amplifier AMP11. The collector terminal of the transistor Q1 is connected to the transistor Q2 that is a load. The base terminal of the transistor Q1 is connected to the output terminal of the voltage amplifier AMP11. The switch circuit SW11 is connected to the non-inverting input terminal of the voltage amplifier AMP11. The output terminal of the current measuring resistor RT is grounded.

  Transistors Q2 and Q3 form a current mirror circuit. Capacitor CT is connected in series with transistor Q3. Transistor Q4 is connected in parallel with capacitor CT. The output terminal of the capacitor CT and the drain terminal of the transistor Q4 are connected to the non-inverting input terminal of the voltage comparator COMP11. The reference voltage setting unit e13R is connected to the inverting input terminal of the voltage comparator COMP11. The output terminal of the voltage comparator COMP11 is connected to the gate terminal of the transistor Q4 and to the output terminal PT of the oscillator OSC. A pulse signal PS is output from the voltage comparator COMP11.

  The operation of the DC-DC converter 1 will be described. The DC-DC converter 1 according to the present invention is a PWM-PFM control type DC-DC converter in which a control method can be switched between PWM fixed control and PFM control. The PWM fixed control is control that fixes the repetition cycle of the operation cycle to a predetermined cycle regardless of the load power amount. Moreover, PFM control is control which expands / contracts the repetition period of an operation cycle according to load electric energy.

  The low pass filter LPF of the averaging circuit 12 is a filter circuit that extracts an average current value of the coil current IL1 flowing through the sense resistor RS1. Here, the average current value of the coil current IL1 represents the output current of the DC-DC converter 1. An output voltage Vr corresponding to the average current value is output from the voltage amplifier AMP2. The voltage comparator COMP4 determines whether the output voltage Vr is larger or smaller than the reference voltage e3. Here, the reference voltage e3 is a voltage value that determines a predetermined current value. The predetermined current value is a value representing the minimum value of the load current, and is appropriately determined according to the power consumption of the load connected to the DC-DC converter. The value of the predetermined current value can be a value equal to or less than the average value of the coil current IL1 in which the bottom value of the sawtooth waveform is zero.

  When the output voltage Vr of the voltage amplifier AMP2 is larger than the reference voltage e3, the voltage comparator COMP4 determines that the output current of the DC-DC converter 1 is larger than the minimum value of the load current and the load is in a heavy load state. Done. The voltage comparator COMP4 outputs a high-level output voltage Vx. On the other hand, when the output voltage Vr is smaller than the reference voltage e3, it is determined that the output current of the DC-DC converter 1 is smaller than the minimum value of the load current and the load is in the no-load state. The voltage comparator COMP4 outputs a low level output voltage Vx.

  The selection between the PWM control and the PFM control is performed based on the comparison result of the voltage comparator COMP4. First, the operation of the DC-DC converter 1 when PWM fixed control is performed under heavy load will be described. When the load is in a heavy load state, a high level output voltage Vx is output from the voltage comparator COMP4. When the high-level output voltage Vx is input to the OR gate circuit OR1, the output voltage Vm of the OR gate circuit OR1 is fixed to the high level. That is, the output voltage Vd of the voltage comparator COMP3 is masked.

  The AND gate circuit AND1 passes the control signal * VQ when the high level output voltage Vm is input from the OR gate circuit OR1, and the control signal * VQ when the low level output voltage Vm is input. This is a circuit for masking. Now, since the high-level output voltage Vm is input to the first input of the AND gate circuit AND1, the control signal * VQ is allowed to pass. As a result, it is understood that when the output voltage Vx is at a high level, the backflow prevention operation of the coil current IL1 by the voltage comparator COMP3 is blocked.

  FIG. 4 shows the waveform of the coil current IL1 when the PWM fixed control is performed. FIG. 4 is an operation waveform in a state where the bottom value Ib of the coil current IL1 reaches a negative value. In the period Ton, the control signal VQ is at a high level, and the transistor FET1 is turned on. Therefore, the coil current IL1 increases at a predetermined time rate. On the other hand, in the period Toff, the control signal * VQ is at the high level, and the transistor FET2 is turned on. Therefore, the coil current IL1 decreases at a predetermined time rate. As a result, the coil current IL1 flows in a sawtooth waveform. An average value of the coil current IL1 flowing in a sawtooth waveform is defined as an average current AI. The average current AI represents the load current flowing through the load.

  Periods (2) and (3) are periods in which the control signal * VQ is at a high level and the transistor FET2 is in a conductive state. (2) In the region, the power accumulated in the choke coil L1 is released to the output terminal Tout side, and the emission is completed when the coil current IL1 reaches zero. In the subsequent region (3), since the conduction state of the transistor FET2 is maintained, the coil current IL1 flows from the output terminal Tout side toward the choke coil L1, and excess power is returned to the choke coil L1.

  (3) After the period ends, the control signal * VQ transitions to a low level, and the control signal VQ transitions to a high level, so that the transistor FET1 enters a conductive state. In the region (4) which is the initial stage of the period, the electric power stored in the choke coil L1 is regenerated to the input terminal Tin side. Here, since the transistor FET1 conducts and regeneration is performed, the power consumption of the coil current IL1 in the regeneration path can be further reduced. When the regeneration is completed, the period (1) is reached, and the coil current IL1 is inverted and the coil current IL1 starts to flow from the input terminal Tin side toward the choke coil L1. Electric power begins to be accumulated in the choke coil L1, and the next cycle is started.

  As described above, the switching control of the transistors FET2 and FET1 during the periods (3) and (4) in FIG. 4 constitutes a step-up DC-DC converter that supplies power from the output terminal Tout to the input terminal IN. It becomes. Therefore, when the DC-DC converter is operating under PWM fixed control, the transistor FET2 can be used to boost the output voltage Vout and return it to the input side. That is, a part of the electric power discharged to the output terminal Tout side is returned to the choke coil L1. As a result, an increase in the output voltage Vout due to excess power accumulated in the output capacitor C1 can be reduced, and the output voltage Vout can be prevented from falling into an uncontrollable state.

  Next, the operation when the load changes from the operating state to the stopped state and is switched from the PWM fixed control to the PFM control will be described. As the load becomes lighter, the average current AI decreases. As the average current AI decreases, the output voltage Vr of the voltage amplifier AMP2 decreases. When the average current AI becomes lower than the predetermined current value PI (FIG. 4), the output voltage Vr becomes lower than the reference voltage e3. Then, the output voltage Vx of the voltage comparator COMP4 transitions from a high level to a low level, and is determined to be in a low load state. Then, switching from PWM fixed control to PFM control is performed according to the low-level output voltage Vx. At this time, as shown in FIG. 4, even when the bottom value of the sawtooth wave of the coil current IL1 is a negative value, it is possible to switch from PWM control to PFM control.

  When the low-level output voltage Vx is input to the OR gate circuit OR1, the output voltage Vm of the OR gate circuit OR1 is not fixed to the high level. That is, the output voltage Vd of the voltage comparator COMP3 is not masked. Then, the voltage comparator COMP3 outputs the low level output voltage Vd when the backflow is detected, thereby fixing the output of the AND gate circuit AND1 to the low level and masking the * control signal VQ. The voltage comparator COMP3 outputs the high-level output voltage Vd when it does not detect backflow, thereby allowing the * control signal VQ to pass. As a result, the voltage comparator COMP3 performs the backflow prevention operation of the coil current IL1. Therefore, the PWM fixed control ends.

  The PFM control will be described. The PFM control is a control that expands and contracts the repetition cycle of the operation cycle according to the load electric energy. In the DC-DC converter 1, the repetition period of the operation cycle is changed by both a method of first changing the oscillation frequency f of the oscillator OSC and a method of skipping switching of the transistor FET1.

  A method of performing PFM control by variably controlling the oscillation frequency f of the oscillator OSC will be described with reference to FIG. When the input output voltage Vx transitions from a high level to a low level, the switch circuit SW11 switches the reference voltage connected to the non-inverting input of the voltage amplifier AMP11 from the reference voltage e11 to e12. Here, the reference voltage e12 is a value lower than e11. The transistor Q1 operates as a constant current circuit so that the voltage generated by the reference voltage of the voltage amplifier AMP11 and the current Irt flowing through the current measurement resistor RT is always constant. Here, since the reference voltage decreases from the reference voltage e11 to e12, the output voltage of the voltage amplifier AMP11 decreases and the base current of the transistor Q1 decreases. When the base current of the transistor Q1 decreases, the current Irt flowing through the current measurement resistor RT also decreases. Therefore, the current Irt decreases to a value corresponding to the reference voltage e12. A current Irt flows through the load transistor Q2. Since transistor Q2 and transistor Q3 form a current mirror circuit, current Irt also flows through transistor Q3.

  Since the capacitor CT is connected in series with the transistor Q3, the capacitor CT is charged with the current Irt flowing through the transistor Q3, and the voltage rises with time. When the voltage of the capacitor CT rises to the voltage e13, the voltage comparator COMP11 outputs a high level pulse signal PS. Then, the transistor Q4 is turned on and the charge of the capacitor CT is discharged, so that the voltage of the capacitor CT becomes 0 (V). When the charge of the capacitor CT becomes 0 (V), the pulse signal PS becomes low level and the transistor Q4 is turned off, so that the capacitor CT is charged again with the current Irt flowing through the transistor Q3. Thus, the oscillator OSC oscillates at a constant oscillation frequency f determined from the capacitance of the capacitor CT and the amount of current Irt.

  Now, as the reference voltage is lowered from the reference voltage e11 to e12, the current Irt is lowered. Therefore, the oscillation frequency f is lowered by changing the reference voltage e11 to e12. As a result, the operating frequency of the DC-DC converter 1 is lowered and the switching frequency is lowered, so that the fixed loss is reduced and the efficiency is increased.

  As described above, it is possible to shift from the PWM fixed control to the PFM control in response to detecting the low load state and setting the output voltage Vx to the low level. Then, the PWM fixed control is stopped in response to the detection of the low load state, and after that, it is detected that the output voltage Vout has risen above the predetermined voltage value, and then the PWM control is shifted to the PFM control. Switching operation with a fast response speed is possible.

  Next, a method for performing PFM control by skipping switching of the transistor FET1 will be described. When the DC-DC converter is forcibly switched at a fixed frequency without allowing reverse flow, when the load is extremely light, the output voltage of the DC-DC converter rises and the DC-DC converter becomes uncontrollable. There is a thing. Therefore, the rise of the output voltage is prevented by PFM control that intermittently inhibits the transistor FET1 from being turned on.

  As described above, since the back-flow prevention operation of the coil current IL1 is performed by the voltage comparator COMP3 in response to the output voltage Vx being set to the low level, the PWM fixed control ends. Then, since the DC-DC converter is switched at a fixed frequency in a state where no backflow is allowed, the output voltage Vout increases.

  In FIG. 2, the error amplifier ERA1 of the control circuit 11 amplifies the difference between the reference voltage e1 and the voltage obtained by dividing the output voltage Vout of the DC-DC converter by the resistance element R1 and the resistance element R2, and outputs the amplified voltage. The voltage Vc is output. Therefore, when the output voltage Vout increases, the difference from the reference voltage e1 decreases, and the output voltage Vc decreases. When the output voltage Vc becomes lower than the reference voltage e2, the voltage comparator COMP2 outputs a low level output voltage Vd2.

  The AND gate circuit AND2 passes the pulse signal PS when the high level output voltage Vd2 is input from the voltage comparator COMP2, and masks the pulse signal PS when the low level output voltage Vd2 is input. It is. Therefore, when the low-level output voltage Vd2 is input, the AND gate circuit AND2 prevents the flip-flop circuit FF from being set by masking the pulse signal PS. As a result, the switching operation of the DC-DC converter is skipped. Thereby, when the load becomes light and the output voltage of the DC-DC converter rises above a specified value, the transistor FET1 is prohibited from being turned on, thereby preventing the output voltage from rising. At this time, since the transistor FET1 is switched at a frequency different from the oscillation frequency f of the oscillator OSC, the PFM control is performed.

  As described above, the PWM control can be switched to the PFM control even in the region where the bottom value of the sawtooth waveform of the coil current IL1 reaches a negative value. Thereby, even when the power supplied to the load is very low and the bottom value of the coil current IL1 becomes negative, the power can be supplied to the load by PWM control. That is, low power can be supplied while keeping the operating frequency of the DC-DC converter constant. Therefore, it is possible to supply power to a load that requires low power supply and is easily affected by fluctuations in the operating frequency of the DC-DC converter.

  When power supply is not necessary, the PWM fixed control can be switched to the PFM control even in the region where the bottom value of the sawtooth waveform of the coil current IL1 reaches a negative value. As a result, the fixed loss can be reduced by the PFM control, so that the efficiency of the DC-DC converter can be increased.

  The present invention is not limited to the above-described embodiment, and it goes without saying that various improvements and modifications can be made without departing from the spirit of the present invention. The oscillator OSC in FIG. 3 is a method of switching the oscillation frequency between two frequencies, but is not limited to this form. FIG. 5 shows a circuit of an oscillator OSC1 that continuously varies the oscillation frequency in accordance with the output current of the DC-DC converter. The voltage amplifier AMP11a provided in the oscillator OSC1 in FIG. 5 is a differential amplifier having two non-inverting input terminals. The reference voltage e11 is input to the first non-inverting input terminal, and the output voltage Vr output from the voltage amplifier AMP2 is input to the second non-inverting input terminal. The voltage amplifier AMP11a has a lower voltage of the reference voltage e11 and the output voltage Vr input to the two non-inverting inputs, and a voltage corresponding to the current Irt flowing through the transistor Q1 input to the inverting input. Amplify the difference.

  In a high load state, the output current of the DC-DC converter is large, and the output voltage Vr of the voltage amplifier AMP2 is higher than the reference voltage e11. Therefore, the voltage amplifier AMP11a amplifies the difference between the reference voltage e11 and the voltage generated by the current Irt. Therefore, the oscillation frequency f of the oscillator OSC1 is a constant value determined by the reference voltage e11.

  When shifting to the low load state, the output current of the DC-DC converter decreases, and the output voltage Vr of the voltage amplifier AMP2 also decreases. When the output voltage Vr becomes lower than the reference voltage e11, the voltage amplifier AMP11a amplifies the difference between the output voltage Vr and the voltage generated by the current Irt. The output voltage Vr decreases as the output current of the DC-DC converter decreases, and the current Irt decreases as the output voltage Vr decreases. When the current Irt decreases, the charging time of the capacitor CT becomes longer, so that the oscillation frequency f of the oscillator OSC1 decreases.

  Thus, the oscillator OSC1 operates at a predetermined oscillation frequency when the output current of the DC-DC converter is larger than the output current determined by the reference voltage e11. When the output current of the DC-DC converter is smaller than the output current determined by the reference voltage e11, the oscillation frequency is lowered according to the shortage. Thereby, PFM control according to the output current value of a DC-DC converter is attained. The reference voltage e11 is set so that the output current determined by the reference voltage e11 is equal to or less than the output current determined by the reference voltage e3 of the voltage comparator COMP4 (output current when switching from PWM control to PFM control). It is desirable to set

  The averaging circuit 12 includes the low-pass filter LPF, but is not limited to this form. FIG. 6 shows a circuit diagram of an averaging circuit 12a which is another configuration example. The averaging circuit 12a includes voltage amplifiers AMP1a and AMP2a, a minimum current detection circuit 30, a minimum / maximum current detection circuit 31, a selection circuit 32, and an arithmetic circuit 33. The input terminal CS1 is connected to the non-inverting input terminal of the voltage amplifier AMP1a, and the input terminal FB1 is connected to the inverting input terminal. The input terminal CS1 is connected to the inverting input terminal of the voltage amplifier AMP2a, and the input terminal FB1 is connected to the non-inverting input terminal. The output terminal of the voltage amplifier AMP1a is connected to the input terminal of the minimum / maximum current detection circuit 31. The peak value Imax output from the minimum / maximum current detection circuit 31 is input to the arithmetic circuit 33. The bottom value Imin1 output from the minimum / maximum current detection circuit 31 is input to the selection circuit 32 and the arithmetic circuit 33. The output terminal of the voltage amplifier AMP2a is connected to the input terminal of the minimum current detection circuit 30. The bottom value Imin2 output from the minimum current detection circuit 30 is input to the selection circuit 32. The bottom value Imin output from the selection circuit 32 is input to the arithmetic circuit 33. An output voltage Vr is output from the arithmetic circuit 33.

  The operation of the averaging circuit 12a will be described. The minimum / maximum current detection circuit 31 is a sample-and-hold circuit that acquires the peak value of the coil current IL1 and outputs it as the peak value Imax, and acquires the positive bottom value of the coil current IL1 and outputs it as the bottom value Imin1. . The bottom value Imin1 is greater than 0 when the bottom value of the coil current IL1 is positive, and is 0 when the bottom value of the coil current IL1 is negative. The minimum current detection circuit 30 is a sample and hold circuit that acquires the absolute value of the negative bottom value of the coil current IL1 and outputs the absolute value as the bottom value Imin2. The bottom value Imin2 is 0 when the bottom value of the coil current IL1 is positive, and is greater than 0 when the bottom value of the coil current IL1 is negative.

  The selection circuit 32 is a circuit that selects one of the bottom values Imin1 and Imin2 according to the sign of the bottom value of the coil current IL1. When the input bottom value Imin1 is greater than 0, the selection circuit 32 determines that the bottom value of the coil current IL1 is positive and selects the bottom value Imin1. On the other hand, when the input bottom value Imin1 is 0, it is determined that the bottom value of the coil current IL1 is negative, and the bottom value Imin2 is selected. Then, the selection circuit 32 outputs the selected bottom value Imin1 or Imin2 to the arithmetic circuit 33 as the bottom value Imin.

  The arithmetic circuit 33 is a circuit that selects an arithmetic method according to the sign of the bottom value of the coil current IL1. When the input bottom value Imin1 is greater than 0, the arithmetic circuit 33 determines that the bottom value of the coil current IL1 is positive, adds the bottom value Imin and the peak value Imax, and then divides by 2 I do. On the other hand, when the input bottom value Imin1 is 0, it is determined that the bottom value of the coil current IL1 is negative, and the calculation is performed by subtracting the bottom value Imin from the peak value Imax and dividing by 2. This is because the bottom value Imin2 output from the minimum current detection circuit 30 is a negative absolute value. Thus, the average current value of the coil current IL1 can be calculated by the averaging circuit 12a.

  Note that the arithmetic circuit 33 may be configured by either digital or analog circuits. In the case where the arithmetic circuit 33 is configured by an analog circuit, an operation by an operational amplifier is provided, and the operation divided by 2 can be performed by setting the gain of the operational amplifier to half. Further, when the arithmetic circuit 33 is constituted by a digital circuit, it is possible to easily perform an operation of dividing by 2 by a 1-bit shift of the counter.

  Further, the control circuit 11 of this embodiment may be configured by a single or a plurality of semiconductor chips. The transistors FET1 and FET2 of this embodiment may be independent discrete power elements, or may be mounted as an LSI in the control circuit 11. Moreover, you may comprise the DC-DC converter 1 with a single or several semiconductor chip. Moreover, you may comprise the DC-DC converter 1 and the control circuit 11 as a module. Needless to say, the DC-DC converter 1 according to this embodiment can be applied to various power supply apparatuses.

  The choke coil L1 is an example of an inductive element, the transistor FET1 is an example of a first switching element, the transistor FET2 is an example of a second switching element, the voltage comparator COMP4 is an example of a comparison circuit, and the voltage comparator COMP3 is an example of a reverse current detection circuit. For example, the minimum / maximum current detection circuit 31 is an example of a peak value acquisition circuit, and the minimum current detection circuit 30 is an example of a bottom value acquisition circuit.

Principle of the present invention Circuit diagram of DC-DC converter 1 Circuit diagram of oscillator OSC Waveform diagram of coil current IL1 Circuit diagram of oscillator OSC1 Circuit diagram of averaging circuit 12a Circuit diagram of DC-DC converter 100 according to the prior art

Explanation of symbols

DESCRIPTION OF SYMBOLS 1 DC-DC converter 11 Control circuit 12 Average circuit AMP1, AMP2 Voltage amplifier AND1, AND2 AND gate circuit COMP1 thru | or COMP4 Voltage comparator FET1, FET2 Transistor IL1 Coil current LPF Low pass filter OR1 OR gate circuit OSC Oscillator

Claims (8)

A DC-DC converter comprising: a first switching element that conducts when storing electric power in an inductive element; and a second switching element that conducts switching under control according to a discharge period of power accumulated in the inductive element to a load. The control between the PWM fixed control in which the repetition cycle of the operation cycle is a predetermined cycle regardless of the load power amount, and the PFM control in which the repetition cycle of the operation cycle is expanded or contracted according to the load power amount in a light load, A control circuit for a DC-DC converter that can be selected,
A current measuring circuit for measuring an inductive element current flowing through the inductive element;
A comparison circuit that compares the inductive element current with a predetermined current value;
A backflow detection circuit that monitors a backflow current due to conduction of the second switching element and detects reversal of the current direction;
When the comparison result of the comparison circuit is input and the inductive element current is smaller than the predetermined current value, the second switching element is made non-conductive according to the detection result of the backflow detection circuit, and the inductive element current is A conduction control circuit that maintains the conduction state of the second switching element until the accumulation of power in the inductive element is started regardless of the detection result when more than a predetermined current value;
An oscillator for determining a repetition period of an operation cycle, comprising: an oscillation circuit that changes an oscillation period in accordance with a measurement result by the current measurement circuit or a comparison result by the comparison circuit; Control circuit.   2. The control circuit for a DC-DC converter according to claim 1, wherein the value of the predetermined current value is equal to or less than an average value of the inductive element currents having a bottom value of zero and is larger than zero. .   The oscillation circuit sets the oscillation period to a predetermined period when the inductive element current is greater than the predetermined current value based on a comparison result by the comparison circuit, and the inductive element current is less than the predetermined current value. 2. The control circuit for a DC-DC converter according to claim 1, wherein the oscillation period is longer than the predetermined period.   The oscillation circuit sets the oscillation period to a predetermined period when the inductive element current is greater than the predetermined current value based on a measurement result by the current measurement circuit, and the inductive element current is greater than the predetermined current value. 2. The control circuit for a DC-DC converter according to claim 1, wherein when the frequency is small, the oscillation period is increased in accordance with a decrease amount of the inductive element current.   The DC-DC converter control circuit according to claim 1, wherein the current measurement circuit includes a low-pass filter. The current measurement circuit includes:
A peak value acquisition circuit for acquiring a peak value of the inductive element current;
A bottom value acquisition circuit for acquiring a bottom value of the inductive element current;
The control circuit for a DC-DC converter according to claim 1, further comprising: an arithmetic circuit that calculates an average value of the peak value and the bottom value. A DC-DC converter comprising: a first switching element that conducts when storing power in the inductive element; and a second switching element that conducts switching under control according to a discharge period of the power accumulated in the inductive element to a load. The control between the PWM fixed control in which the repetition cycle of the operation cycle is a predetermined cycle regardless of the load power amount, and the PFM control in which the repetition cycle of the operation cycle is expanded or contracted according to the load power amount in a light load, A selectable DC-DC converter,
A current measuring circuit for measuring an inductive element current flowing through the inductive element;
A comparison circuit that compares the inductive element current with a predetermined current value;
A backflow detection circuit that monitors a backflow current due to conduction of the second switching element and detects reversal of the current direction;
When the comparison result of the comparison circuit is input and the inductive element current is less than the predetermined current value, the second switching element is made non-conductive according to the detection result of the backflow detection circuit, and the inductive element current is A conduction control circuit that maintains the conduction state of the second switching element until the accumulation of power in the inductive element is started regardless of the detection result when more than a predetermined current value;
An oscillator for determining a repetition period of an operation cycle, comprising: an oscillation circuit that changes an oscillation period in accordance with a measurement result by the current measurement circuit or a comparison result by the comparison circuit; . A DC-DC converter comprising: a first distribution path that conducts when storing power in the inductive element; and a second distribution path that conducts switching under control according to a discharge period of the power accumulated in the inductive element to a load. The control between the PWM fixed control in which the repetition cycle of the operation cycle is a predetermined cycle regardless of the load power amount, and the PFM control in which the repetition cycle of the operation cycle is expanded or contracted according to the load power amount in a light load, A control method for a DC-DC converter that is selectable,
Measuring an inductive element current flowing through the inductive element;
Comparing the inductive element current to a predetermined current value determined in advance;
Monitoring a backflow current due to conduction of the second flow path and detecting reversal of the current direction;
When the comparison result of the comparing step is input and the inductive element current is less than the predetermined current value, the second flow path is made non-conductive according to the detection result of the backflow detection circuit, and the inductive element current is Maintaining the conduction state of the second flow path until the accumulation of power to the inductive element is started regardless of the detection result when the current value is greater than the predetermined current value;
An oscillator that determines a repetition period of an operation cycle, comprising: a step of changing an oscillation period according to a measurement result of the measuring step or a comparison result of the comparing step Control method.

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