JP2006230079A - Inverter control system - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide an inverter control system capable of smoothly changing a motor output between a non-deformed mode and a one-pulse mode by suppressing pulse disappearance from a PWM signal near a zero cross in an overmodulated mode. <P>SOLUTION: When a modulation rate Am of pulse-modulated three-phase fundamental voltage commands Vu, Vv, Vw exceeds 1, the overmodulated mode is executed. In the overmodulated mode, three-phase voltage commands U, V, W are converted into staircase voltages with a high-level value (duty ratio 100%), a low-level value (duty ratio 0%) and a middle-level value (duty ratio 50%). Thus, noise and vibration can be suppressed along with improvement in a voltage utilization factor. <P>COPYRIGHT: (C)2006,JPO&NCIPI

Description

  The present invention relates to an inverter control system that converts DC power into AC power and supplies power to an AC motor.

  In conventional PWM (pulse width modulation) control of an inverter that drives and controls an AC motor, the following Patent Document 1 discloses an asynchronous overmodulation mode, synchronization in order to increase the output voltage of the inverter beyond the sine wave mode. It is proposed to switch to the one-pulse mode of the equation. The synchronous type here means that the phase of the waveform of the voltage command to be pulse width modulated and the waveform of the carrier voltage are matched. Hereinafter, this inverter control method is referred to as a three-mode switching type inverter control method.

  In this three-mode switching inverter control method, the overmodulation mode is performed when the modulation factor Am becomes 1 or more. In this overmodulation mode, a high level period is set in which the voltage command is fixed to a high level value (duty ratio 100% in pulse width modulation) near the maximum value of the voltage command, and the voltage command is set low near the minimum value of the voltage command. A low level period that is fixed to a level value (duty ratio 0% in pulse width modulation) is set, and the voltage command is monotonously changed between the high level period and the low level period to change the high level value and the low level period. The transition period is set to be an inclined waveform that is linked to the level value. However, in this overmodulation mode, the inverter output voltage is changed in accordance with the voltage command by increasing the high level period and the low level period as the voltage command increases.

Note that the sine wave mode is a mode in which pulse width modulation is performed without giving the above-described deformation to the voltage command of the input sine wave waveform when the maximum amplitude value of the voltage command is less than 1. In the one-pulse mode, the modulation rate Am reaches approximately 1.27, and the input voltage command is converted into a pulse voltage that transitions between the maximum value and the minimum value of the carrier voltage in the pulse width modulation.
JP 2003-125597 A

  However, in the inverter control method of Patent Document 1 described above, the waveform of the voltage command during the transition period of the overmodulation mode becomes steep, so that the pulse width modulation is performed in the vicinity of the zero cross where the polarity of the voltage command changes particularly in the high rotation range. Originally generated pulse component disappears, causing discontinuity of the motor voltage, or inconvenience that DC current flows to the motor on average, or between the above sine wave mode and 1 pulse mode As a result, the motor output could not be changed continuously.

  The present invention has been made in view of the above problems, and suppresses the disappearance of a pulse from a PWM signal in the vicinity of the zero cross in the overmodulation mode, and smoothly changes the motor output between the sine wave mode and the one-pulse mode. It is an object of the present invention to provide an inverter control method that can be performed.

The inverter control system of the present invention that solves the above-described problems is characterized in that the amplitude of the voltage command of each phase to be output by pulse width modulation to the gate driver that controls the switching element of the inverter for AC motor drive control is X, and the pulse width Modulation rate calculation means for calculating X / Y as a modulation rate Am when the carrier voltage amplitude used for modulation is Y, and a sine wave mode when the modulation rate Am is less than 1, a predetermined modulation rate greater than 1 Control mode selection means for selecting a predetermined overmodulation mode at a modulation factor Am between the sine wave mode and the one pulse mode, and the input voltage command as the sine wave. Output as it is in the mode, convert the input voltage command into a binary pulse voltage that transitions between the maximum value and the minimum value of the carrier voltage in the one-pulse mode, and outputs it. The input phase voltage command has a high level value (duty ratio 100% in pulse width modulation) near the maximum value of the voltage command in the overmodulation mode, and a low level value near the minimum value of the phase voltage command. Voltage command generating means for converting and outputting an overmodulated voltage having a duty ratio (0% in pulse width modulation), and generating a PWM signal for pulse width modulating the output voltage of the voltage command generating means and outputting it to the gate driver The voltage command generating means includes a high level period during which the high level value is output in the overmodulation mode and a low level period during which the low level value is output along with an increase in amplitude of the phase voltage command. An extended inverter control system,
The voltage command generating means is a substantially constant value separated from the high level value and the low level value by a predetermined value in a transition period that is a period between the high level period and the low level period of the overmodulation mode. It has a middle level period in which a middle level value is output to the PWM signal generating means.

  That is, in the inverter control system of the present invention, during the transition period of the overmodulation mode, a middle level value that is a substantially constant value separated from the high level value and the low level value by a predetermined value is output to the PWM signal generating means as a voltage command value. To do.

  In this way, when performing the overmodulation mode, the carrier voltage of the pulse width modulation can be crossed at a middle level value which is a substantially constant level during the transition period. It is possible to prevent the pulse component from disappearing due to a phenomenon (see FIG. 12) that occurs when the slope of the transition period intersects with a steep voltage command. Thereby, it is possible to prevent the positive current component and the negative current component of the inverter from becoming unbalanced as a result of the disappearance of the pulse component. Preferably, the middle level period of a predetermined phase is formed temporally simultaneously with the high level period and the low level period of another phase.

  Note that the “middle level value” referred to above is 13 to 87%, more preferably 25 to 75%, more preferably 40 to 60%, and more preferably in terms of the duty ratio of pulse width modulation. It shall mean a value of 48-52%.

  In a preferred aspect, the middle level value is set to a median value corresponding to a duty ratio of 50% in the pulse width modulation. For example, when this middle level value in the middle level period of the transition period is set to a median value (0 value) corresponding to a PWM duty ratio of 50%, the phase voltage command of one phase is the period of electrical angles 0 to θ. 0 (Duty 50%), electrical angle θ to π−θ is 1 (Duty 100%), electrical angle θ−π to π + θ is 0 (Duty 50%), electrical angle π + θ to 2π−θ is −1 (Duty 0%), the period of electrical angle 2π−θ˜2π is 0 (Duty 50%). Thereby, the voltage utilization factor can be continuously shifted from the sine wave mode to the one-pulse mode by changing the electrical angle (hereinafter also referred to as phase angle) θ.

  In other words, the maximum value of the voltage utilization factor when the voltage command of the staircase voltage waveform that always outputs 0 (Duty 50%) is pulse width modulated during the transition period is 78%, and the voltage command of the sine wave is pulse width modulated. In this case, the maximum value of the voltage utilization factor is about 61%, so that the inverter output can be continuously changed in the overmodulation mode in which the voltage utilization factor is 61 to 78%. In addition, when the above-described intermediate steady value output period is shorter than the above-described transition period, the intermediate steady value output period is preferably arranged at the temporal center position of the transition period. In other words, pulse disappearance in the PWM signal can be prevented and a PWM signal with a duty ratio of 50% can be reliably output in this middle level value output period, so that it is applied to the motor in this middle level value output period. The DC component contained in the phase voltage to be applied can be reduced.

  In a preferred aspect, the middle level period occupies substantially all of the transition period. In this way, circuit processing for synthesizing the voltage command waveform in the transition period can be simplified.

  In a preferred aspect, the voltage command generation means outputs a three-phase voltage command of two-phase modulation to the PWM signal generation means in the sine wave mode. This makes it possible to reduce the voltage utilization range using the overmodulation mode by using a voltage utilization factor that is higher in two-phase modulation than in the case of using a three-phase voltage command having a sinusoidal waveform, A sudden voltage change in switching between the modulation mode and the overmodulation mode can also be prevented.

  In a preferred aspect, the voltage command generation means outputs a three-phase voltage command having a sinusoidal waveform on which a third harmonic voltage component is superimposed to the PWM signal generation means. In this case, overmodulation is achieved by using a higher voltage utilization factor of the three-phase voltage command of the sine wave waveform on which the third-order harmonic voltage component is superimposed than when using the three-phase voltage command of the sine wave waveform. The voltage utilization rate range using the mode can be reduced, and a sudden voltage change in switching between the sine wave mode (two-phase modulation mode) and the overmodulation mode can be prevented.

  In a preferred aspect, the voltage command generation means switches between the sine wave mode and the overmodulation mode, and the three-phase voltage command output by the voltage command generation means is set to a duty ratio of 0%, 50%, and 100%. This is performed when the corresponding high level value, low level value, and middle level value are reached. As a result, sudden changes in voltage can be reduced, so that increase in noise and vibration associated with sudden changes in voltage can be prevented.

  In a preferred aspect, the voltage command generation means and the PWM signal generation means are one phase at an electrical angle of 0 degrees of the motor in the overmodulation mode in a synchronization state where the synchronization number is 6n (n is a positive integer). Synchronous control is performed so that the median temporal value of the middle level period of the voltage command substantially coincides with the maximum or minimum point of the carrier voltage of the PWM signal generating means, and the synchronization number is 6n (n is a positive integer) If not, the synchronization control is stopped. As a result, pulse disappearance can be prevented even at a high motor speed.

  In a preferred aspect, the voltage command generating means switches the overmodulation mode to the one-pulse mode when the motor rotational speed is higher than a predetermined threshold rotational speed Mref, and the modulation factor is smaller than 1 and close to 1. Perform at a predetermined threshold. Thereby, even if the motor rotation speed becomes high and the middle level period is shortened, the overmodulation mode can be stably implemented by reliably detecting the zero cross point in the overmodulation mode. Note that this method is not limited to the overmodulation mode control of the present invention that uses the staircase voltage described above, but can also be implemented in conventional overmodulation mode control that uses a monotonic ramp voltage. That is, the following configuration of the invention can be employed.

When the amplitude of the voltage command of each phase to be output by pulse width modulation to the gate driver that controls the switching element of the inverter for AC motor drive control is X, and the amplitude of the carrier voltage used for the pulse width modulation is Y, Modulation rate calculation means for calculating X / Y as the modulation rate Am, the sine wave mode when the modulation rate Am is less than 1, and the 1 pulse mode when the predetermined modulation rate value greater than 1 is reached. And a control mode selection means for selecting a predetermined overmodulation mode at a modulation factor Am between the 1 pulse mode and the input voltage command as it is in the sine wave mode, and the input voltage command Is converted into a binary pulse voltage that transitions between the maximum value and the minimum value of the carrier voltage in the one-pulse mode, and the input phase voltage command is output in the overmodulation mode. Over-modulation having a high level value (duty ratio 100% in pulse width modulation) near the maximum value of the voltage command and a low level value (duty ratio 0% in pulse width modulation) near the minimum value of the phase voltage command A voltage command generating means for converting the voltage output to a voltage, and a PWM signal generating means for pulse width modulating the output voltage of the voltage command generating means and outputting the voltage to the gate driver. An inverter control system that extends a high level period in which the high level value is output in a modulation mode and a low level period in which the low level value is output as the amplitude of the phase voltage command increases,
The voltage command generation means sets the switching from the overmodulation mode to the one-pulse mode when the motor rotation speed is higher than a predetermined threshold rotation speed Mref, with a modulation rate voltage utilization rate smaller than 1 and close to 1. Inverter control system, characterized by being carried out at the threshold.

  Hereinafter, preferred embodiments of the three-phase AC motor control using the inverter control device of the present invention will be described. However, the present invention is not limited to the following embodiments, and the technical idea of the present invention may be constituted by other known techniques or a combination of other techniques.

  A motor control system using the inverter control device of Embodiment 1 will be described with reference to the drawings.

(Overall circuit)
The configuration of this motor control system is shown in FIG. 1 which is a block diagram. 1 is a DC power source, 2 is a driving device (also called a PWM inverter), 3 is a three-phase synchronous motor (also called a three-phase motor), 4 and 5 are two current sensors for detecting a phase current, and 6 is a motor rotation angle. It is a motor rotation position detection means to detect. The driving device 2 converts the DC power supplied from the DC power source 1 by PWM control of the switching elements into three-phase AC power and supplies the three-phase synchronous motor 3 to each other, and the switching elements of the inverter 7 are intermittently connected. And a control circuit 8 for controlling.

(Control circuit 8)
The configuration and basic operation of the control circuit 8 will be described with reference to FIG. FIG. 2 is a block circuit diagram showing a functional configuration of the control circuit 8.

  The control circuit 8 includes a motor rotation speed calculation means 81, a dq axis current generation means 82, a dq axis current command generation means 83, a dq axis voltage command generation means 84, a coordinate axis conversion means 85, a modulation factor calculation means 86, a control mode determination means. 87, voltage command generating means 88, PWM signal generating means 89, and switching gate driver 90.

  The motor rotation number calculation means 81 calculates the motor rotation number based on the rotation angle θ of the motor rotor input from the motor rotation position detection means 6 and outputs it to the dq axis current command generation means 83. The dq-axis current command generation means 83 is based on the motor rotation speed, the voltage Vb of the DC power source 1 input from the input voltage detection means 9, and the torque command trq * indicating the magnitude and direction of the torque input from the outside. A dq-axis current command that is a d-axis current id * and a q-axis current iq * as currents that should flow through the three-phase motor 3 is calculated. The dq-axis current generation means 82 converts the actual currents Iv and Iw input from the current sensors, that is, the motor current detection means 4 and 5, and calculates the d-axis current id 0 and the q-axis current iq 0 that are actual currents, respectively. And output to the dq-axis voltage command generation means 84.

  The dq-axis voltage command generation means 84 obtains and obtains the current deviation Δid (= d-axis current id * −d-axis current id0) and Δiq (= q-axis current iq * −q-axis current iq0) for each coordinate axis. The dq axis voltage commands Vd and Vq corresponding to the current deviations Δid and Δiq are calculated to converge the current deviations Δid and Δiq to 0, and are output to the coordinate axis conversion unit 85 and the modulation factor calculation unit 86. The coordinate axis conversion means 85 converts the input dq axis voltage commands Vd and Vq into three-phase basic voltage commands Vu, Vv and Vw, and inputs them to the voltage command generation means 88.

  The voltage command generation means 88 adjusts the three-phase basic voltage commands Vu, Vv, Vw by input signals to be described later, calculates the three-phase voltage commands U, V, W, and outputs them to the PWM signal generation means 89. The PWM signal generating means 89 generates three-phase PWM voltages Uout, Vout, Vout corresponding to the input three-phase voltage commands U, V, W, and these three-phase PWM voltages Uout, Vout, Vout are the switching gate driver 90. The signal is amplified by 6 and is made into six signal voltages UU, UL, VU, VL, WU, WL, and then individually applied to the gate electrodes of the switching elements of the corresponding three-phase inverter 7.

  The modulation factor calculation means 86 calculates the modulation factor Am based on the dq axis voltage commands Vd and Vq output from the dq axis voltage command generation means 84, and outputs them to the control mode determination means 87 and the voltage command generation means 88. . The control mode determination unit 87 currently selects from a plurality of predetermined modes stored in advance based on the motor rotation number input from the motor rotation number calculation unit 81 and the modulation factor Am input from the modulation factor calculation unit 86. The power mode is selected and commanded to the voltage command generating means 88. The voltage command generator 88 adjusts the three-phase basic voltage commands Vu, Vv, and Vw based on the rotation angle θ of the motor rotor, the modulation factor Am, and the selected mode, and outputs them to the PWM signal generator 89. The power three-phase voltage commands U, V, and W are determined.

  In the control circuit 8 described above, the configuration and operation of the circuit blocks other than the modulation factor calculation means 86, the control mode determination means 87, and the voltage command generation means 88 are the same as the three-phase voltage applied from the coordinate axis conversion means 85 to the PWM signal generation means 89. Since it is the same as an inverter control circuit for torque control type motor control that performs normal dq axis conversion that outputs three-phase basic voltage commands Vu, Vv, and Vw as commands U, V, and W, description thereof is omitted. Hereinafter, the modulation factor calculation means 86, the control mode determination means 87, and the voltage command generation means 88 are also referred to as mode control circuits.

(Description of mode control circuit)
Next, the operation of the mode control circuit that characterizes this embodiment will be described in detail.

(Description of mode)
First, the three modes employed in this embodiment will be described below with reference to FIG. However, in the following description, the three-phase basic voltage commands Vu, Vv, and Vw are described as being limited to sinusoidal voltages, but are not limited thereto as described later.

(Sine wave mode)
This sine wave mode corresponds to the sine wave mode referred to in the present invention. However, since the voltage command subjected to pulse width modulation is a sine wave, it is particularly referred to as a sine wave mode.

  (A), (b), and (c) of FIG. 3 are output from the U-phase three-phase basic voltage command Vu that is input to the voltage command generation means 88 and the voltage command generation means 88 between the electrical angles 2π. 6 is a waveform diagram showing a U-phase three-phase voltage command U, a carrier voltage formed in PWM signal generation means 89, and a U-phase three-phase PWM voltage Uout output from PWM signal generation means 89. FIG. However, in FIG. 3C, illustration of the carrier voltage and the U-phase three-phase PWM voltage Uout is omitted.

  The minimum value of the three-phase PWM voltages Uout, Vout, Vout output from the PWM signal generation means 89 (a negative maximum value when regarded as an AC voltage) is a positive maximum value with a duty ratio of 0% of the three-phase PWM voltage. A state in which the duty ratio of the three-phase PWM voltage is 100% (a positive maximum value when regarded as an AC voltage) is regarded as a modulation rate Am = 1. That is, at the modulation rate Am = 1, the maximum values of the three-phase basic voltage commands Vu, Vv, Vw (and the three-phase voltage commands U, V, W) coincide with the maximum value of the carrier voltage of the PWM signal generating means 89. .

  When the maximum value of the three-phase basic voltage commands Vu, Vv, Vw is smaller than the maximum value of the carrier voltage of the PWM signal generating means 89 (more strictly speaking, the absolute value of the amplitude of the three-phase basic voltage commands Vu, Vv, Vw) When the value is smaller than the absolute value of the amplitude of the carrier voltage of the PWM signal generating means 89), that is, when the modulation factor Am is less than 1, the voltage command generating means 88 is supplied with the three-phase basis inputted from the coordinate axis converting means 85. The voltage commands Vu, Vv, Vw are output as they are to the PWM signal generating means 89 as three-phase voltage commands U, V, W. Therefore, in this case, the PWM signal generating means 89 outputs three-phase PWM voltages Uout, Vout, Vout corresponding to the three-phase basic voltage commands Vu, Vv, Vw which are sine wave voltages. Hereinafter, this is referred to as a sine wave mode.

(Overmodulation mode)
FIG. 3A shows, for example, a U-phase voltage command U, a basic voltage command Vu, and a carrier voltage at a modulation rate Am = 1. When the modulation factor Am becomes 1 or more, the voltage command generation means 88 executes the overmodulation mode. In this overmodulation mode, the voltage command generator 88 corresponds to a period during which a high level value corresponding to a duty ratio of 100% in pulse width modulation is output (high level period) and a duty ratio of 0% in pulse width modulation. By setting the middle level period in which the middle level value is output in substantially all transition periods between the period in which the low level value is output (low level period), the high level value and the low level value are A voltage command of a ternary step waveform (staircase waveform) having a middle level value is generated during the period of time, and is output to the PWM signal generating means 89 as three-phase voltage commands U, V, W.

  The high-level period of the three-phase voltage commands U, V, W is a waveform that extends equally before and after the positive peak value (maximum value) of the three-phase basic voltage commands Vu, Vv, Vw as a temporal center point. In the low-level period of the three-phase voltage commands U, V, and W, the negative peak value (minimum value) of the three-phase basic voltage commands Vu, Vv, and Vw is the time center point, and the waveform extends equally before and after that. It has become.

  This is centered around the zero point of the amplitude of the three-phase basic voltage commands Vu, Vv, Vw (corresponding to 50% of the pulse width modulation), and the middle level value (pulse width modulation) with a phase angle θx before and after that and a total time width of 2θx Equivalent to 50% duty ratio). The phase angle θx means an electrical angle range that is half the middle level period in which the staircase voltage is 0, that is, the duty ratio of the PWM signal is 50%, that is, the middle level value.

  Increasing the phase angle θx decreases the effective voltage output from the inverter, and decreasing the phase angle θx increases the effective voltage output from the inverter. Therefore, in the overmodulation mode, by adjusting the phase angle θx according to the magnitude of the input three-phase basic voltage commands Vu, Vv, Vw, the sinusoidal waveform three-phase basic voltage commands Vu, Vv, Vw are set. Equivalent three-phase voltage commands U, V, W of staircase wave voltage are generated. The “three-phase voltage commands U, V, and W of the staircase voltage equivalent to the three-phase basic voltage commands Vu, Vv, and Vw having a sine wave waveform” are referred to as the three-phase basic voltage commands Vu, Vv, and Vw. This means that the torque generated by the motor is equal in the positive and negative half-wave periods of the three-phase voltage commands U, V, and W of the staircase voltage.

  In this embodiment, the phase angle θx is calculated using the following equation.

(4 / π) COSθx = Am
However, it is assumed that the amplitudes of the three-phase basic voltage commands Vu, Vv, and Vw are 1 when the duty ratio corresponds to 100% in the pulse width modulation. Therefore, by using the calculated phase angle θx, the phase angle θx has passed from the point (zero cross point) at which the three-phase basic voltage commands Vu, Vv, Vw become 0 level (duty ratio in pulse width modulation 50%). Middle level value and high level value (three-phase basic voltage commands Vu, Vv, Vw are positive half-wave periods) or low-level value (three-phase basic voltage) of the three-phase voltage commands U, V, W of the staircase voltage at the time The three-phase voltage commands U, V, and W of the staircase voltage can be created by shifting the commands Vu, Vv, and Vw to the negative half-wave period).

  Alternatively, the three-phase basic voltage commands Vu, Vv, and Vw may be input to a map stored in advance to obtain a corresponding phase angle range (also simply referred to as a phase angle) θx. The phase angle θx may be calculated by substituting the three-phase basic voltage commands Vu, Vv, and Vw into other arithmetic expressions stored in advance. Since the calculation itself of the phase angle θx based on the three-phase basic voltage commands Vu, Vv, and Vw is not the gist of the present invention and is well known, further explanation is omitted.

  Thus, in this overmodulation mode, the U-phase three-phase PWM voltage Uout is, as shown in FIG. 3A, a high-level period during which the high-level value of the three-phase voltage command U is output, and before and after The staircase voltage waveform has a middle level period in which the middle level value is output and a low level period in which the low level value of the three-phase voltage command U is output. The other phases are the same except that the phases are different by 2π / 3. The middle level period is assumed to be 2θx. When the modulation factor Am = 1, the phase angle θx is 0.21π. As the modulation factor Am increases, the middle level period 2θx is shortened, the high level period and the low level period are extended, and the average AC output current of the inverter increases. FIG. 3B shows the voltages of the respective parts when the modulation factor Am = 1.2.

(1 pulse mode)
When the modulation factor Am reaches 1.27, the middle level period becomes 0, and the three-phase voltage commands U, V, W are divided into a high level period in which a high level value is output and a low level period in which a low level value is output. Only have. FIG. 3C shows waveforms of the U-phase three-phase basic voltage command Vu and the three-phase voltage command U at this time. This state is referred to as a 1 pulse mode. In the 1-pulse mode, the three-phase PWM voltages Uout, Vout, and Vout have substantially the same waveform as the three-phase voltage commands U, V, and W.

  That is, in this embodiment, one of the three modes of the sine wave mode, the overmodulation mode, and the one-pulse mode is selected depending on the magnitude of the modulation factor Am, and in the overmodulation mode, the three-phase basic voltage commands Vu, Vv, Vw are selected. Is converted to a staircase voltage that is equivalent to a three-value step waveform on the inverter output basis, and in the 1-pulse mode, the three-phase basic voltage commands Vu, Vv, and Vw are converted to an equivalent binary pulse voltage waveform on the motor output basis. Output to the PWM signal generating means 89.

  The modulation factor calculation means 86 calculates the modulation factor Am. That is, the modulation factor calculating means 86 may calculate the modulation factor Am by obtaining a combined voltage command from the dq axis voltage commands Vd and Vq and dividing this by the voltage value at the modulation factor Am = 1 stored in advance. Also, other substantially equivalent calculation methods for the modulation factor Am may be employed.

  The control mode determination unit 87 selects the mode described above. If it demonstrates concretely, the modulation factor Am input will be compared with the mode switching threshold value stored beforehand, and mode switching will be performed.

  The voltage command generation unit 88 converts the three-phase basic voltage commands Vu, Vv, and Vw into the three-phase PWM voltages Uout, Vout, and Vout in the three modes described above according to the mode commanded from the control mode determination unit 87.

  An example in which the control circuit 8 is configured by software processing by a microcomputer will be described with reference to a flowchart shown in FIG. Step S100 is a subroutine that constitutes the motor speed calculation means 81, dq-axis current generation means 82, dq-axis current command generation means 83, dq-axis voltage command generation means 84, and coordinate axis conversion means 85, and rotates as described above. Based on the angle θ, the external torque command Trq *, and the motor currents Iv, Iw, three-phase basic voltage commands Vu, Vv, Vw are generated.

  Next, step S200 is performed. This step is the modulation rate calculation means 86, and calculates the modulation rate Am by the above calculation method. Next, step S300 is performed. This step is control mode determination means 87, which determines a control mode to be executed based on the value of the modulation factor Am. In this embodiment, the overmodulation mode has a synchronous control type overmodulation mode and an asynchronous control type overmodulation mode as will be described later, and the control mode determination means 87 determines the motor rotation speed from the motor rotation speed calculation means 81. When the overmodulation mode is selected based on the reading and the motor rotation speed, either the synchronous control overmodulation mode or the asynchronous overmodulation mode is selected. The synchronous overmodulation mode and the asynchronous overmodulation mode will be described later.

  Next, step S400 is performed. This step is a voltage command generating means 88, which sets the three-phase basic voltage commands Vu, Vv, Vw to predetermined values for each rotation angle θ according to the input selection mode, and sets the three-phase voltage commands U, V, W. The three-phase voltage commands U, V, W calculated in this way are compared with a carrier voltage of a predetermined frequency by the PWM signal generation means 89 (triangular wave voltage as in the normal case), and the three-phase PWM voltages Uout, Vout are compared. , Vout. The PWM signal generating means 89 can also be processed by software by a microcomputer, but it is easier to process by a simple hardware circuit.

(Switching between sine wave mode and overmodulation mode)
Next, switching timing from the sine wave mode using the sine wave voltage as the three-phase basic voltage commands Vu, Vv, and Vw to the overmodulation mode using the staircase voltage will be described with reference to FIG. This is essentially the same when switching from overmodulation mode to sinusoidal mode.

  In FIG. 5, t1 is a time point when the three-phase basic voltage command Vu zero-crosses from − to +, t2 is a time point when the three-phase basic voltage command Vw zero-crosses from + to −, t3 is a time point when the three-phase basic voltage command Vv is −− + T4 is the time when the three-phase basic voltage command Vu is zero-crossed from + to-, t5 is the time when the three-phase basic voltage command Vw is zero-crossed from-to +, t6 is the time when the three-phase basic voltage command Vv is + This is the time when zero crossing occurs. Each time point t1 to t6 is shifted by an electrical angle π / 3.

  Since the sine wave voltage waveform in the sine wave mode and the staircase voltage waveform in the overmodulation mode are greatly different, the three-phase voltage commands U, V, and W greatly change at the time of switching. The continuity of the PWM voltages Uout, Vout, and Vout is disturbed, causing problems such as disturbance of the motor current, vibration and noise associated therewith. Therefore, in this embodiment, in order to suppress this problem as much as possible, the values of the sine wave voltages of the three-phase voltage commands U, V, W in the sine wave mode and the three-phase voltage commands U, V, W in the overmodulation mode are used. At the zero-cross point of the three-phase basic voltage commands Vu, Vv, and Vw in the sine wave mode (at the time when the duty ratio is 50% in the pulse width modulation), which is the time point (phase) at which the value of the staircase wave voltage is well approximated The continuity is suitably maintained by switching.

  More specifically, the voltage command generation means 88 is instructed when the control mode determination means 87 is instructed to switch from the sine wave mode to the overmodulation mode or from the overmodulation mode to the sine wave mode. The zero crossing point of the phase basic voltage commands Vu, Vv, Vw is detected and the switching is performed. Thereby, a sudden voltage change can be suppressed. That is, when any one of the zero cross points t1 to t6 shown in FIG. 5 is detected or the zero cross point t1 is detected, the transition between the overmodulation mode and the sine wave mode is performed.

(Synchronous overmodulation mode and asynchronous overmodulation mode)
In this embodiment, two overmodulation modes, a synchronous overmodulation mode and an asynchronous overmodulation mode, are used as the overmodulation mode. Hereinafter, the synchronous overmodulation mode and the asynchronous overmodulation mode will be described with reference to FIG. However, the following explanation is based on three-phase AC.

  In a synchronous state where the frequency of the carrier voltage is an integral multiple of the frequency of the three-phase voltage commands U, V, and W, the carrier voltage frequency is 6n of the frequency of the three-phase voltage commands U, V, W (n is a positive integer) ) Times, n times the period of the n triangular wave pulses of the carrier voltage coincides with the electrical angle π / 3 of the motor. In the overmodulation mode, the middle level period of each phase is generated sequentially for each electrical angle π / 3.

  Therefore, when the 6n synchronization state is detected in the overmodulation mode, the three-phase voltage command U, the motor electrical angle 0 ° determined by the rising edge of the NM (North Maker) signal output from the motor rotation position detection means 6 By matching 0 degree of a predetermined phase (U phase in this embodiment) of V and W, the deviation of the three-phase voltage commands U, V, and W due to calculation errors can be removed, and synchronization is achieved. Can be taken.

  In this embodiment, n times the n triangular wave pulse period of the carrier voltage coincides with the electric angle π / 3 of the motor, and in the overmodulation mode, the middle level period of each phase is the phase every electric angle π / 3. Even if the so-called zero cross point at which the middle level values of the three-phase voltage commands U, V, W coincide with the zero level point of the carrier voltage matches the electrical angle 0 degree of the U-phase voltage command U. Good. In this way, in the pulse width modulation of the three-phase voltage commands U, V, and W, the zero-cross point of each phase is set at the temporal center position of the middle level period. Even if the modulation rate Am increases until the middle level period decreases, the zero-cross point between the three-phase voltage commands U, V, W and the carrier voltage can be accurately secured, and the three-phase voltage commands U, V, The positive and negative imbalance of W can be reduced.

  FIG. 6 shows the waveform of the U-phase voltage command U, the waveform of the carrier voltage, and the waveform of the NM signal in the synchronous overmodulation mode in the overmodulation mode when n = 1. θoffset is a phase delay of the U-phase voltage command U caused by a circuit processing delay or the like. In this embodiment, the electrical angle 0 of the U-phase voltage command is originally generated at a motor rotation angle of 0 degrees. However, when there is a predetermined angular deviation Δθ, this angle is known in advance. What is necessary is just to correct | amend deviation (DELTA) (theta).

  The control of the synchronous overmodulation mode described above is shown as a flowchart in FIG. As described above, the three-phase voltage commands U, V, and W in the overmodulation mode that is the staircase voltage are formed when the phase angle θx and the electrical angle of the three-phase voltage commands U, V, and W are known to 0 degrees. be able to. However, if the electrical angle 0 degrees of the three-phase voltage commands U, V, and W is deviated from the original position, noise and vibration due to this error increase. Therefore, it is first determined whether or not the 6n synchronization state described above is present, then it is determined whether or not the current state is the overmodulation mode, and then the position of the motor electrical angle 0 degree and the electrical angle θ of the voltage command for the U phase are determined. The angle difference θoffset from (0) is calculated, and the three-phase voltage commands U, V, W are phase-shifted by this angle difference θoffset. More specifically, the above-described three-phase voltage commands U, V, W The three-phase voltage commands U, V, and W in the overmodulation mode are created by shifting the phase angle θx serving as a reference for creating the above by the angle difference θoffset. Furthermore, the phase of the carrier voltage is adjusted so that the zero crossing from − to + of the carrier voltage occurs at an electrical angle of 0 degrees of the three-phase voltage commands U, V, and W. Thereby, noise and vibration can be suppressed and stable synchronization can be ensured.

  When the number of synchronizations is not 6n (n is a positive integer), the above-described three-phase voltage commands U, V, and W of the synchronous staircase voltage are not formed. This is because even if it is performed at a certain point in time, the phase shift between the three-phase voltage commands U, V, W and the carrier voltage increases with the passage of time.

  In the above description, the phase shift is corrected using the NM signal indicating the motor electrical angle of 0 degrees output from the motor rotational position detecting means 6, but various detections are made according to the output structure of the motor rotational position detecting means 6. Obviously, the angle can be used as well.

(Modification)
In the above-described embodiment, the frequency of the carrier voltage is constant. However, if the frequency of the carrier voltage is changed in accordance with the motor rotation speed, the carrier voltage decreases at low speeds, and noise and vibration are prevented from increasing. Because.

  However, the frequency of the carrier voltage may be changed so that the above 6n synchronization is always established with respect to the motor rotation speed only in the overmodulation mode in which vibration and noise are originally large. In this way, the 6n synchronous overmodulation mode described above can always be implemented in the overmodulation mode.

  Further, even in the sine wave mode, the carrier voltage is set so that the above-mentioned 6n synchronization is always established with respect to the motor rotation speed at a motor rotation speed of a predetermined rotation speed or higher where noise and vibration due to a decrease in the frequency of the carrier voltage are not a problem. The frequency may be changed.

  Control related to switching between the overmodulation mode and the one-pulse mode will be described with reference to the flowchart shown in FIG.

  In the above-described embodiment, it has been described that the overmodulation mode and the 1-pulse mode are switched when the modulation rate is 1 (phase angle θx = 0). It is obvious that the overmodulation mode and the one-pulse mode may be switched at Am.

  This embodiment is made based on this knowledge, and it is checked whether the motor rotation speed N has become a high rotation exceeding a predetermined threshold rotation speed Nref. Whether or not a small predetermined value (in this embodiment, the modulation factor is set to 0.95) is checked, and if it exceeds, the overmodulation mode is switched to the one-pulse mode at an early stage.

  In this case, since the rotational speed is high, it is necessary to set the zero cross point using a very short middle level period to create the three-phase voltage commands U, V, and W of the staircase voltage waveform. Therefore, the control stability can be improved. In this embodiment, the early switching is performed when switching from the overmodulation mode to the one-pulse mode. However, the switching (resulting in delay switching) is also performed when switching from the one-pulse mode to the overmodulation mode. You may go.

  The inverter control system of Example 3 will be described below. In this embodiment, instead of the three-phase voltage commands U, V, W of the three-phase modulation sine wave waveform used in the sine wave mode of the first embodiment, the three-phase voltage commands U, V, W of the two-phase modulation waveform are changed. The feature is the point adopted in the sine wave mode.

  The waveforms of the three-phase voltage commands U, V, W in the two-phase modulation sine wave mode are shown in FIG. However, the modulation factor Am is 1 in FIG.

  In this two-phase modulation, one phase closest to the high level or the low level among the three-phase basic voltage commands Vu, Vv, Vw is sequentially fixed to the high level value or the low level value, and the change of the interphase voltage caused thereby. This is a known inverter control system for level-shifting the other two-phase voltage commands to prevent this. In order to perform this two-phase modulation, the dq-axis voltage command generation means 84 shown in FIG. 2 may be changed to a known two-phase voltage command generation means 91. However, since the circuit configuration and operation of the inverter control of the two-phase modulation method are already well known, further explanation is omitted. The switching between the two-phase modulation type sine wave mode and the step modulation voltage overmodulation mode described above has the following advantages.

  First, at the modulation rate Am = 1, when switching between the two-phase modulation voltage shown in FIG. 9A and the staircase voltage shown in FIG. 9C, in both cases, the low level is set for each electrical angle π / 3. A combination of a value (duty ratio 0%), a high level value (duty ratio 100%) and a middle level value (50%) appears. Therefore, in switching between the sine wave mode (more precisely, the two-phase modulation mode) and the overmodulation mode, the two phases of the three-phase voltage commands U, V, and W are low level values (duty ratio 0%) immediately before and after the switching. ) And a high level value (duty ratio 100%), and the remaining one phase becomes a middle level value (50%), so that the duty ratios of all phases of the three-phase PWM voltages Uout, Vout, and Vout are not changed suddenly. Compared with the zero-crossing point of the three-phase voltage commands U, V, and W having a sinusoidal waveform performed in the first embodiment, the voltage continuity at the time of mode switching can be remarkably improved.

  More specifically, in the switching between the sine wave mode and the overmodulation mode described in the first embodiment, one phase of the three-phase voltage commands U, V, and W which are sine wave voltages has the same voltage, but the other two phases The voltage command is non-continuous because the voltage value is different. In contrast, when switching from the two-phase modulation sine wave mode to the over-modulation mode, the three-phase voltage commands U, V, and W have a duty ratio of 0%, 50%, and 100% when the modulation factor Am = 1. By switching to the staircase voltage in the overmodulation mode at the time, voltage continuity can be maintained in all three-phase staircase voltages. Of course, this is the same when switching from the overmodulation mode of the staircase voltage to the sine wave mode of the two-phase modulation.

  Also, according to the combination of the sine wave mode of the two-phase modulation and the over modulation mode, the voltage utilization factor of the sine wave mode of the two-phase modulation waveform is larger than the voltage utilization factor of the sine wave mode of the sine wave waveform. The voltage utilization factor at the switching point between the sine wave mode and the overmodulation mode can be set high. As a result, the voltage distortion is large compared to the sine wave mode of the sine wave waveform and the sine wave mode of the two-phase modulation. The voltage utilization ratio range in which the overmodulation mode in which the duty ratio of Uout, Vout, and Vout has a large sudden change can be reduced, and there is an advantage that noise and vibration can be reduced by that amount.

(Modification)
As a modification, a sine wave mode using a sinusoidal waveform voltage is used until the modulation rate Am becomes 1, and a sine wave mode of a two-phase modulation mode is used when the modulation rate Am increases further. If it increases, the overmodulation mode can be used, and if the modulation factor Am further increases, the 1-pulse mode can also be used.

  In the above embodiment, the three-phase voltage commands U, V, and W of the sine wave are used in the sine wave mode. Instead, as shown in FIG. Three-phase voltage commands U, V, and W having a waveform with superimposed components may be used.

  In this case, as can be seen from FIG. 9B, at the modulation rate Am = 1 at which switching occurs, the two-phase modulation voltage is high level value (duty ratio 100%) or low level value for each electrical angle π / 3 of the motor. (Duty ratio 0%) or a middle level value (50%), and by switching at this timing, the low level value (for each electrical angle π / 3, as in the third embodiment) (Duty ratio 0%), high level value (Duty ratio 100%) and middle level value (50%) are switched at the timing when the combination occurs, and as a result, the two-phase modulation voltage and staircase voltage of Example 3 As in the case of switching, the three-phase voltage commands U, V, and W do not change suddenly.

(Modification)
As a modification, a sine wave mode using a sinusoidal waveform voltage is used until the modulation rate Am becomes 1, and a third harmonic voltage component superimposed sine wave mode is used when the modulation rate Am increases further. When Am increases, the overmodulation mode can be used, and when the modulation factor Am further increases, the 1-pulse mode can also be used.

(Modification)
In the above embodiment, the middle level value of the three-phase voltage commands U, V, W of the staircase voltage waveform is set to a constant value of 0 level (duty ratio 50%). For example, as shown in FIG. The level value may be set to a constant value at another level. Also in this case, the phase angle θx from the cross point where the values of the three-phase basic voltage commands Vu, Vv, Vw coincide with the middle level value is calculated to calculate the three-phase voltage commands U, V, W of the staircase voltage waveform. Can be created. The middle level value between the high level value (duty ratio 100%) and the low level value (duty ratio 0%) of each phase voltage of the three-phase voltage commands U, V, W of the staircase voltage is one. Instead of a constant value, a plurality of constant values may be used as shown in FIGS.

(Modification)
Further, in the above embodiment, in the entire transition period between the high level value (duty ratio 100%) and the low level value (duty ratio 0%) of the three-phase voltage commands U, V, W in the overmodulation mode. The middle level period for setting the middle level value, which is one or a plurality of intermediate steady values, is set, but this middle level period is set as a part of the transition period, and the middle level period is set between the high level period and the middle level period. A period during which the voltage changes may be between the low level period and the middle level period.

1 is a block circuit diagram illustrating a configuration of a motor control system according to a first embodiment. FIG. 2 is a block circuit diagram illustrating a detailed configuration of a control circuit in FIG. 1. It is a time chart which shows the waveform of the voltage command of the staircase waveform in overmodulation mode. It is a flowchart which shows operation | movement of the control circuit of FIG. It is a wave form diagram which shows the waveform of the three-phase voltage command of a staircase voltage. It is a time chart which shows a 6n synchronous state. It is a flowchart which shows an example of a synchronous control operation | movement. It is a flowchart which shows an example of switching operation | movement between 1 pulse mode and overmodulation mode. It is a wave form diagram which shows the waveform of a two-phase modulation voltage, a 3rd harmonic superimposed sine wave voltage, and a staircase voltage. It is a wave form diagram which shows the deformation | transformation aspect of the staircase voltage in overmodulation mode. It is a wave form diagram which shows the deformation | transformation aspect of the staircase voltage in overmodulation mode. It is a time chart which shows the conventional pulse missing state.

Explanation of symbols

DESCRIPTION OF SYMBOLS 1 DC power supply 2 Drive device 3 Three-phase synchronous motor 4 Motor current detection means 5 Motor current detection means 6 Motor rotation position detection means 7 Inverter 8 Control circuit 9 Input voltage detection means 81 Motor rotation speed calculation means 82 Axis current generation means 83 Axis Current command generating means 84 Axis voltage command generating means 85 Coordinate axis converting means 86 Modulation rate calculating means 87 Control mode determining means 88 Voltage command generating means 89 Signal generating means 90 Switching gate driver 91 Phase voltage command generating means

Claims (9)

When the amplitude of the voltage command of each phase to be output by pulse width modulation to the gate driver that controls the switching element of the inverter for AC motor drive control is X, and the amplitude of the carrier voltage used for the pulse width modulation is Y, Modulation rate calculation means for calculating X / Y as the modulation rate Am;
When the modulation factor Am is less than 1, the sine wave mode is set. When the predetermined modulation rate value greater than 1 is reached, the 1 pulse mode is set, and the modulation rate Am between the sine wave mode and the 1 pulse mode is set to a predetermined excess. Control mode selection means for selecting a modulation mode;
The input voltage command is output as it is in the sine wave mode, and the input voltage command is converted into a binary pulse voltage that transitions between the maximum value and the minimum value of the carrier voltage in the one-pulse mode. The input phase voltage command has a high level value (duty ratio 100% in pulse width modulation) near the maximum value of the voltage command in the overmodulation mode, and near the minimum value of the phase voltage command. A voltage command generating means for converting to an overmodulation voltage having a low level value (duty ratio 0% in pulse width modulation) and outputting the voltage,
PWM signal generating means for pulse-width modulating the output voltage of the voltage command generating means and outputting it to the gate driver;
With
The voltage command generating means is an inverter control system that extends a high level period in which the high level value is output in the overmodulation mode and a low level period in which the low level value is output as the amplitude of the phase voltage command increases. Because
The voltage command generating means is a substantially constant value separated from the high level value and the low level value by a predetermined value in a transition period that is a period between the high level period and the low level period of the overmodulation mode. An inverter control system comprising a middle level period during which a middle level value is output to the PWM signal generating means. In the inverter control system according to claim 1,
The inverter control system characterized in that the middle level value is set in the vicinity of a median value corresponding to a duty ratio of 50% in the pulse width modulation. In the inverter control system according to claim 1 or 2,
The middle level period occupies the entire transition period. In the inverter control system according to any one of claims 1 to 3,
The voltage command generation means outputs a three-phase voltage command of two-phase modulation to the PWM signal generation means in the sine wave mode. In the inverter control system according to any one of claims 1 to 4,
The inverter control system characterized in that the voltage command generating means outputs a three-phase voltage command having a sine wave waveform on which a third harmonic voltage component is superimposed to the PWM signal generating means. In the inverter control system according to claim 4 or 5,
The voltage command generation means switches between the sine wave mode and the overmodulation mode when the three-phase voltage command in the sine wave mode reaches the high level value, the low level value, and the middle level value. An inverter control system characterized by being implemented in In the inverter control system according to any one of claims 1 to 6,
In the overmodulation mode, the voltage command generating means and the PWM signal generating means are the middle of the voltage command that is one phase at an electrical angle of 0 degrees in the motor in a synchronous state where the synchronization number is 6n (n is a positive integer). Synchronous control is performed so that the temporal median of the level period and the maximum value or minimum value of the carrier voltage of the PWM signal generating means substantially match, and when the number of synchronizations is not 6n (n is a positive integer) An inverter control system characterized by stopping synchronous control. In the inverter control system according to any one of claims 1 to 7,
The voltage command generating means switches the overmodulation mode to the one-pulse mode when the motor rotational speed is higher than a predetermined threshold rotational speed Mref to a predetermined threshold close to 1 with a modulation factor smaller than 1. An inverter control system characterized by being implemented. When the amplitude of the voltage command of each phase to be output by pulse width modulation to the gate driver that controls the switching element of the inverter for AC motor drive control is X, and the amplitude of the carrier voltage used for the pulse width modulation is Y, Modulation rate calculation means for calculating X / Y as the modulation rate Am;
When the modulation factor Am is less than 1, the sine wave mode is set. When the predetermined modulation rate value greater than 1 is reached, the 1 pulse mode is set, and the modulation rate Am between the sine wave mode and the 1 pulse mode is set to a predetermined excess. Control mode selection means for selecting a modulation mode;
The input voltage command is output as it is in the sine wave mode, and the input voltage command is converted into a binary pulse voltage that transitions between the maximum value and the minimum value of the carrier voltage in the one-pulse mode. The input phase voltage command has a high level value (duty ratio 100% in pulse width modulation) near the maximum value of the voltage command in the overmodulation mode, and near the minimum value of the phase voltage command. A voltage command generating means for converting to an overmodulation voltage having a low level value (duty ratio 0% in pulse width modulation) and outputting the voltage,
PWM signal generating means for pulse-width modulating the output voltage of the voltage command generating means and outputting it to the gate driver;
With
The voltage command generating means is an inverter control system that extends a high level period in which the high level value is output in the overmodulation mode and a low level period in which the low level value is output as the amplitude of the phase voltage command increases. Because
The voltage command generating means switches the overmodulation mode to the one-pulse mode when the motor rotational speed is higher than a predetermined threshold rotational speed Mref to a predetermined threshold close to 1 with a modulation factor smaller than 1. An inverter control system characterized by being implemented.

Images