JPH07303302A - Inverter controller - Google Patents

Abstract

PURPOSE:To provide an inverter controller which adopts such a control system that successively fixes conducting states to the terminals of different phases of a load to prescribed values at every different phase period and, at the same time, does not distort much the waveform of the voltage applied across the load. CONSTITUTION:Three-phase signal voltages Su, Sv, and Sw are combined based on the differences between commanded current values iu and iw obtained by means of subtracting circuits 11 and 13 and detected currents values iu' and iw' and phase discriminator 2 discriminates a voltage fixing phase period based on the levels of the voltages Su, Sv, and Sw. Then an adding circuit 7 decides the offset values required at every voltage fixing phase period and a multiplexer 3 forms three-phase PWM signals by selecting the offset values based on the output of the phase discriminator 2 and comparing a triangular-wave voltage T with the voltages Su, Sv, and Sw after adding the selected offset values to the voltage T.

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to an inverter control device, and more particularly to a voltage duty of a phase of an inverter.
The present invention relates to a two-phase modulation type inverter control device for fixing ratio) to 0% or 100% at a predetermined phase.

[0002]

2. Description of the Related Art Conventionally, in a PWM control device for a three-phase inverter, there has been proposed a two-phase modulation method capable of reducing electromagnetic noise and switching loss and improving power conversion efficiency. This two-phase modulation method is a method in which the voltage level of one of the three phases is fixed at a high level or a low level and the remaining two phases are used for modulation.
It is disclosed in P4641075.

In the two-phase modulation method of this publication, the switching elements in the output stage of each phase of the three-phase inverter are controlled to open / close by the current deviation between the current command value of each phase and the detected current value of each phase, and each phase is controlled. The duty ratio of the three-phase voltage applied from the output stage to the terminals of each phase of the three-phase load is controlled. In this case, the generation of an undesired voltage vector is reduced by fixing the current command value of each phase to the maximum value in different predetermined phase periods.

Further, in the two-phase modulation system of the above publication, the power factor of the load is regarded as constant, and the ideal phase voltage corresponding to the current command value can be easily estimated from the output current command value. Then, assuming a period in which the ideal phase voltage estimated from the power factor Θv of the load, which is considered to be a known constant value, peaks, the amplitude of the current command value of that phase is fixed to the maximum value during this period, During the remaining period, the current command value for each phase is a sine waveform. As a result, the voltage of the phase for which the current command value is fixed during the period when the ideal phase voltage peaks is fixed to the maximum value, so the applied voltage of each phase is slightly distorted, but in the form of a three-phase AC voltage. As long as the detected current values of the remaining two phases closely follow the current command value, the detected current value of the phase with the fixed current command value should also be a sine waveform.

[0005]

[Problems to be Solved by the Invention] However, in this way,
Estimate the situation under a certain assumption, and based on this estimation,
The two-phase modulation method of the above publication, which fixes the current of a certain phase in a certain period to the maximum value, has the following problems. Primarily,
The phase angle (also called the phase difference) between the load current applied to the load and the applied voltage to the load is the amount of load current (the internal impedance of the motor changes depending on the amount of load current) and load Since it fluctuates according to the frequency of the applied voltage (the internal impedance of the motor changes depending on the frequency) applied to the load, the voltage applied to the load should peak when the phase difference is assumed to be constant as described above. The phase voltage of the load cannot be fixed to the maximum value during the period, and as a result, the distortion of the voltage waveform applied to the load (motor) increases, resulting in lower efficiency, heat generation, and noise increase. In other words, a good result is obtained with the phase difference assumed to be constant, but depending on the power factor of the motor, the duty cycle of the voltage is increased.
The voltage of the phase is fixed to the maximum value when the ratio should not be set to a high level.

Secondly, the above-mentioned first problem is an explanation when this control system is treated as an AC circuit for processing a sine wave signal. However, in reality, the processing waveform is a non-sinusoidal waveform due to the voltage fixing operation described above. Therefore, in an impedance system including reactance such as a motor, it is necessary to accurately consider the response delay of the transfer function. That is, due to the delay of the detected current based on the transmission phase delay of the control system with respect to the current command value, the phase of the phase voltage and the phase of the detected current deviated from the phase of the current command value of a certain phase by a predetermined (apparent) phase difference. On the other hand, the degree of the phase shift from the phase of the actual phase voltage shifted by the actual phase difference is further increased.

To further explain the above problem, in the conventional two-phase modulation method, the duty ratio of the inverter is controlled by the deviation between the current command value, which is usually a sine waveform, and the detected current (also called the actual current value). , The voltage applied to the motor is controlled. As described above, in the impedance system including the reactance of the motor, the detected current (actual current value) is delayed by a predetermined phase angle with respect to the applied voltage of the sine waveform, and a predetermined response delay is generated in the impulse response. It has a characteristic that the responsiveness of the deviation, that is, the responsiveness of the detected current to the change of the applied voltage is considerably deteriorated.

For this reason, conventionally, the above-mentioned deviation (or a load proportional thereto), which is the difference between the current command value and the detected current, is advanced so that the voltage of a certain phase advances by a predetermined fixed phase difference with respect to the current command value of that phase. The phase of the applied voltage) is shifted, but this is due to the deviation (or the deviation) in consideration of the problem of deviation from the phase difference described above and the problem of impulse response delay of the actual current value described above. Since the phase of (proportional applied voltage) is not shifted, there is a problem that this response delay cannot be compensated in terms of phase. As a result, the distortion of the applied voltage waveform increases, and the actual current value also distorts and noise. However, there is a problem that the efficiency is lowered.

The present invention has been made in view of the above problems, and employs a control system in which the energization state of each phase terminal of the load is sequentially fixed to a predetermined value for each phase period different from each other, and the voltage applied to the load is applied. The purpose is to realize an inverter control device with small waveform distortion. In order to achieve this object, in the present invention, in the inverter in which the duty ratio of the output voltage is determined based on the deviation signal, there is little phase shift between the deviation signal and the output voltage. By utilizing the deviation signal, the timing for fixing the duty ratio is determined.

[0010]

The first structure of the present invention is as follows.
An inverter that applies a voltage of each phase to the terminals of each phase of the multi-phase AC motor, a current detection unit that detects a current that flows to at least two phase terminals of the motor, and a current command value that should be supplied to the motor The current command value generating means for outputting and the opening and closing control of the inverter based on the deviation signal according to the deviation between the detected current and the current command value, and the inverter applies the voltage to each phase terminal of the motor. Signal generating means for determining the waveform of the phase voltage, and for each mode indicating different phase periods determined in relation to the deviation signal, the operating state of the output stage of each phase of the inverter, sequentially,
It is characterized in that it is provided with signal changing means for fixing the signal at a fixed level.

In a second configuration of the present invention, in addition to the first configuration, the signal changing means further sets the phase period based on each phase signal obtained by amplifying the deviation between the detected current and the current command value. It is characterized by determining the mode to be shown. A third configuration of the present invention is characterized in that, in the second configuration, the current command value has a sine waveform.

In a fourth configuration of the present invention, in addition to the first configuration, the signal changing means sets one output state of the output stage of each phase of the inverter for each mode indicating the phase period. It is characterized in that it is fixed to the maximum voltage level or the minimum voltage level, that is, either peak level. In a fifth configuration of the present invention, in addition to the second configuration, the signal changing means further detects a deviation between the detected current and the current command value for each phase, , And the phase of the deviation whose positive and negative signs are opposite to those of the other two phases is fixed.

According to a sixth aspect of the present invention, in addition to the above-mentioned first configuration, the signal changing means sequentially fixes the operating state of the output stage of each phase of the inverter to a constant level, and fixes the operating state. It is characterized in that the voltage waveforms of the other phases are changed according to the amount of According to a seventh aspect of the present invention, in addition to the first configuration, the signal generating means further comprises pulse width modulation signal generating means for determining a waveform of each phase voltage based on a comparison between the deviation signal and a reference waveform. And the signal changing means fixes the voltage waveform by switching the level of the reference waveform,
The feature is that the voltage waveforms of other phases are changed.

According to an eighth aspect of the present invention, in addition to the first configuration, the signal changing means further comprises difference calculating means for calculating a difference between the deviation signal of the fixed phase and the peak value of the reference waveform. A switching means for offsetting the reference waveform according to the difference calculated by the difference calculating means, and based on a comparison between the offset reference waveform and the deviation signal of the other phase, the duty of the other phase It is characterized by generating a signal.

A ninth structure of the present invention is characterized in that, in addition to the above-mentioned first structure, a signal change prohibiting means for prohibiting fixing of the voltage waveform by the signal changing means is further provided. A tenth structure of the present invention has an inverter composed of a switching element, applies a pulse-width-modulated three-phase AC voltage to the motor, and changes the frequency of the three-phase AC voltage. In an inverter control device of a two-phase modulation system, which controls a rotation speed of a motor and further fixes a predetermined switching element among the switching elements to either a full conduction state or a full interruption state in sequence. Means for outputting an accelerator signal for determining the degree of acceleration of the motor, means for detecting the rotational speed of the motor, and current for detecting at least two phase currents flowing from the inverter to the motor. Detecting means, currents of at least two phases which are determined according to the accelerator signal of the cross section for accelerating the motor and the rotational speed of the motor, and which are sine wave signals having mutually different predetermined angular phases. Means for generating a command value, a deviation signal detecting means for detecting a deviation signal of each phase between the current command value and the detected current value detected by the current detecting means, and the current phase of the deviation signal. Phase determination means for generating a mode signal by determining whether or not it corresponds to a mode, means for generating a triangular wave signal, and a shift amount for each mode from the peak value of the triangular wave signal and the deviation signal. Means to do
Means for selecting a specific shift amount from the shift amounts according to the mode signal; means for calculating a pulse width modulation signal from the specific shift amount, the triangular wave signal, and the deviation signal. , The pulse width modulated signal to the inverter
And a gate driving means for making a gate signal given to the computer.

According to an eleventh structure of the present invention, in addition to the tenth structure, the means for selecting the shift amount sets the shift amount to zero, thereby inhibiting the two-phase modulation and performing the three-phase modulation. It is characterized by having means for selecting.

[0017]

According to the inverter control device of the first configuration of the present invention, the waveform of each phase voltage is controlled by opening / closing the inverter based on the deviation signal according to the deviation between the current command value and the detected current. To decide. Further, according to the two-phase modulation method of the present invention, the operating states of the output stages of the respective phases of the inverters of the respective phases are sequentially fixed to a constant level for different phase periods according to the deviation signal.

Therefore, the current command due to the fluctuation of the phase difference between the applied voltage (phase voltage) and the energized current (detected current) and the electric response delay of the motor, which has been a serious problem in the conventional technology. It is possible to significantly reduce the occurrence of a problem due to the phase shift between the value and the detected current. Further, since the phase period is determined based on the waveform of the deviation including the response delay of the detected current with respect to the current command value, and the voltage fixing control is performed for each period, the influence of the response delay can be reduced.

According to the second configuration of the present invention, in the first configuration, the signal changing means further includes the output state of each phase of the inverter, that is, the output voltage value, based on the deviation between the detected current and the current command value. Determine the phase period to fix. With this configuration, since the waveform of the voltage of each phase is originally determined by the deviation, the circuit configuration becomes complicated by determining the phase period for fixing the voltage of each phase based on the deviation. The voltage can be fixed without causing a phase shift between the original waveform of the voltage of each phase and the voltage-fixed waveform, and eventually a two-phase modulation system without a phase shift is realized.

According to the third structure of the present invention, in the first structure, the current command value has a sinusoidal waveform which does not require the voltage fixing operation as disclosed in the prior art. This facilitates the process of combining the current command values. According to the fourth configuration of the present invention, in the first configuration, the voltage of each phase is fixed to the maximum voltage level or the minimum voltage level for each phase period. Since the voltage of each phase has a small rate of change in amplitude at its peak, the distortion of the voltage of each phase output from the inverter is small. In a further preferred mode, six voltage-fixing phase periods are set in one cycle for the three-phase AC voltage. The phase period for voltage fixing is applied to each peak period (π / 3 or less) of the maximum value or the minimum value (which means a negative maximum value in terms of AC) of the original voltage waveform of each phase that is a sine waveform. Is set. In this way, the two-phase modulation method can be realized while reducing the distortion of the three-phase AC voltage.

According to the fifth configuration of the present invention, the voltage fixed phase and its phase period can be more easily determined in the second configuration. According to the sixth aspect of the present invention,
Further, in the first configuration, since the voltage waveforms of the other phases are changed according to the amount for fixing the fixed phase, the voltage change amount of the fixed phase can be supplemented by the voltage waveforms of the other phases. Therefore, the controllability of the motor can be improved.

According to the seventh structure of the present invention, in the first structure, the voltage waveform of each phase is determined by determining the voltage waveform based on the reference waveform such as a triangular wave and switching the level of the reference waveform. Since it is changed, it is possible to easily and reliably switch the voltage for each phase at the same time, so that the voltage change amount of the fixed phase can be compensated by the voltage waveforms of the other phases.

According to the eighth structure of the present invention, the voltage waveform is changed by further offsetting the reference waveform in the seventh structure, and the voltage waveform of each phase can be easily changed. . According to the ninth configuration of the present invention, in addition to the first configuration, a signal change inhibiting means for inhibiting the fixing of the voltage waveform is further provided, so that the two-phase modulation system is switched to the three-phase modulation system at any timing. Can be controlled.

According to a tenth structure of the present invention, the first
The inverter control device of the independent invention can be realized easily and surely. According to the eleventh configuration of the present invention, the inverter control device having the tenth configuration can be easily realized.

[0025]

DESCRIPTION OF THE PREFERRED EMBODIMENTS An electric vehicle using the inverter control device of the present invention will be described below with reference to the drawings.
This electric vehicle 60, as shown in FIG. 1, a main battery 61, an inverter 62, an accelerator sensor 63, a control unit ECU 64, a gear 65, a star-connected cage type induction motor (hereinafter also simply referred to as a motor) 66, a rotation sensor. 67, suspension 68, running tire 69
Equipped with.

As shown in FIG. 2, the inverter 62 includes six transistors (IGBT) which are switching elements.
620-625 and three negative logic elements 626-628
And a smoothing capacitor 629, the ECU
It is controlled by control signals U, V, W of each phase output from (inverter control device in the present invention) 64. The waveforms of the control signals U, V, W of each phase are formed by controlling the duty ratio (pulse width / pulse period) of the pulse signal of a predetermined carrier frequency.

An example of the ECU 64 will be described with reference to FIG. Reference numeral 11 is a U-phase deviation calculator that calculates the deviation between the U-phase current command value iu and the actual current value (detected current value) iu ′ detected by the current sensor 8, and similarly 13 is the W-phase deviation calculator. The current command value iw and the actual current value i detected by the current sensor 8
It is a W-phase deviation calculator that calculates the deviation from w '.

Reference numeral 21 denotes a U-phase PI calculator for calculating the deviation calculated by the U-phase deviation calculator 11, which outputs a U-phase signal voltage S _u . Similarly, 23 is the W-phase deviation calculator 13
The deviation calculated by a W-phase PI calculation part for PI calculation, and outputs the W-phase signal voltage S _w. The V-phase signal voltage S _v given to the V-phase is obtained by calculating the sum of the U-phase signal voltage S _u and the W-phase signal voltage S _w and inverting the sign by the inverting adder 29.

Further, in this embodiment, the ECU 64 includes a phase determiner 2 for determining the phase states of the applied voltages Vu, Vv, Vw of the respective phases applied to the induction motor 66 based on the signal voltages Su, Sv, Sw. , An adder circuit 7, a multiplexer 3 that generates an offset voltage OFV based on the determination signal from the phase determiner 2 and the signal from the adder circuit 7, a triangular wave voltage generation circuit 5, an adder 6, and a controller 640. , A gate drive circuit 34 and a current detector 9.

The controller 640 has a frequency and an amplitude determined according to an accelerator signal from an accelerator sensor (not shown) and a rotation speed of the motor 66 from the rotation sensor 67, and the phases are 120 degrees different from each other. Signal or current signal) current command values iu, iw
This is a circuit for generating a signal, and its gist is different from that of the present invention.

The triangular wave voltage generating circuit 5 is a circuit for generating a triangular wave voltage (which may be a sawtooth wave voltage) T having a predetermined carrier (sampling) frequency. The adder 6 is an adder circuit that adds the triangular wave voltage T and the offset voltage OFV and applies the result to the + input terminals of the comparators 31 to 33.
The current detector 9 amplifies a signal corresponding to the U-phase and W-phase currents supplied to the motor 66 detected by the current sensor 8 to a magnitude comparable to the current command values iu and iw.

Next, the operation of the circuit shown in FIG. 3 will be described. Deviations between the U-phase current command value iu and the W-phase current command value iw and the actual current values iu ′ and iw ′ are calculated for each phase and PI amplified to obtain the U-phase and W-phase signal voltages Su and Sw. V-phase signal voltage Sv is formed by adding and inverting them.
Then, the respective phase signal voltages Su, Sv, Sw are applied to the-input ends of the comparators 31-33, respectively. Note that only the low frequency component of the deviation may be taken out through the filter circuit and PI amplified. Further, only the low frequency component of the detected current may be similarly extracted. Incidentally, it is preferable to cut off a high frequency component of 6 times or more, preferably 4 times or more of the frequency of the current command value from the deviation or the detected current. What is important here is that the voltage level is not fixed or Even if the applied voltage V of each phase of the motor
When the applied voltages Vu, Vv, Vw of the respective phases are sequentially fixed to the maximum value or the minimum value for each predetermined phase period in accordance with the peak period or the bottom period of u, Vv, Vw, each of the voltage-fixed processing is performed. Distortion of the applied voltages Vu, Vv, Vw of the phases is not so great, and it can be regarded as an approximately three-phase AC voltage and processed. That is, the signal voltages Su, Sv, Sw corresponding to the above deviations are the phase voltages V of the motor applied voltage.
That is, the signals have the same phase as u, Vv, and Vw. Therefore, if the phase relationship of the signal voltages Su, Sv, and Sw is discriminated to determine the period for fixing the voltage level, the phase shift as in the conventional case is small and the applied voltage V of each phase is small.
The waveform distortion of u, Vv, and Vw should be able to be reduced significantly compared with the conventional one. Based on this knowledge, in this embodiment, the phase determiner 2 determines the phase relationship between the signal voltages Su, Sv, Sw, and the multiplexer 3 generates the offset voltage OFV for fixing the voltage level.

Hereinafter, a method of determining the phase by the phase determiner 22 and a method of fixing the voltage by the offset voltage OFV from the multiplexer 3 will be separately described. (Phase Judgment Operation) With respect to the phase judgment operation by the phase judgment device 2, an idealized model will be described for simplification of description.

Now, assuming that the influence of the voltage fixation can be ignored from the detected currents (actual current values) iu 'and iw',
Since the current command values iu and iv are sine waves having a phase difference of 120 degrees and equal amplitudes, the detected currents iu ′ and iw ′ can also be regarded as sine waves, and as a result, the three-phase signal voltages Su, Sv, Sw.
Can be roughly regarded as a sine wave. FIG. 4A shows ideal waveforms of the three-phase signal voltages Su, Sv, Sw.

From the amplitude states of these three-phase signal voltages Su, Sv, Sw, one cycle can be divided into six periods of modes 1 to 6. That is, as shown in FIG. 4B, depending on whether the instantaneous values of the AC components of the three-phase signal voltages Su, Sv, and Sw are positive or negative, there are six modes as shown in FIG. 4B. Recognize. Since each of the three phases is represented by 0 or 1, all modes are represented by a 3-bit signal.

There are eight states represented by 3 bits, and in addition to FIG. 4B, there may be cases where all the phases are positive and where all the phases are negative. The code representing the mode of will be used for another purpose and will be described later. Further, the phase determiner 2 for generating the 3-bit signal has three-phase signal voltages Su, Sv,
It can be constituted by three comparators (not shown) that compare Sw with a reference voltage corresponding to an instantaneous value of zero.

Since the detected currents iu 'and iw' are distorted, each mode 1 actually deviates from the above ideal model, but this is because the three-phase signal voltages Su, Sv, Sw include the above distortion. However, it is rather convenient in that the voltage can be fixed according to the waveforms of the actual three-phase signal voltages Su, Sv, Sw. (Fixed voltage operation) During the period of each mode determined by the phase determiner 2, different phase voltages among the applied voltages Vu, Vv, Vw of each phase are sequentially selected, and the selected voltage is set to the maximum value or the minimum value. The fixing operation will be described below.

In this embodiment, the two-phase modulation system disclosed in "Inverter control device (Japanese Patent Application No. 5-262665)" filed on Oct. 20, 1993 by the present applicant is adopted. Specifically, the three-phase signal voltages Su, Sv,
Offset voltage OFV of the triangular wave voltage T compared with Sw
By changing, the voltage is fixed, that is, two-phase modulation is performed.

That is, when only the triangular wave is originally applied to the input of the comparator, the offset voltage (shift amount) OFV is added to the triangular wave to make the input of the comparator, and the duty of the comparator output is added. -The ratio is fixed at a duty ratio of 100% or 0% for the particular phase.
The relationship between the phase to be fixed and the mode is shown in the parentheses in FIG.

The two-phase modulation method will be described below. As shown in FIG. 5A, also in the conventional three-phase modulation, the sinusoidal phase signal voltage of each phase is compared with the triangular wave voltage, and the PWM waveform is generated according to which level is higher. . FIG. 5B is a partially enlarged view of FIG. For simplicity, the output S of the U-phase PI arithmetic unit is actually a curve.
Let u be represented by a straight line a in FIG. 5B, and let this potential be Va. At this time, in order to perform the two-phase modulation by setting the duty ratio of the PWM signal to 100% as in the present invention, the offset voltage is added so that the positive peak value of the triangular wave (the output of the adder 6 in FIG. 3) is Va. Just fix it. Therefore, when the shift amount is ΔVa, the shift amount ΔVa is obtained by the following equation. Note that Vp is a positive peak value of the triangular wave voltage T, and -Vp is its negative peak value.

In other words, as shown in FIG. 5A, when the positive peak value of the triangular wave slightly exceeds the level of the U phase, the duty ratio is made large, and the negative peak value of the triangular wave becomes large. The duty ratio is made small when the level of the U-phase voltage is slightly exceeded. Therefore, in order to set the duty ratio to 100%, the positive peak value of the triangular wave should not exceed the level of the U phase, and in order to set it to 0%, the negative peak value of the triangular wave should be set as follows. It is advisable not to exceed the level of the U phase.

[0042]

## EQU1 ## .DELTA.Va = Va-Vp = Va + (-Vp) That is, the shift amount .DELTA.V at which the duty ratio is 100%.
a is the negative peak value -Vp of the triangular wave voltage T and the PI output Su
It is calculated by adding and. Next, the output Sv of the V-phase PI calculator
Is represented by a straight line b, and this potential is Vb. In this case, the shift amount Δ for which the duty ratio should be 0%
When Vb is calculated, it can be calculated by the following equation as in the above case.

[0043]

## EQU2 ## ΔVb = Vb − (− Vp) = Vb + Vp, that is, the shift amount ΔVb is the positive peak value V of the triangular wave T.
It is obtained by adding b and the voltage Vp of the PI output Sv. The same applies to the other phases, and the duty ratio is 10
PI output Sv, Sw is 0% or the shift amount fixed to 0%
It can be seen that it can be obtained by adding the potential of (3) and the ± peak value (Vp or -Vp) of the triangular wave T. This is shown in the table for each mode in FIG.

The adder 7 for performing this addition can be realized by a circuit composed of a simple resistor set as shown in FIG. 6, and if the shift amounts of all the modes 1 to 6 described above are obtained and an appropriate shift amount is selected. I know it's good. The shift amount for each mode is as shown in FIG. The code in FIG. 7 is the code shown in FIG.
This is bit information and indicates one of modes 1 to 6.
From among these, the multiplexer 3 selects the shift amount suitable for the mode. For example, when the mode is [011], the multiplexer 3 selects the input terminal 2 and the shift amount is Su-Vp.

The shift amount (offset voltage) OFV output from the multiplexer 3 is input to the adder 6,
By adding to the triangular wave voltage T by the adder 6, the corrected triangular wave voltage having an offset is synthesized. The triangular wave voltage T on which the offset voltage OFV is superimposed is
The three-phase signal voltage S is input to the comparators 31 to 33.
u, Sv, Sw are individually compared and converted into a duty ratio, that is, a PWM signal.

In this way, the duty ratio of the comparator 32 is 0 in mode 1 as shown in the table of FIG.
%, That is, fixed to a low level, the duty ratio of the comparator 31 is fixed to 100%, that is, a high level in the mode 2, and the comparator 33 in mode 3
The duty ratio of is fixed at 0%, that is, low level,
In mode 4, the duty ratio of the comparator 32 is 1
The duty ratio of the comparator 31 is fixed to 0%, that is, the high level in mode 5, and the duty ratio of the comparator 31 is fixed to 0%, that is, the low level in mode 5.
The duty ratio of is fixed to 100%, that is, a high level, and a two-phase modulation system having six different voltage fixing patterns is realized.

As a result, the voltage command by the PWM signal is applied to the three-phase inverter 62 via the gate drive circuit 34, and the input current of the motor 66 flows according to the current command value. The output of the multiplexer 3 (offset voltage O
The FV) is stepwise, and the duty ratio of the comparator may suddenly change to 100% or 0%, which may adversely affect the motor 66. In order to prevent this, the output of the multiplexer 3 may be blunted by a low pass filter or an integrating circuit.

In the control of the three-phase inverter 62, the two-phase modulation is not effective in all cases, and it may be better to control the three-phase modulation. In such a case, switching between two-phase modulation and three-phase modulation can be easily performed in the embodiment described below. In this embodiment, a method of shifting a triangular wave at the stage of creating a PWM waveform of two-phase modulation is used.
That is, when this shift amount is set to 0, there is no difference from the motor control by the PWM waveform of the general three-phase modulation method. Therefore, on the output side of the phase determiner 2, as shown in FIG.
One AND circuit 44, 45, 46 is provided, and a circuit 47 to which the output of the phase determiner 2 is input is provided for each AND circuit 44, 45, 46. In this circuit 47, the AND circuits 44, 45, and 46 are connected to a power supply (not shown) via a resistor 48, or a "Lo" signal is connected to the ground via a switch SW.
The "w" signal is input. Therefore, by turning on the switch SW in FIG. 26 according to a command from the ECU 10, the “Low” signal can be input to the three AND circuits 44, 45, 46 to output the code signal [000]. In the multiplexer 3, a mode in which the shift amount is 0, that is,
The mode that does not provide the offset phase is [00
0] and the mode of the shift amount 0 is selected in advance, switching between three-phase modulation and two-phase modulation can be performed at arbitrary timing.

According to the above configuration, a waveform with less distortion can be obtained regardless of fluctuations in the power factor of the load due to changes in frequency and current, and the effect of reducing switching loss due to two-phase modulation can be obtained in all regions. be able to. In the above embodiment, an example has been described in which the triangular wave that is one input of the comparator is offset and two-phase modulated, but the three-phase signal voltage S that is the other input of the comparator is used.
It is also possible to give an offset to each of u, Sv, and Sw for two-phase modulation. Hereinafter, this embodiment will be described with reference to FIG.

The circuit of FIG. 9 corresponds to the multiplexer 3 of FIG.
The configurations of the adder circuit 7 and the adder 6 are changed. 7
1 to 76 are adders. Reference numeral 71 is a circuit that adds the positive peak value + Vp of the triangular wave voltage T and the signal voltage Sw to generate the shift amount in the mode 3 of FIG.

Reference numeral 72 denotes a negative peak value -Vp of the triangular wave voltage T.
And a signal voltage Sw are added to generate a shift amount in the mode 6 of FIG. 73 adds the positive peak value + Vp of the triangular wave voltage T and the signal voltage Sv to obtain the mode 1 of FIG.
Is a circuit for generating the shift amount of. 74 adds the negative peak value −Vp of the triangular wave voltage T and the signal voltage Sv, and FIG.
It is a circuit for generating the shift amount of mode 4 of.

75 is a positive peak value of the triangular wave voltage T + Vp
And a signal voltage Su are added to generate a shift amount in the mode 5 of FIG. 74 adds the negative peak value −Vp of the triangular wave voltage T and the signal voltage Su to obtain the mode 2 of FIG.
Is a circuit for generating the shift amount of. 3a, 3b, and 3c are multiplexers.

The multiplexer 3a sends the output of the adder 71 to the adder 61 when the 3-bit output signal from the phase determiner 2 is 001, and when the 3-bit output signal from the phase determiner 2 is 110. The output of the adder 72 is added to the adder 6
Send to 1, turn off otherwise. As a result, the adder 6
1 outputs the triangular wave voltage T shifted by the offset voltage OFV required in the modes 3 and 6 to the comparator 33.

The multiplexer 3b sends the output of the adder 73 to the adder 62 when the 3-bit output signal from the phase determiner 2 is 010, and when the 3-bit output signal from the phase determiner 2 is 101. The output of the adder 74 to the adder 6
Send to 2, turn off otherwise. As a result, the adder 6
2 outputs to the comparator 32 the triangular wave voltage T shifted by the offset voltage OFV required in the modes 1 and 4.

The multiplexer 3c sends the output of the adder 75 to the adder 63 when the 3-bit output signal from the phase determiner 2 is 100, and when the 3-bit output signal from the phase determiner 2 is 011. The output of the adder 76 is added to the adder 6
Send to 3 and turn off otherwise. As a result, the adder 6
3 outputs to the comparator 32 the triangular wave voltage T shifted by the offset voltage OFV required in the modes 5 and 2.

Then, the output voltages of the comparators 31 to 33 are fixed at the high level or the low level only during the period in which the triangular wave voltage T on which the offset voltages OFV are superposed is applied to the minus input terminal, and the triangular wave is fixed in the other periods. Voltage T
Accordingly, the three-phase signal voltages Su, Sv, Sw are converted into PWM signals.

[Brief description of drawings]

FIG. 1 is a block diagram of an electric vehicle that employs an inverter control device according to an embodiment of the present invention.

FIG. 2 is a circuit diagram of the inverter shown in FIG.

FIG. 3 is a block circuit diagram mainly of an ECU (inverter control device in the present invention) of FIG.

FIG. 4A is a diagram showing the relationship between the waveform of the signal voltage of each phase and the mode in the above embodiment. (B) is a diagram showing a code signal of bits representing a mode.

FIG. 5 is a diagram for explaining modulation by PWM, and FIG. 5A is a diagram showing a relationship between a waveform of a signal voltage of each phase and a triangular wave and a modulated pulse train according to a conventional method. FIG. 5B is a partial enlarged view of FIG. 5A corresponding to the partially enlarged schematic view of FIG. 5A, for explaining the shift of the triangular wave in the above one embodiment.

FIG. 6 is a diagram showing details of the adder shown in FIG. 3;

FIG. 7 is a diagram showing a relationship between each mode and a shift amount and a duty ratio of a comparator output in the above one embodiment.

FIG. 8 is a partial electric circuit diagram showing an example in which a switching circuit for switching between a two-phase modulation system and a three-phase modulation system is added according to the second embodiment of the present invention.

FIG. 9 is a partial electric circuit diagram showing a main part of the third embodiment.

[Explanation of symbols]

 66 motor (multi-phase AC motor) 62 three-phase inverter (inverter) 8 current sensor (current detecting means) 640 controller (current command value generating means) 31-33 comparator (signal generating means) 2 phase determiner (of signal changing means) Part 3) Multiplexer (part of signal changing means) 6 Adder (part of signal changing means) 7 Adder circuit (part of signal changing means)

Claims (11)

[Claims] 1. An inverter for applying a voltage of each phase to a terminal of each phase of a multi-phase AC motor, a current detecting means for detecting a current supplied to at least two phase terminals of the motor, and an electric current for energizing the motor. Current command value generating means for outputting a current command value, and opening / closing control of the inverter based on a deviation signal corresponding to a deviation between the detected current and the current command value so that the inverter is a terminal for each phase of the motor. Signal generating means for determining the waveform of the voltage of each phase to be applied to the inverter, and the operation of the output stage of each phase of the inverter for each mode indicating different phase periods determined in relation to the deviation signal. An inverter control device, comprising: a signal changing unit that sequentially fixes states to a constant level. 2. The signal changing means determines the mode indicating the phase period based on each phase signal obtained by amplifying the deviation between the detected current and the current command value. Inverter control device. 3. The inverter control device according to claim 2, wherein the current command value has a sine waveform. 4. The signal changing means fixes one output state of an output stage of each phase of the inverter to a maximum voltage level or a minimum voltage level, that is, either peak level for each mode indicating the phase period. The inverter control device according to claim 1, which is configured to: 5. The signal changing means detects a deviation between the detected current and the current command value for each phase, and among the deviations for each phase, the deviations of the other two phases are positive and negative. The inverter control device according to claim 2, wherein opposite phases of deviation are fixed. 6. The signal changing means sequentially fixes the operating state of the output stage of each phase of the inverter to a constant level, and changes the voltage waveforms of the other phases according to the fixed amount. The inverter control device according to claim 1, which is one. 7. The signal generating means comprises pulse width modulation signal generating means for determining the waveform of each phase voltage based on a comparison between the deviation signal and a reference waveform, and the signal changing means includes the reference waveform of the reference waveform. The inverter control device according to claim 1, wherein the voltage waveform is fixed by switching the level, and the voltage waveforms of other phases are changed. 8. The signal changing means calculates a difference between a deviation signal of a fixed phase and a peak value of the reference waveform, and the reference according to the difference calculated by the difference calculating means. 8. The inverter control device according to claim 7, further comprising switching means for offsetting the waveform, the duty signal for the other phase being generated based on a comparison between the offset reference waveform and the deviation signal for the other phase. . 9. The inverter control device according to claim 1, further comprising signal change prohibiting means for prohibiting fixing of the voltage waveform by the signal changing means. 10. An inverter composed of a switching element.
A pulse-width-modulated three-phase AC voltage is applied to the motor, and the frequency of the three-phase AC voltage is changed to control the rotation speed of the motor. In a two-phase modulation type inverter control device for sequentially fixing a predetermined switching element to either a full conduction state or a full interruption state, a means for outputting an accelerator signal for determining an acceleration degree of the motor, Means for detecting the number of rotations of the motor; current detecting means for detecting the current of at least two phases flowing from the inverter to the motor; and the accelerator signal of the cross section for accelerating the motor. Means for generating current command values of at least two phases, which are determined according to the number of rotations of the motor, and which are sinusoidal signals having mutually different predetermined angular phases, the current command value and the current detection means. Deviation signal detecting means for detecting a deviation signal of each phase from the detected current value detected by the above, and phase judging means for judging which mode the phase of the deviation signal corresponds to at present and generating a mode signal. A means for generating a triangular wave signal; a means for calculating a shift amount for each mode from the peak value of the triangular wave signal and the deviation signal; and a means for calculating the shift amount according to the mode signal. Means for selecting a specific shift amount, means for calculating a pulse width modulation signal from the specific shift amount, the triangular wave signal, and the deviation signal, and a gate for applying the pulse width modulation signal to the inverter. An inverter control device comprising: a gate drive means for providing a gate signal. 11. The means for selecting the shift amount sets the shift amount to zero to inhibit the two-phase modulation and
11. The inverter control device according to claim 10, further comprising means for selecting a phase modulation.