JP2005312120A - Switching power supply circuit - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To solve the problem of increasing ripple by including an arrangement for improving power factor in a switching power supply. <P>SOLUTION: In a compound resonance converter combining a partial resonance voltage circuit for a half-bridge coupling system current resonance converter, power factor is improved by a power factor improving circuit 10 of such an arrangement as the primary series resonance current obtained in a primary series resonance circuit is regenerated as power and fed back to a smoothing capacitor Ci (power regeneration system). Under such an arrangement, flux density of a power isolation transformer PIT is set not higher than a predetermined level so that the rectification operation current of a secondary both wave rectification circuit is in a continuous mode regardless of load variation thus suppressing increase in ripple voltage of the secondary DC output voltage caused by power regeneration. Furthermore, the number of components is decreased by sharing a diode performing rectification operation and a diode performing switching operation for a commercial AC power supply. A higher power factor can be attained by an arrangement where a power factor improving capacitor C30 is added to the power factor improving circuit. <P>COPYRIGHT: (C)2006,JPO&NCIPI

Description

  The present invention relates to a switching power supply circuit provided as a power supply for various electronic devices.

The present applicant has previously proposed various power supply circuits including a resonance type converter on the primary side. Various power supply circuits configured with a power factor correction circuit for improving the power factor of the resonant converter have also been proposed.
10 and 11 are circuit diagrams each showing an example of a switching power supply circuit configured based on the invention previously filed by the present applicant.

  First, the power supply circuit of FIG. 10 is configured to include a power factor correction circuit 20 for a self-excited current resonance type switching converter. The switching converter in this power supply circuit is a switching converter that combines a current resonance type converter using a half-bridge coupling method and a partial voltage resonance circuit that performs voltage resonance only when the semiconductor switch (switching element) is turned off.

The power supply circuit shown in FIG. 10 includes a bridge rectifier circuit Di that performs full-wave rectification on the commercial AC power supply AC. In this case, the rectified output rectified by the bridge rectifier circuit Di is charged to the smoothing capacitor Ci via the power factor correction circuit 20, and both ends of the smoothing capacitor Ci correspond to a level equal to the AC input voltage VAC. The rectified and smoothed voltage Ei is obtained.
The power factor correction circuit 20 will be described later.

The power supply circuit includes a self-excited current resonance converter that uses a rectified and smoothed voltage Ei that is a voltage across the smoothing capacitor Ci as an operation power supply.
In this current resonance type converter, as shown in the figure, the switching elements Q1 and Q2 by two bipolar transistors are half-bridge coupled and then connected so as to be inserted between the positive side of the smoothing capacitor Ci and the primary side ground. Has been.
Starting resistors RS1 and RS2 are inserted between the collectors and bases of the switching elements Q1 and Q2, respectively. The resistors RB1 and RB2 connected to the bases of the switching elements Q1 and Q2 set the base current (drive current) of the switching elements Q1 and Q2.
Clamp diodes DD1 and DD2 are inserted between the bases and emitters of the switching elements Q1 and Q2, respectively. The clamp diodes DD1 and DD2 form a current path of a clamp current that flows through the base-emitter during the period when the switching elements Q1 and Q2 are turned off, respectively.
The resonance capacitors CB1 and CB2 together with drive windings NB1 and NB2 of the drive transformer PRT described below form a series resonance circuit (self-oscillation drive circuit) for self-oscillation, and switching elements Q1,. Determine the switching frequency of Q2.

The drive transformer PRT (Power Regulating Transformer) in this case is provided for driving the switching elements Q1 and Q2 and performing constant voltage control by variably controlling the switching frequency. The drive transformer PRT in the case of this figure is an orthogonal saturable reactor in which the drive windings NB1 and NB2 are wound and the control winding NC is wound in a direction perpendicular to the respective windings. ing.
One end of the drive winding NB1 of the drive transformer PRT is connected to the base of the switching element Q1 via a series connection of a resistor RB1 and a resonance capacitor CB1. The other end side of the drive winding NB1 is a tap point that is a connection point with the end of the resonance current detection winding ND. The other end (tap point) of the drive winding NB1 is the switching element Q1. Connected to the emitter.
One end of the drive winding NB2 is grounded, and the other end is connected to the base of the switching element Q2 via a series connection of a resistor RB2 and a resonance capacitor CB2.
The drive winding NB1 and the drive winding NB2 are wound so that voltages having opposite polarities are generated.

An insulating converter transformer PIT (Power Isolation Transformer) transmits the switching outputs of the switching elements Q1 and Q2 to the secondary side.
One end of the primary winding N1 of the insulating converter transformer PIT is connected to the contact (switching output point) of the emitter of the switching element Q1 and the collector of the switching element Q2 via the resonance current detection winding ND, so that the switching output Will be obtained.
The other end of the primary winding N1 is connected to the cathode of the fast recovery diode D1 in the power factor correction circuit 20 through the series resonance capacitor C1.

  In this case, the series resonant capacitor C1 and the primary winding N1 are connected in series, but the capacitance of the series resonant capacitor C1 and the leakage inductance of the insulating converter transformer PIT including the primary winding N1 (series resonant winding) ( A primary side current resonance circuit for making the operation of the switching converter into a current resonance type is formed by the leakage inductance component L1.

A partial resonance capacitor Cp is connected in parallel with the collector-emitter of the switching element Q2.
By connecting the partial resonance capacitor Cp, the voltage resonance operation can be obtained only when the switching elements Q1 and Q2 are turned off by the capacitance of the partial resonance capacitor Cp and the leakage inductance component L1 of the primary winding N1. That is, a partial voltage resonance circuit is formed.

On the secondary side of the insulating converter transformer PIT in this figure, a center tap is provided for the secondary winding N2, and rectifier diodes DO1, DO2, DO3, DO4 and smoothing capacitors CO1, CO2 are connected as shown in the figure. As a result, two sets of both-wave rectifier circuits are provided, each consisting of a set of [rectifier diodes DO1, DO2, smoothing capacitor CO1] and a set of [rectifier diodes DO3, DO4, smoothing capacitor CO2]. A double-wave rectifier circuit composed of [rectifier diodes DO1, DO2, smoothing capacitor CO1] generates a DC output voltage EO1, and a double-wave rectifier circuit composed of [rectifier diodes DO3, DO4, smoothing capacitor CO2] generates a DC output voltage EO2. To do.
In this case, the DC output voltage E01 and the DC output voltage E02 are also branched and input to the control circuit 1. In the control circuit 1, the DC output voltage EO1 is used as a detection voltage, and the DC output voltage EO2 is used as an operating power source.
For the rectifier diodes DO1 and DO2 and the rectifier diodes DO3 and DO4, for example, Schottky diodes are selected in accordance with the high-speed switching operation (rectification operation) corresponding to the switching cycle.

  The control circuit 1 performs constant voltage control by supplying, as a control current, a direct current whose level is varied according to the level of the secondary side direct-current output voltage E01, for example, to the control winding NC of the drive transformer PRT. .

As a switching operation of the power supply circuit having the above configuration, when commercial AC power is first turned on, for example, a starting current is supplied to the bases of the switching elements Q1 and Q2 via the starting resistors RS1 and RS2, for example. If the switching element Q1 is turned on first, the switching element Q2 is controlled to be turned off. Then, as an output of the switching element Q1, a resonance current flows from the resonance current detection winding ND → the primary winding N1 → the series resonance capacitor C1, but when the resonance current becomes 0, the switching element Q2 is turned on and the switching element Q1 is turned on. Controlled to be off. Then, a resonant current in the opposite direction flows through the switching element Q2. Thereafter, a self-excited switching operation in which the switching elements Q1, Q2 are alternately turned on is started.
In this way, the switching elements Q1 and Q2 alternately open and close using the terminal voltage of the smoothing capacitor Ci as the operating power supply, thereby supplying a drive current close to the resonance current waveform to the primary winding N1 of the insulating converter transformer PIT. An alternating output is obtained at the next winding N2.

As described above, the control circuit 1 is determined by supplying, for example, a DC current whose level is variable according to the level of the secondary side DC output voltage E01 as the control current to the control winding NC of the drive transformer PRT. Perform voltage control.
That is, by passing a control current according to the level of the DC voltage output EO1 through the control winding NC, the inductances of the drive windings NB1 and NB2 are changed, thereby changing the inductance value forming the self-excited oscillation circuit. To be. If the inductance value is varied in this way, the oscillation frequency of the self-excited oscillation circuit changes, and as a result, the switching frequency is variably controlled. Thus, the drive current supplied to the primary winding N1 of the primary side series resonance circuit is controlled by variably controlling the switching frequency of the switching elements Q1, Q2 in accordance with the level of the DC output voltage EO1. By controlling the energy transmitted to the secondary side, constant voltage control of the secondary side DC output voltage is achieved.
Hereinafter, the constant voltage control method using the above method will be referred to as a “switching frequency control method”.

Next, the configuration of the power factor correction circuit 20 will be described.
This power factor correction circuit 20 adopts a configuration as a power regenerative system by a magnetic coupling type.
In the power factor correction circuit 20, a filter choke coil LN, a fast recovery diode D1, and a high frequency inductor L10 are connected in series between the positive output terminal of the bridge rectifier circuit Di and the positive terminal of the smoothing capacitor Ci. .
The filter capacitor CN is inserted between the anode side of the fast recovery diode D1 and the positive terminal of the smoothing capacitor Ci, thereby forming a normal mode low-pass filter together with the filter choke coil LN.
Further, since the parallel resonant capacitor C20 is provided in parallel with the high frequency inductor L10, the parallel resonant capacitor C20 and the high frequency inductor L10 constitute a parallel resonant circuit. Thereby, it has the effect | action which suppresses the raise of the rectification | straightening smoothing voltage Ei when a load becomes light.

  The power factor correction circuit 20 is connected to the above-described primary-side current resonance circuit (N1, C1) with respect to the connection point between the cathode of the fast recovery diode D1 and the choke coil LS. The current / voltage by the switching output obtained in the resonance circuit is fed back.

In such a power factor correction circuit 20, the primary side series resonance current flowing in the primary side series resonance circuit (C 1 -N 1 (L 1)) is regenerated as electric power according to the switching output of the switching elements Q 1 and Q 2, By feeding back to the smoothing capacitor Ci through the parallel connection of the inductance L10 // capacitor C20, the fast recovery diode D1 is switched when the positive / negative absolute value of the AC input voltage VAC is 1/2 or more. Will work.
As a result, the charging current to the smoothing capacitor Ci flows even during a period in which the rectified output voltage level is lower than the voltage across the smoothing capacitor Ci.
As a result, the average waveform of the AC input current approaches the waveform of the AC input voltage, and the conduction angle of the AC input current is expanded. As a result, the power factor is improved.

Next, FIG. 11 shows another configuration example of the switching power supply circuit configured based on the invention previously filed by the present applicant.
The power supply circuit shown in FIG. 11 is also provided with a current resonance type converter in which two switching elements are half-bridge coupled, and the drive system is a separate excitation type. Also in this case, the power factor improving circuit 21 for improving the power factor is provided.
Note that the same parts as those in FIG.

In the primary side current resonance type converter shown in this figure, for example, a MOS-FET is selected as the two switching elements Q1 and Q2 connected by the half bridge coupling method.
Here, the drain of the switching element Q1 is connected to the line of the rectified smoothing voltage Ei, the source of the switching element Q1 and the drain of the switching element Q2 are connected, and the source of the switching element Q2 is connected to the primary side ground. The half-bridge connection is compatible with the excitation type.
These switching elements Q1 and Q2 are switched and driven by the oscillation drive circuit 2 so that the on / off operation is alternately repeated, and the rectified and smoothed voltage Ei is intermittently used as a switching output.
In this case, clamp diodes DD1 and DD2 connected in the direction shown in the figure are provided between the drains and sources of the switching elements Q1 and Q2.

  In this case, one end of the primary winding N1 of the insulating converter transformer PIT is connected to the connection point (switching output point) of the source and drain of the switching elements Q1 and Q2 to the primary winding N1. In contrast, a switching output is provided. The other end of the primary winding N1 is connected to the anode point of the fast recovery diode D1 of the power factor correction circuit 21 via the series resonant capacitor C1.

Also in this case, a current resonance circuit in which the switching power supply circuit is a current resonance type is formed by the capacitance of the series resonance capacitor C1 and the leakage inductance component L1 of the insulating converter transformer PIT including the primary winding N1.
A partial voltage resonance circuit is formed by the partial resonance capacitor Cp connected in parallel between the drain and source of the switching element Q2 and the leakage inductance component L1 of the primary winding N1.

In this case, the control circuit 1 outputs, for example, a control signal having a level corresponding to the fluctuation of the DC output voltage EO1 to the oscillation drive circuit 2. The oscillation drive circuit 2 varies the switching frequency by changing the frequency of the switching drive signal supplied from the oscillation drive circuit 2 to each gate of the switching elements Q1 and Q2 based on the control signal supplied from the control circuit 1. I have to. Thereby, the constant voltage control is performed as in the case of FIG.
The start circuit 3 is provided to detect the voltage or current obtained on the rectifying and smoothing line immediately after the power is turned on to start the oscillation drive circuit 2, and is additionally wound around the insulating converter transformer PIT. A low-level DC voltage obtained by rectifying N4 with a rectifier diode D30 and a smoothing capacitor C30 is used as an operating power source.

The power factor correction circuit 21 shown in this figure adopts the configuration of a power factor correction circuit as a power regeneration method using an electrostatic coupling type.
In the power factor correction circuit 21, a high frequency inductor L10 and a fast recovery type diode D1 are connected in series between the positive output terminal of the bridge rectifier circuit Di and the positive terminal of the smoothing capacitor Ci. Here, the filter capacitor CN is provided in parallel to the series connection circuit of the high frequency inductor L10 and the fast recovery type diode D1. Even in such a connection configuration, the filter capacitor CN forms a normal mode low-pass filter together with the filter high-frequency inductor L10.
The resonant capacitor C20 is provided in parallel with the fast recovery diode D1.
In addition, a current resonance circuit (N1, C1) is connected to the power factor correction circuit 21 at a connection point between the high frequency inductor L10 and the anode of the fast recovery diode D1.

Also in the power factor correction circuit 21 formed in this manner, the primary-side series resonance current is regenerated as power and fed back to the smoothing capacitor Ci through the capacitor C20, whereby the fast recovery diode D1 is The switching operation is performed when the positive and negative absolute values of the AC input voltage VAC are 1/2 or more.
As a result, the charging current to the smoothing capacitor Ci flows even during the period when the rectified output voltage level is lower than the voltage across the smoothing capacitor Ci, and the average waveform of the AC input current is the waveform of the AC input voltage. As a result, the conduction angle of the AC input current is expanded, so that the power factor is improved.

Here, the insulating converter transformer PIT provided in the power supply circuit shown in FIGS. 10 and 11 has the following structure, for example. First, as the core, an EE type core obtained by combining an E type core made of a ferrite material is provided. Then, after dividing the winding part on the primary side and the secondary side, the primary winding N1 and the secondary winding are wound around the central magnetic leg of the EE core.
In addition, a gap of about 1.0 mm is formed with respect to the central magnetic leg of the EE type core of the insulating converter transformer PIT, and the induced voltage per 1T (turn) of the secondary winding (N2). The number of turns of the secondary winding (N2) and the primary winding (N1) is set so that the level becomes 5V (5V / T).
As a result, a coupling coefficient of about 0.80 to 0.85 is obtained between the primary side winding (N1) and the secondary side winding (N2), and the required leakage inductance (L1). Is to be obtained.

  Further, as a technique for improving the power factor other than those shown in FIGS. 10 and 11, for example, as shown in FIG. 12, a configuration in which a power choke coil PCH is inserted into a commercial AC power supply line is also known. . Such power factor improvement by the power choke coil is also called a choke input method.

FIG. 13 shows the characteristics of the power supply circuit configured by inserting the power choke coil PCH into the commercial AC power supply line as shown in FIG. Each characteristic of the rate PF is shown. This figure also shows the characteristics of a power supply circuit having a configuration in which the power choke coil PCH is not inserted (power factor improvement is not performed) for comparison. In this case, the subsequent stage of the power supply circuit shown in FIG. 12 may have a circuit configuration in which the power factor correction circuits 20 and 21 are omitted from the power supply circuit of FIG. 10 or FIG.
In practice, the power choke coil PCH shown in FIG. 12 has a power factor PF of 0 at the maximum load power (in this case, Po = 125 W) as the load power Po, as in the power factor characteristic shown by the solid line in FIG. The value of the inductance Lc is set so as to be .75 or more. Thereby, the regulation value of the harmonic distortion regulation class D of the current home appliances / general-purpose electronic devices can be satisfied, for example. Note that 10 mH was selected as the actual value of the inductance Lc.

Japanese Patent Laid-Open No. 2003-189614 (FIGS. 6, 7, and 11)

However, as shown in FIGS. 10 and 11, the power supply circuit having the power factor correction circuit has the following problems.
In the power supply circuits shown in these figures, the magnetic feedback type and the electrostatic coupling type are adopted as the power feedback (regeneration) methods by the power factor correction circuits 20 and 21, respectively. The primary side series resonance current flowing through the power is regenerated and fed back to the smoothing capacitor Ci. This means that the rectification current path of the commercial AC power supply and the primary side series resonance circuit to which the switching output is supplied are connected to form a feedback path. For this reason, the current of the commercial AC current cycle is superimposed on the primary side series resonance current flowing in the primary winding N1 of the insulating converter transformer PIT.
As a result, the ripple voltage of the commercial power supply cycle of the DC output voltages E01 and E02 on the secondary side increases from before the power factor improvement. For example, in the case of a circuit configuration in which the circuit portions of the power factor improvement circuits 20 and 21 are not provided in FIGS. 10 and 11, the power factor PF = 0.55, but the power factor PF = 0.5 as the circuit of FIGS. When configured to obtain about 0.8, the ripple voltage increases 5 to 6 times.

  As a countermeasure, the capacitances of the smoothing capacitors C01 and C02 for smoothing the DC output voltage must be increased 5 to 6 times. That is, even if the gain of the control circuit 1 is increased as much as possible, in order to make it equivalent to the circuit before the power factor improvement, it is necessary to increase the capacitances of the smoothing capacitors C01 and C02 by 5 to 6 times. The cost will be increased, and practical application will be impractical. Since such a countermeasure is not practical, it is difficult to employ the circuits as shown in FIGS. 10 and 11 in a device that strictly requires, for example, the ripple voltage to be below a certain level.

Therefore, if the power factor is improved by the choke input method as shown in FIG. 12, the above-described ripple problem is solved.
However, the power choke coil PCH required in such a choke input system is problematic in that it increases in size and increases in weight, size, and cost as load power increases.
Furthermore, a place where there is no influence of the leakage magnetic flux must be selected as an arrangement position of the power choke coil PCH. Alternatively, since measures to prevent the influence of leakage magnetic flux are required, problems such as difficulty in layout design on the board and an increase in board size and weight due to the need for a shield member arise. .
In other words, as a power factor improvement technology, the power regeneration method currently has a problem of increased ripple, but the choke input method has problems such as an increase in circuit weight, size and cost due to the power choke coil PCH. The situation of becoming. From this, it can be said that there is a demand for a configuration for improving the power factor that can solve the problem of ripple increase, and that can be reduced in size and weight.

Therefore, in the present invention, in view of the above problems, the switching power supply circuit is configured as follows.
That is, first, a rectifying / smoothing means for generating a rectified and smoothed voltage corresponding to the commercial AC power supply by a rectifying circuit that performs a rectifying operation by inputting a commercial AC power supply and a smoothing capacitor that smoothes a rectified output by the rectifying circuit. Prepare.
The switching device includes a switching element that includes the switching element that performs switching by inputting the rectified and smoothed voltage as a DC input voltage, and switching driving means that performs switching driving of the switching element.
Further, at least a primary winding to which a switching output obtained by the switching operation of the switching means is supplied and a secondary winding in which an alternating voltage is excited by the switching output obtained in the primary winding are wound. An insulating converter transformer formed at least, a leakage inductance component of a primary winding of the insulating converter transformer, and a capacitance of a primary side series resonant capacitor connected in series to the primary winding, A primary-side series resonance circuit whose operation is a current resonance type.
A secondary-side DC output voltage generating means configured to generate a secondary-side DC output voltage by performing an rectifying operation by inputting an alternating voltage excited to the secondary winding of the insulating converter transformer; The constant voltage control for the secondary side DC output voltage is performed by controlling the switching drive means according to the level of the secondary side DC output voltage and varying the switching frequency of the switching means. Constant power control means, and power factor improving means for improving the power factor.
In addition, the magnetic flux density of the insulating converter transformer is set to be equal to or less than a predetermined value so that the secondary side rectified current is in a continuous mode regardless of the fluctuation of the secondary side DC output voltage.
Furthermore, as the power factor improving means, first, as a diode element forming the rectifying circuit in the rectifying and smoothing means, it is inserted in series for each rectification current path formed every half cycle of the commercial AC power supply, A first switching diode and a second switching diode configured to switch a rectified current flowing through each rectified current path are provided.
Further, the first power factor improving capacitor, one end of which is connected to the positive electrode terminal of the smoothing capacitor in the rectifying / smoothing means, and the rectified current flow commonly in each half cycle of the commercial AC power source. And a second power factor improving capacitor having one end connected to a connection point between the first switching diode and the second switching diode.
And a filter capacitor inserted in parallel with the line of the commercial AC power supply.
Then, the end of the primary side series resonance circuit is connected to a connection point between the other end of the first power factor improving capacitor and the other end of the second power factor improving capacitor. It is what is done.

The switching power supply circuit of the present invention having the above configuration includes a current resonance type converter as the primary side switching converter. In addition, the power factor is improved using a power regeneration system.
Then, as described above, the magnetic flux density of the insulating converter transformer is set to be lower than the required value, so that the fluctuation of the secondary side DC output voltage, that is, the fluctuation of the load and the level fluctuation of the commercial AC power supply (AC input voltage). Regardless of this, the secondary side rectification operation is always in a continuous mode that does not produce a period in which the secondary rectification current becomes discontinuous.
In this way, if the magnetic flux density of the insulating converter transformer is kept below the required level and the secondary side rectification operation is set to the continuous mode, it is superimposed on the primary side series resonance circuit, which was supposed to occur when the power factor is improved by the power regeneration method. The ripple voltage of the commercial AC power supply cycle superimposed on the secondary side DC output voltage, which is generated along with the ripple of the commercial AC power supply cycle, is suppressed.
In addition, according to the configuration of the power factor improving means according to the present invention, unlike the conventional electrostatic coupling type configuration, the diode element for performing the rectifying operation for the commercial AC power supply is rectified to improve the power factor. It is also used as a diode element for switching current. According to this, the number of circuit components can be reduced as compared with the conventional configuration in which a switching diode for power factor improvement is separately provided.
In the power factor improving means of the present invention, in addition to the conventional power factor improving capacitor inserted between the positive terminal of the smoothing capacitor and the end of the primary side series resonant circuit, A second power factor improving capacitor is also inserted between the connection point of the two switching diodes. According to this, the experimental result which can make the conduction angle of alternating current input into an expansion tendency rather than the case where only one conventional power factor improvement capacitor is provided was obtained. That is, a higher power factor can be obtained by adding the power factor improving capacitor in this way.

As described above, the switching power supply circuit of the present invention has a configuration for improving the power factor by the power regeneration method, and the ripple voltage of the secondary side DC output voltage is suppressed. The capacitance of the secondary side rectifying / smoothing capacitor for generating the side DC output voltage can be within a practical range. In other words, it is possible to more easily realize and push forward the practical use of a power supply circuit having a power factor improving configuration by a power regeneration method.
This also eliminates the need for the choke input method as an alternative power factor improvement technique, but this has resulted in a significant reduction in circuit board size and weight as a power circuit having a power factor improvement function. Means that.

Further, like the power factor improving means of the present invention, the diode element for performing the rectifying operation for the commercial AC power supply and the diode element for switching the rectified current for improving the power factor are combined. Accordingly, the number of parts constituting the power factor improving means can be reduced accordingly.
Further, in the present invention, the second power factor improving capacitor is provided in addition to the conventional power factor improving capacitor (first power factor improving capacitor), thereby further increasing the conduction angle of the AC input current. Can be achieved, and a higher power factor can be obtained.

Hereinafter, the best mode for carrying out the invention (hereinafter referred to as an embodiment) will be described.
FIG. 1 is a circuit diagram showing a configuration example of the switching power supply circuit according to the first embodiment of the present invention. The power supply circuit shown in this figure employs a configuration in which a partial voltage resonance circuit is combined with a current resonance type converter using a half-bridge coupling method by a separate excitation type as a basic configuration on the primary side.

  In the power supply circuit shown in FIG. 1, first, a noise filter composed of an across capacitor CL and a common mode choke coil CMC is connected to the line of the commercial AC power supply AC. In the line of the commercial AC power supply AC, a set of filter capacitors CN is connected in parallel to the subsequent stage of the noise filter. The filter capacitor CN is for suppressing normal mode noise generated in the rectified output line of the bridge rectifier circuit Di described below.

In this case, the rectifier circuit system that generates a rectified smoothed voltage (DC input voltage) Ei from the commercial AC power supply AC includes a bridge rectifier circuit Di and a smoothing capacitor Ci illustrated.
The bridge rectifier circuit Di is composed of four rectifier diodes, rectifier diodes D1 to D4. In this case, a connection point between the rectifier diode D1 and the rectifier diode D2 is a positive input terminal, a connection point between the rectifier diode D2 and the rectifier diode D4 is a positive output terminal, and a connection point between the rectifier diode D4 and the rectifier diode D3 is a negative input terminal. If the connection point between the rectifier diode D3 and the rectifier diode D1 is the negative output terminal, the positive input terminal of the bridge rectifier circuit Di is connected to the positive line of the commercial AC power supply AC. Connected. The positive output terminal is connected to the positive terminal of the smoothing capacitor Ci.
The negative input terminal is connected to the negative line of the commercial AC power supply AC and connected to the other end of the filter capacitor CN, and the negative output terminal is connected to the primary side ground.

  In this case, as an operation of the power factor correction circuit 10 described later, the rectifier diodes D1 to D4 forming the bridge rectifier circuit Di are provided in order to flow a rectification current so as to perform switching corresponding to the switching period. High-speed type (high-speed recovery type) has been selected.

  When this full-wave rectifying / smoothing circuit inputs a commercial AC power supply AC and performs a full-wave rectifying operation, a rectified and smoothed voltage Ei (DC input voltage) is obtained at both ends of the smoothing capacitor Ci. In this case, the rectified and smoothed voltage Ei is at a level corresponding to an equal magnification of the AC input voltage VAC.

  As shown in the figure, the current resonance type converter for switching (intermittently) by inputting the DC input voltage includes a switching circuit in which two switching elements Q1 and Q2 by MOS-FETs are connected by half bridge coupling. Damper diodes DD1 and DD2 are connected in parallel between the drains and sources of the switching elements Q1 and Q2. The anode and cathode of the damper diode DD1 are connected to the source and drain of the switching element Q1, respectively. Similarly, the anode and cathode of the damper diode DD2 are connected to the source and drain of the switching element Q2, respectively. The damper diodes DD1 and DD2 are body diodes provided in the switching elements Q1 and Q2, respectively.

  A partial resonance capacitor Cp is connected in parallel between the drain and source of the switching element Q2. A parallel resonance circuit (partial voltage resonance circuit) is formed by the capacitance of the partial resonance capacitor Cp and the leakage inductance L1 of the primary winding N1. A partial voltage resonance operation in which voltage resonance occurs only when the switching elements Q1, Q2 are turned off is obtained.

  In this power supply circuit, an oscillation / drive circuit 2 is provided to drive the switching elements Q1, Q2 in a switching manner. The oscillation / drive circuit 2 includes an oscillation circuit and a drive circuit, and for example, a general-purpose IC can be used. Then, a drive signal (gate voltage) having a required frequency is applied to the gates of the switching elements Q1 and Q2 by the oscillation circuit and the drive circuit in the oscillation / drive circuit 2. Thereby, the switching elements Q1 and Q2 perform the switching operation so as to be alternately turned on / off at a required switching frequency.

The insulating converter transformer PIT is provided to transmit the switching outputs of the switching elements Q1 and Q2 to the secondary side.
One end of the primary winding N1 of the insulating transformer PIT is connected to the power factor improving capacitor C20 and the power factor improving capacitor C30 in the power factor improving circuit 10 through a series connection of the primary side series resonant capacitor C1. Connected to the connection point. The other end of the primary winding N1 is connected to a connection point (switching output point) between the source of the switching element Q1 and the drain of the switching element Q2, so that the switching output is transmitted. Yes.

  Here, the insulating converter transformer PIT generates a required leakage inductance L1 in the primary winding N1 of the insulating converter transformer PIT by a structure described later. The leakage inductance L1 and at least the capacitance of the primary side series resonance capacitor C1 form a primary side series resonance circuit. According to the connection mode described above, the switching outputs of the switching elements Q1, Q2 are transmitted to the primary side series resonance circuit. The primary side series resonance circuit performs a resonance operation by the transmitted switching output, thereby making the operation of the primary side switching converter a current resonance type.

According to the above description, the primary side switching converter shown in this figure has the operation as the current resonance type by the primary side series resonance circuit (L1-C1) and the part by the partial voltage resonance circuit (Cp // L1) described above. A voltage resonance operation is obtained.
That is, the power supply circuit shown in this figure has a configuration as a complex resonance type converter in which a resonance circuit for making the primary side switching converter a resonance type is combined with another resonance circuit.

  An alternating voltage corresponding to the switching output transmitted to the primary winding N1 is excited in the secondary winding of the insulating converter transformer PIT. In this case, as the secondary winding N2, secondary windings N2A and N2B which are divided into two parts by providing a center tap are provided. In this case, the secondary windings N2A and N2B have the same predetermined number of turns.

The secondary windings N2A and N2B are provided with a synchronous rectification circuit by double-wave rectification, which includes N-channel MOS-FETs Q3, Q4, Q5, and Q6 as rectification elements as shown in the figure.
For these MOS-FETs Q3 to Q6, for example, a low breakdown voltage trench structure is selected to obtain a low on-resistance.

  The center tap output of the secondary winding N2 of the insulating converter transformer PIT is connected to the connection point of the positive terminal of the smoothing capacitor Co via a series connection of the inductor Ld as shown.

One end of the secondary winding N2 (the end on the secondary winding N2B side) is connected to a connection point between the drain of the MOS-FET Q3 and the drain of the MOS-FET Q5. The connection points of the sources of these MOS-FETs Q3 and Q5 are connected to the secondary side ground.
Similarly, the other end of the secondary winding N2 (the end on the secondary winding N2A side) is connected to the connection point between the drain of the MOS-FET Q4 and the drain of the MOS-FET Q6. The connection point of the sources of the FETs Q4 and Q6 is connected to the secondary side ground.
Note that body diodes DD3, DD4, DD5, and DD6 are connected in parallel to the drain-sources of the MOS-FETs Q3, Q4, Q5, and Q6, respectively.

  According to such a connection form, in the rectification current path including the secondary winding N2B, the parallel connection circuit of the MOS-FET Q3 // MOS-FET Q5 which is the rectification element is inserted in series. In the rectified current path including the secondary winding N2A, a parallel connection circuit of MOS-FET Q4 // MOS-FET Q6, which is also a rectifying element, is inserted in series.

Further, in the synchronous rectifier circuit shown in this figure, the drive circuit for driving the MOS-FET Q3 and the MOS-FET Q5 includes the end of the secondary winding N2A on the side not center-tapped and the gates of the MOS-FETs Q3 and Q5. The gate resistor Rg1 is connected so as to be inserted in common.
Similarly, the driving circuit for driving the MOS-FETs Q4 and Q6 is inserted so as to be common between the end portion of the secondary winding N2B which is not center-tapped and the gates of the MOS-FETs Q4 and Q6. It is formed with a gate resistance Rg2.
That is, in this case, the MOS-FETs Q3 and Q5 are turned on (conductive) / off (non-conductive) at the same timing when the alternating voltage excited by the secondary winding N2A is detected by the gate resistor Rg1. The MOS-FETs Q4 and Q6 are also turned on (conducting) / off (nonconducting) at the same timing when the alternating voltage excited by the secondary winding N2B is detected by the gate resistor Rg2. It is what has been.
Also, the pair of MOS-FETs Q3 and Q5 is turned on / off according to the voltage at the end of the secondary winding N2A that is not center-tapped, and the pair of MOS-FETs Q4 and Q6 is the center tap of the secondary winding N2B. Turning on / off in accordance with the voltage at the end that is not performed means that each pair of MOS-FETs is turned on / off in accordance with an alternating voltage having a reverse polarity. That is, the MOS-FETs Q3 and Q5 and the MOS-FETs Q4 and Q6 are rectified so that they are alternately turned on / off according to the timing at which the alternating voltage excited by the secondary winding N2 is inverted. The operation (switching operation) is performed.

Here, when an ON voltage is applied to the gate of the MOS-FET, the drain-source is equivalent to a mere resistor, so that current flows in both directions. If this is to function as a secondary side rectifying element, it is necessary to pass a current only in the direction in which the positive terminals of the secondary side smoothing capacitors (smoothing capacitors Co1 to Co4) are charged. If a current flows in the opposite direction, a discharge current flows from the secondary side smoothing capacitor to the insulating converter transformer PIT side, and power cannot be effectively transmitted to the load side. Further, the MOS-FET generates heat and noise due to the reverse current, resulting in switching loss on the primary side.
Based on the detection of the voltage of the secondary winding, the drive circuit described above allows only the current in the direction of charging the positive terminal of the secondary-side smoothing capacitor (that is, in this case, the direction of source → drain) to flow. This is a circuit for switching the MOS-FETs Q3 to Q6. In other words, the circuit configuration of the synchronous rectification circuit in this case employs a configuration in which the MOS-FET is driven on / off in synchronization with the rectification current by the winding voltage detection method.

  In this case, for the gate resistance Rg1 that forms the drive circuit system of the MOS-FETs Q3 and Q5 and the gate resistance Rg2 that forms the drive circuit system of the MOS-FETs Q4 and Q6, A Schottky diode Dg1 and a Schottky diode Dg2 are connected in parallel in the direction shown in the figure. By inserting the Schottky diodes Dg1 and Dg2 in this way, the charges stored in the gate input capacitances of the MOS-FETs Q3, Q5, Q4, and Q6 correspond to the Schottky diodes Dg1 and Schottky corresponding to these turn-off times, respectively. A path for discharging via the key diode Dg2 is formed. As a result, these MOS-FETs are reliably turned off to obtain good switching characteristics.

Further, as described above, in the power supply circuit shown in FIG. 1, the inductor Ld is inserted in series between the center tap of the secondary winding N2 and the secondary side smoothing capacitor. That is, in this case, the inductor Ld is inserted in the secondary side rectified current path with respect to the line through which the rectified current flows in common during each period in which the secondary side alternating voltage is positive / negative.
By inserting the inductor Ld in this way, it is possible to suppress noise that was supposed to be generated in the secondary side DC output voltage Eo.
When a secondary rectifier circuit is provided with a synchronous rectifier circuit using a MOS-FET, high-frequency noise is easily superimposed on the secondary-side DC output voltage Eo due to the influence of switching noise or the like caused by the MOS-FET. Thus, by inserting the inductor Ld into the rectified current path in this way, the high frequency noise component is suppressed by smoothing with the impedance component.
Further, depending on the inductor Ld inserted in the rectified current path as described above, an effect of suppressing the generation of the reverse current that is supposed to be generated in the secondary side rectified current can be obtained as described later.

Depending on the synchronous rectifier circuit having the circuit configuration described so far, an operation of charging a rectified current obtained by rectifying the secondary-side smoothing capacitor by both-wave rectification can be obtained.
That is, in one half cycle of the alternating voltage excited on the secondary side, the current flowing from the secondary winding N2B conducts through the parallel connection circuit of the MOS-FET Q3 // Q5 in the source-to-drain direction, and the smoothing capacitor Charged to Co. Further, in the other half cycle of the alternating voltage, the current flowing through the secondary winding N2A is conducted in the source-to-drain direction through the parallel connection circuit of the MOS-FET Q3 // Q5, and the smoothing capacitor Co is charged. As a result, a double-wave rectification operation in which the smoothing capacitor Co is charged in a period in which the alternating voltage is positive / negative can be obtained.
A secondary side DC output voltage Eo as shown in the figure is obtained as the voltage across the smoothing capacitor. The secondary side DC output voltage Eo is supplied to a load side (not shown) and is also branched and input as a detection voltage for the control circuit 1 described below.

In the present embodiment, the reason why the rectifier elements (MOS-FETs) of the synchronous rectifier circuit are connected in parallel as described above is as follows.
As an example here, as a load condition that the power supply circuit of the present embodiment should actually cope with, the load current varies from 30 A to 0 A with respect to the secondary side DC output voltage Eo = 5V. This is a so-called low voltage / large current load condition in which the fluctuation range is substantially wide. Therefore, if the load tends to be heavy, a considerably large current is also applied to the secondary side rectifier circuit. Flowing. Therefore, with respect to the MOS-FET that is a rectifying element, a configuration in which a plurality of MOS-FETs are connected in parallel reduces the burden on the element when a large current flows, and ensures high reliability. .

The control circuit 1 supplies a detection output corresponding to the level change of the secondary side DC output voltage Eo to the oscillation / drive circuit 2. The oscillation / drive circuit 2 drives the switching elements Q1 and Q2 such that the switching frequency is varied according to the input detection output of the control circuit 1. By changing the switching frequency of the switching elements Q1 and Q2, the resonance impedance of the primary side series resonance circuit is changed and transmitted from the primary winding N1 of the insulating converter transformer PIT to the secondary winding N2 (N2A, N2B) side. Although the amount of electric power to be changed also changes, it operates so as to stabilize the level of the secondary side DC output voltage Eo.
For example, when the secondary-side DC output voltage Eo decreases due to a heavy load tendency, the switching frequency is controlled to be lowered. However, this reduces the resonant impedance. The secondary side DC output voltage Eo is raised. On the other hand, when the secondary side DC output voltage Eo rises due to a light load tendency, the resonance impedance is increased by controlling the switching frequency to be increased, and the secondary side The DC output voltage Eo is reduced.

Next, the configuration of the power factor correction circuit 10 will be described.
The power factor improving circuit 10 includes a bridge rectifier circuit Di that performs a rectifying operation by inputting a commercial AC power source AC, a filter capacitor CN that is inserted in parallel with the line of the commercial AC power source AC, and a power factor improving circuit that is illustrated. The capacitor C20 and the power factor improving capacitor C30 are included. And as a power regeneration system, the structure of the power factor improvement circuit which belongs to an electrostatic coupling type is taken.

In the power factor correction circuit 10, as described above, the connection point between the power factor improving capacitor C20 and the power factor improving capacitor C30 is connected to the insulating converter through the series connection of the primary side series resonant capacitor C1. One end of the primary winding N1 of the transformer PIT is connected. That is, in the power factor correction circuit 10, one end of the primary side series resonance circuit (L1-C1) is connected to the connection point between the power factor correction capacitor C20 and the power factor correction capacitor C30. is there.
Thus, the primary side series resonance current corresponding to the output of the primary side switching converter is input to the power factor correction circuit 10.

As shown in the figure, the end of the power factor improving capacitor C30 opposite to the series resonant capacitor C1 is connected to the positive input terminal (the connection point between D1 and D2) of the bridge rectifier circuit Di described above. Connected to each other. One end of the power factor improving capacitor C20 opposite to the series resonant capacitor C1 is connected to a connection point between the positive terminal of the smoothing capacitor Ci and the drain of the switching element Q1.
Thus, the primary side series resonance current flows into the smoothing capacitor Ci through the power factor improving capacitor C20, and the route into the smoothing capacitor Ci after being superimposed on the rectified current through the power factor improving capacitor C30. It will branch and flow.

Here, as described above, the power factor correction circuit 10 in this case includes the bridge rectification circuit Di that performs the rectification operation on the commercial AC power supply AC. When such a bridge rectifier circuit Di performs a rectification operation, the rectified current in this case flows through a path of [rectifier diode D2 → smoothing capacitor Ci → rectifier diode D3] in one half cycle of the AC input voltage VAC.
Further, the other half cycle of the AC input voltage VAC flows through the path [rectifier diode D4 → smoothing capacitor Ci → rectifier diode D1].
According to such a rectified current path, a pair of rectifier diodes D2 and D3 forming the bridge rectifier circuit Di performs a rectification operation in one half cycle of the AC input voltage VAC. In the other half cycle of the AC input voltage VAC, the pair of rectifier diodes D1 and D4 performs a rectification operation.

Further, according to the rectified current path, it can be seen that the rectified current flows through the positive input terminal (the connection point of D1 and D2) of the bridge rectifier circuit Di in both positive and negative half cycles of the AC input voltage VAC.
According to this, as described above, as the power factor improving capacitor C30 connected to the positive input terminal, a rectified current is commonly used in each positive and negative half cycle of the AC input voltage VAC (commercial AC power supply AC). It is connected to the flowing point.
Since the positive input terminal and the power factor improving capacitor C30 are thus connected, the line on the power factor improving capacitor C30 side is in a period in which the rectified current flows by the rectifying operation of each rectifier diode. Correspondingly, the primary side series resonance current flows.

  Thus, the primary side series resonance current flowing through the power factor improving capacitor C30 is superimposed on the rectified current flowing by the rectifying operation by the rectifier diodes D1 to D4 and flows into the smoothing capacitor Ci. That is, in the rectified current path, the primary side series resonance current component superimposed on the rectified current in this way flows through the filter capacitor CN inserted in parallel with the line of the commercial AC power supply AC, thereby passing through the rectified current path. It is made to circulate and flows into the smoothing capacitor Ci.

In the power factor correction circuit 10 shown in FIG. 1, the switching output (primary side series resonance current) obtained in the primary side series resonance circuit is regenerated as power, the path through the power factor improvement capacitor C20, and the above An operation of returning to the smoothing capacitor Ci through a path via the power factor improving capacitor C30 and the rectifier diode (D2 · D3 pair or D4 · D1 pair) forming the bridge rectifier circuit Di is obtained.
In this case, since the capacitances of the power factor improving capacitor C20 and the power factor improving capacitor C30 are interposed between the smoothing capacitor Ci and the primary side series resonance circuit, it is considered that the power regeneration is performed by electrostatic coupling. be able to.

According to such a connection configuration, the power factor improving capacitor C20 and the power factor improving capacitor C30 described above are connected in series to the primary winding N1 through the series connection of the primary side series resonant capacitor C1. Will be connected.
According to this, when the primary side series resonance circuit is viewed from the switching output point on the primary side to the positive terminal of the smoothing capacitor Ci, these power factor improving capacitors C20 and C30 together with the primary side series resonance capacitor C1 are on the primary side. It can be viewed as acting as a series resonant capacitor that forms the capacitance of the series resonant circuit.

In the waveform diagram of FIG. 3, the operation of the power factor correction circuit 10 is shown by the commercial AC power supply cycle. FIG. 3 shows the operation waveforms of the main part when the AC input voltage VAC = 230 V and the load power Po = 150 W.
Here, on the assumption that the AC input voltage VAC is input as shown in the figure, power regeneration is performed from the primary side series resonance circuit to the power factor correction circuit 10 side as described above. Become. Correspondingly, the voltage V1 between the positive output terminal of the bridge rectifier circuit Di (connection point of D1 and D2) and the power factor improving capacitor C30 and the primary side ground is switched as shown in the figure. Is obtained as a waveform in which the bottom level rises as the AC input voltage VAC level increases and the peak level decreases as the AC input voltage VAC level decreases. .

The voltage V1 having such a waveform is applied to the positive input terminal of the bridge rectifier circuit Di. Correspondingly, the rectifier diodes D1 and D2, which are of the fast recovery type, perform switching operations by applying an alternating voltage component in the voltage V1, and are inserted into the rectification current path in series with D1 and D2 respectively in each half cycle. Similarly, the rectifier diodes D3 and D4, which are also high-speed types, are similarly switched so that the rectified current is interrupted.
This is also indicated by the fact that an alternating waveform with a switching period is obtained as shown in the figure as the waveform of the current I1 (rectified current) flowing through the positive input terminal of the bridge rectifier circuit Di.

According to this, the pair of rectifier diodes D2 and D3 that perform rectification operation in one half cycle of the AC input voltage VAC (commercial AC power supply) can simultaneously function as a switching diode that interrupts the rectification current during this period. Understandable (first switching diode).
The pair of rectifier diodes D4 and D1 that perform rectification operation in the other half cycle of the commercial AC power supply also functions as a switching diode that interrupts the rectification current in this half cycle (second switching diode).
In this way, the diode element that performs rectification operation for the commercial AC power supply and the switching element that interrupts the rectification current for power factor improvement are combined. In this case, a switching diode for power factor improvement is used. This eliminates the need for a separate component, thereby reducing circuit components.

When the pair of rectifier diodes D2 and D3 and the pair of rectifier diodes D1 and D4 perform a switching operation, a rectification current flows through the smoothing capacitor Ci as described above. That is, the conduction period of such rectified current corresponds to the period during which the rectifier diodes D2 and D3 and the rectifier diodes D1 and D4 are switched, and further, the conduction of the AC input current IAC shown in FIG. The period is almost the same as the conduction period of the rectified current.
According to the experimental results shown in this figure, the period during which the pair of rectifier diodes D2 and D3 and the pair of D1 and D4 perform the switching operation in this case is as shown by referring to the waveform of the current I1. There is almost no rest period. Since the rectifier diodes D2 and D3 and the rectifier diodes D1 and D4 perform the switching operation with almost no rest period as described above, the rectification current flows with almost no rest period that changes to 0 level. Accordingly, the AC input current IAC is allowed to flow almost without a rest period as shown in the figure.
In other words, in this case, the conduction period of the AC input current IAC is substantially equal to the half cycle of the AC input voltage VAC, and therefore the conduction angle of the AC input current IAC is increased to a maximum. From this, it is understood that the power factor is improved.

In the power factor improving circuit 10 shown in FIG. 1, when the power factor improving capacitor C30 is omitted and only the power factor improving capacitor C20 is provided, the conduction angle of the AC input current IAC is as described above. Did not produce results that were nearly maximized. From this, it can be said that in the circuit shown in FIG. 1, the conduction angle of the AC input current IAC is further expanded by adding the power factor improving capacitor C30.
Incidentally, FIG. 3 shows an operation waveform when the load power Po = 150 W (at the maximum load), but experimental results are obtained in which the conduction angle of the AC input current IAC tends to decrease as the load power Po decreases. It has been.

  In FIG. 3, the waveform of the current Ic3 flowing through the power factor improving capacitor C30 is shown. As shown in the figure, as this current Ic3, an alternating waveform according to the switching period is obtained, and the primary side series resonance current is obtained. It can be seen that a waveform corresponding to is obtained. In addition, the positive and negative amplitude levels of the current Ic3 increase in accordance with the increase (absolute value) of the peak level of the current I1. From such a waveform of the current Ic3, the primary side series resonance current flowing into the power factor correction circuit 10 is accompanied by the switching operation by the rectifier diodes D2 and D3 and the rectifier diodes D1 and D4, and the power factor correction capacitor. It can be understood that the flow is also diverted to the C30 side.

Here, as understood from the above description, the power factor correction circuit 10 according to the embodiment is a conventional power factor correction circuit 21 of the same electrostatic coupling type as shown in FIG. As compared with the above, a configuration in which the high-frequency inductor L10 is omitted is adopted.
In this way, the power factor correction circuit 10 in which the conventional high-frequency inductor L10 is omitted can obtain the following merits.

First, since the high-frequency inductor L10 is omitted in this way, a higher power factor can be set.
That is, in the conventional electrostatic coupling type configuration, the power factor is set by setting the inductance of the high-frequency inductor L10, and the capacitances of the power factor improving series resonance capacitor C20 and the primary side series resonance capacitor C1. In order to obtain a higher power factor, the inductance value of the high-frequency inductor L10 is set to be larger.
However, as a high value is set as the inductance of the high-frequency inductor L10, the copper loss tends to increase, and the power loss at this portion also increases. For this reason, in the configuration including the conventional high-frequency inductor L10, it is difficult to set a high power factor.
On the other hand, in the configuration of the embodiment in which the high frequency inductor L10 is omitted as described above, it is not necessary to set the inductance for setting the power factor, and the power loss as described above does not occur. The rate can be set.

According to the above description, in the embodiment, regarding the setting of the power factor, the setting of the capacitance of the power factor improving capacitor (power factor improving series resonant capacitor) C20, the power factor improving capacitor C30, and the primary side series resonant capacitor C1 is performed. Will only be done with it.
For example, depending on the setting of the respective capacitances of the power factor improving capacitor C20 and the power factor improving capacitor C30, the impedance components generated in the respective capacitors are changed, whereby the power factor improving capacitor C30 is used to improve the power factor. The level of the primary side series resonance current to be fed back can be set. That is, the power level to be fed back for power factor improvement (the peak level of the voltage V1 in FIG. 3) can be set. And since the setting about the electric power level which should be returned in this way is attained, the setting of a power factor is attained.
Thus, if the power factor can be set only by setting the capacitances of the power factor improving capacitor C20, the power factor improving capacitor C30, and the primary side series resonant capacitor C1, the inductance of the high frequency inductor L10 is also set. The circuit design can be facilitated as compared with the conventional case that must be present.

Furthermore, if the power factor can be set only by setting the capacitance of each capacitor in this way, variations in the value of the power factor can be suppressed.
Conventionally, since setting of the inductance of the high frequency inductor L10 is necessary for setting the power factor as described above, variations in the power factor value are caused by the power factor improving series resonance capacitor C20, the primary side series resonance capacitor. Such a variation in the inductance of the high-frequency inductor L10 is included in addition to the variation in the capacitance of C1.
On the other hand, in the circuit of the embodiment, such an inductance variation does not occur, and therefore the power factor variation is only the capacitance variation of each of the power factor improving capacitors C20 and C30 and the primary side series resonance capacitor C1. Will depend on.
For example, in this case, the variation in capacitance of the power factor improving capacitor C20 (C30) and the primary side series resonance capacitor C1 is about ± 3% in both the circuit of FIG. 11 and the circuit of the embodiment, and the high frequency inductor L10 in FIG. Therefore, in the case of the circuit of the embodiment, theoretically, the power factor variation can be suppressed by about ± 2%.

By the way, in the configuration of the switching converter as in the power supply circuit shown in FIGS. 10 and 11, the secondary side DC output voltage is stabilized by variable control of the switching frequency of the primary side switching element. Has been. In the case of adopting such a configuration, for example, in a state where the load tends to be light, stabilization is achieved by controlling the switching frequency to be high. In this state, the secondary-side rectifier circuit operates in a so-called continuous mode in which the period during which the secondary-side rectified current flows through the secondary-side smoothing capacitor is continuous and there is no period of pause.
On the other hand, if the primary side switching frequency is controlled to decrease as the secondary side DC output voltage drops due to heavy load, the secondary side rectification is performed with respect to the secondary side smoothing capacitor. A transition is made to a so-called discontinuous mode in which a current stops flowing and a current discontinuous period occurs. That is, there is a state in which the discontinuous mode is set according to the load variation as the secondary-side double-wave rectification operation.
The secondary side DC output voltage varies depending on the commercial AC power supply AC (AC input voltage VAC), and a constant voltage control operation corresponding to this is also performed, so that the secondary side DC output voltage also depends on the level of the AC input voltage VAC. There will be a state of discontinuous mode.

In the configuration in which the power factor is improved by the power regeneration method (magnetic coupling method, electrostatic coupling type) like the power supply circuit shown in FIG. 10 and FIG. 11, the ripple of the commercial AC power supply cycle is included in the primary side series resonance current. It has been described that, by superimposing, the ripple voltage of the commercial AC power source period superimposed on the secondary side DC output voltage is significantly increased as compared with the case where the power factor correction circuit is not provided.
As described above, this is mainly caused by the secondary side rectification operation being in the discontinuous mode due to, for example, load fluctuations and fluctuations in the AC input voltage VAC. In other words, the cause is eliminated if the continuous mode is maintained as the secondary side rectification operation regardless of the load fluctuation and the fluctuation of the AC input voltage VAC. This means that an increase in the ripple voltage of the commercial AC power supply cycle superimposed on the side DC output voltage can be effectively suppressed.

Therefore, in the present embodiment, the power supply circuit shown in FIG. 1 has the following configuration.
FIG. 2 is a cross-sectional view showing a structural example of an insulating converter transformer PIT included in the power supply circuit of FIG.
As shown in this figure, the insulating converter transformer PIT includes an EE type core (EE-shaped core) in which E-type cores CR1 and CR2 made of a ferrite material are combined so that their magnetic legs face each other.
And the bobbin B formed with the shape which divided | segmented so that it might mutually become independent about the winding part of a primary side and a secondary side, for example with a resin etc. is provided. The primary winding N1 is wound around one winding portion of the bobbin B. Further, the secondary winding N2 (N2A, N2B) is wound around the other winding portion. By attaching the bobbin B on which the primary side winding and the secondary side winding are wound in this way to the EE type cores (CR1, CR2), the primary side winding and the secondary side winding are different from each other. By the winding area, the center magnetic leg of the EE core is wound. In this way, the structure of the insulating converter transformer PIT as a whole is obtained. In this case, as an actual EE type core, for example, EER-40 is selected on the premise that it corresponds to a load condition as described later and a rated level condition of an AC input voltage.

  In addition, a gap G having a gap length of about 1.6 mm is formed on the central magnetic leg of the EE type core as shown in the figure. Thereby, as the coupling coefficient k, for example, a loosely coupled state with k = 0.75 or less is obtained. That is, as a conventional example, it is in a more loosely coupled state than the insulating converter transformer PIT of the power supply circuit shown in FIGS. Note that k = 0.75 was set as the actual coupling coefficient k. The gap G can be formed by making the central magnetic legs of the E-type cores CR1 and CR2 shorter than the two outer magnetic legs.

  Further, the induced voltage level per 1T (turn) of the secondary winding is also lower than that of the power supply circuit shown in FIGS. 10 and 11, so that the primary winding N1 and the secondary windings N2A and N2B Set the number of windings (turns). For example, by setting the primary winding N1 = 65T and the secondary winding N2A = N2B = 2T, the induced voltage level per 1T (turn) of the secondary winding is 2.5 V / T or less.

By setting the structure of the insulating converter transformer PIT and the number of turns of the primary winding N1 and the secondary winding (N2A, N2B), the magnetic flux density in the core of the insulating converter transformer PIT in this case is reduced. Thus, the leakage inductance in the insulating converter transformer PIT increases as compared with the power supply circuits shown in FIGS.
If the magnetic flux density generated in the core of the insulating converter transformer PIT can be reduced to a required level or less in this way, the electric power excited on the secondary side also changes, thereby causing a heavy load and a low AC input voltage. Even in this condition, the secondary side rectification operation can be set to the continuous mode.

  In this way, the continuous mode is obtained even under heavy load and low AC input voltage states, regardless of changes in the secondary side DC output voltage Eo due to load fluctuations, AC input voltage fluctuations, etc. This means that the secondary side rectification operation is performed in the continuous mode. As a result, in the case of adopting the configuration including the power factor correction circuit based on the power regeneration system as in the present embodiment, it is assumed that the secondary side DC output voltage Eo is superposed by the ripple superposed on the primary side series resonance current. In addition, an increase in the ripple of the commercial AC power supply cycle is greatly suppressed.

  Note that the ripple of the commercial AC power supply cycle, which was supposed to be superimposed on the secondary side DC output voltage Eo in this way, can be suppressed because the secondary side rectification operation becomes a continuous mode as described above. This is because the peak level of the secondary side rectified current is suppressed. In addition, as described with reference to FIG. 2, the magnetic flux density of the insulating converter transformer is set below the required value, so that the power transmission from the primary side to the secondary side also changes, and accordingly, the power is transferred from the primary side to the secondary side. This is also because the influence of the ripple on the commercial AC power supply period generated in the primary side series resonance current is reduced.

  As a result, the level of the ripple voltage of the commercial AC power supply cycle superimposed on the secondary side DC output voltage also decreases, and it is not necessary to increase the capacitance of the secondary side smoothing capacitor. That is, the practical use of a switching power supply circuit including a power factor correction circuit based on a power regeneration system can be easily realized. For example, in the power supply circuits shown in FIGS. 10 and 11, the ripple voltage has increased 5 to 6 times as compared with the case where the power factor correction circuit is not provided, but in the present embodiment, it is doubled. It is suppressed to the extent.

In addition, by adopting a power regenerative power factor correction circuit in a switching power supply circuit in this way, a power choke coil is inserted into a commercial AC power supply line as a means for improving the power factor. There is no need to do.
As mentioned earlier, the power choke coil is considerably large among power circuit components, and the influence of leakage magnetic flux cannot be ignored, so it is possible to omit such a power choke coil. For example, it is possible to solve problems such as an increase in the size of the power circuit board, an increase in weight, and a problem in layout design on the board.

As a specific example, when the power choke coil PCH is inserted instead of the power factor correction circuits 20 and 21 based on the configuration of the switching converter shown in FIGS. 10 and 11, for example, the weight of the power choke coil PCH Is about 153 g, the occupied volume is 32.4 cubic centimeters, and the mounting area on the printed circuit board is about 10.8 square centimeters.
On the other hand, in the present embodiment, it is not necessary to provide an inductor element as the high frequency inductor L10 as a component constituting the power factor improving circuit 10, and a filter capacitor CN, a power factor improving capacitor C20, a power factor improving capacitor. It can be constituted only by comparatively small and light component elements such as C30, switching diode D1, and switching diode D2. For this reason, the power factor improving circuit 10 as a whole is significantly smaller than the weight, occupied volume, and mounting area of the power choke coil PCH.

  In addition, as described above, in the embodiment, an inductor element is not required for the power factor correction circuit 10, so that it is not necessary to design the arrangement on the substrate in consideration of the influence of leakage magnetic flux, and the magnetic There is no need to apply a shield. This also facilitates reduction in size and weight of the power circuit board.

  Here, the power supply circuit of the present embodiment shown in FIG. 1 includes a synchronous rectifier circuit using a low-on-resistance MOS-FET as a rectifier element as a secondary side rectifier circuit. This is closely related to the fact that the secondary-side double-wave rectification operation is set to the continuous mode regardless of load fluctuations and AC input voltage VAC level fluctuations as described. This point will be described.

Here, the description starts with the power supply circuit shown in FIGS.
As described above, in the power supply circuit adopting the switching frequency control method based on the configuration shown in FIGS. 10 and 11, for example, when the load tends to be heavy or the AC input voltage VAC tends to decrease. The secondary side rectified current flowing through the secondary-side double-wave rectifier circuit becomes a discontinuous mode in which the current does not flow continuously.
Such a discontinuous mode state can be said to be a state in which the secondary side rectified current flows in a period shorter than the period in which the primary side series resonance current flows. And since the rectified current flows in a short period in this way, the peak level of the rectified current at this time becomes relatively high, and accordingly, the conduction loss of each rectifier diode on the secondary side is relatively large. turn into.
In the circuits shown in FIG. 10 and FIG. 11, a corresponding power loss occurs on the secondary side due to the conduction loss of the rectifier diode due to such a discontinuous mode.

  Therefore, as one of the techniques for reducing the secondary-side power loss due to the conduction loss of such a rectifier diode, a synchronous circuit using a low-on-resistance MOS-FET as a rectifier for the secondary-side double-wave rectifier circuit. It is known as a rectifier circuit. For example, a Schottky diode is selected as the secondary side rectifier diode of the power supply circuit shown in FIGS. 10 and 11. For example, a MOS-FET having a trench structure has a much smaller on-resistance. Therefore, by making the secondary side rectifier circuit a synchronous rectifier circuit, it is possible to reduce conduction loss in the rectifier element and reduce power loss on the secondary side.

  As such a synchronous rectifier circuit, for example, a resistance element for detecting an alternating voltage obtained at the secondary winding N2 (secondary winding N2A, N2B) of the insulating converter transformer PIT is provided, and the detected voltage serves as a rectifier element. The MOS-FET is turned on / off. This is also called a winding voltage detection method.

However, in the discontinuous mode state, even after the charging current to the smoothing capacitor becomes 0 level, the primary side series resonance current having the same polarity flows in the primary winding N1 during the discontinuous period. The polarity of the induced voltage of the winding N2 is not reversed, and during this period, the MOS-FET is not completely turned off but remains on.
Since the MOS-FET is turned on even after the charging current to the smoothing capacitor becomes 0 level in this way, a reverse current flows as a rectified current during this period, and the invalidity due to the reverse current is lost. Electricity is generated.
As a result, in the case of a synchronous rectifier circuit that employs the winding voltage detection method, although conduction loss in the rectifier element is reduced, the generation of reactive power due to such reverse current effectively increases the overall power conversion efficiency. It is difficult to plan.

  Therefore, as a technique for solving the problem of generation of reactive power due to the reverse rectified current as described above, a synchronous rectifier circuit based on a rectified current detection method is known. This rectified current detection method is a technique for turning off the MOS-FET before the rectified current charged in the secondary-side smoothing capacitor Co becomes 0 level.

As a circuit configuration for this purpose, for example, a current (rectified current) flowing through the secondary winding N2 is detected by a current transformer or the like. The current detected by the current transformer is output as a voltage (detection voltage), and this detection voltage is compared with a predetermined reference voltage by a comparator.
Here, when the rectified current starts flowing so as to charge the smoothing capacitor Co, the rectified current is detected by the current transformer, and a detection voltage corresponding to the rectified current level is input to the comparator. The comparator compares the reference voltage with the detection voltage, and outputs an H level when the detection voltage exceeds the reference voltage, for example. This H level output is applied as an on-voltage from the buffer to the gate of the MOS-FET which is a rectifying element to turn on the MOS-FET. As a result, the rectified current flows in the source-to-drain direction of the MOS-FET.
When the level of the rectified current decreases with the passage of time and the detection voltage, which is the output of the current transformer, becomes lower than the reference voltage accordingly, the comparator inverts the output. By outputting this inverted output through the buffer, the gate capacitance of the MOS-FET which is a rectifying element is discharged, and the MOS-FET is turned off.

  With such an operation, the MOS-FET which is a rectifying element is turned off at a timing before the rectified current becomes 0 level. As a result, like the synchronous rectification circuit using the winding voltage detection method, the reverse current does not flow through the MOS-FET in the period in which the rectification current is discontinuous, and no reactive power is generated. Conversion efficiency is increased.

However, as can be seen from the above description, in such a synchronous rectification circuit of the rectification current detection system, at least one set of current transformers and one output of the current transformer correspond to one MOS-FET, and the MOS-FET Requires a relatively complicated drive circuit system for driving the drive. As a result, the circuit configuration becomes complicated, resulting in inconveniences such as reduced manufacturing efficiency, increased cost, and increased circuit board size.
In particular, when the configuration of a current resonance type switching converter is based on the power supply circuit shown in FIGS. 10 and 11, for example, the synchronous rectification circuit needs to be a double-wave rectification circuit. Therefore, two sets of the above-described current transformer and driving circuit system are required corresponding to each half-wave period, and the above-described problem is further increased.

Considering the adoption of a synchronous rectifier circuit on the premise that there is a state of discontinuous mode as the secondary side rectification operation in this way, the merits of the winding voltage detection method and the rectification current detection method are Must be in a trade-off relationship. That is, the winding voltage detection method is disadvantageous in terms of power conversion efficiency, but the circuit configuration is simplified. In contrast, the rectified current detection method is advantageous in terms of power conversion efficiency because reactive power is not generated, but the circuit configuration is complicated.
In other words, as long as there is a condition for the discontinuous mode as the secondary side rectification operation, for example, even if an attempt is made to adopt a synchronous rectification circuit in consideration of power conversion efficiency, a rectification current detection method must be adopted. As a result, the circuit configuration becomes complicated.

On the other hand, in the present embodiment, the configuration of the insulating converter transformer PIT described above allows the secondary-side double-wave rectification operation to be performed in the continuous mode regardless of load fluctuations and AC input voltage fluctuations. It is said.
As described above, the synchronous rectifier circuit provided in the power supply circuit shown in FIG. 1 employs the winding voltage detection method. However, since the continuous mode is always obtained as described above, There is no reactive power. In other words, the present embodiment includes a winding voltage detection type synchronous rectifier circuit, which suppresses an increase in circuit scale as a simple circuit configuration and further avoids an increase in cost, while also reducing current consumption. This effectively solves the problem of reduction in power conversion efficiency caused by reactive power in continuous periods.

In summary, the power supply circuit according to the present embodiment firstly sets the insulating converter transformer PIT in a loosely coupled state with respect to the problem of an increase in ripple when a power factor improvement circuit using a power regeneration system is provided. The problem is solved by setting the secondary-side rectification operation to the discontinuous mode.
Furthermore, the secondary-side double-wave rectification operation is set to the discontinuous mode, so that reactive power during the current discontinuity period does not occur in the secondary-side rectifier circuit. A synchronous rectifier circuit is provided. This effectively reduces power loss in the secondary side rectifier circuit by adding a small number of components.

  In the present embodiment, the inductor Ld is inserted into the secondary side rectified current path. However, when the inductance value is set appropriately, the impedance component of the inductor Ld causes the rectified current. An operation of suppressing the reverse current component is obtained. That is, the insertion of the inductor Ld further enhances the suppression of the reverse current during the current discontinuous period.

For reference, the power factor PF and the change characteristics of the rectified smoothing voltage Ei (DC input voltage) are shown in FIGS. 4 and 5 as a result obtained by actually conducting experiments on the power supply circuit shown in FIG. I will show you.
In these figures, FIG. 4 shows each characteristic with respect to fluctuations in the load power Po = 150 W to 0 W when the AC input voltage VAC = 230 V is constant, and FIG. 5 shows the load power Po = 150 W constant. In this case, each characteristic with respect to fluctuation of the AC input voltage VAC = 170V to 264V is shown.
In order to obtain the experimental results shown in these figures, components were selected for the main part of the power supply circuit shown in FIG. 1 as follows.

Insulating converter transformer PIT: EER-40 ferrite core, gap G = 1.6mm, primary winding N1 = 65T, secondary windings N2A, N2B = 2T, coupling coefficient k = 0.75
Primary side series resonant capacitor C1 = 0.033μF
Power factor improving capacitor C20 = 0.01 μF
Power factor improving capacitor C30 = 0.1 μF
Filter capacitor CN = 1μF

First, for the power factor PF, the characteristics shown in FIGS. 4 and 5 are obtained. In particular, in FIG. 4, the power factor PF with respect to the load fluctuation has a characteristic that its value increases as the load tends to be heavy. When the maximum load power Po = 150 W, the power factor PF is about 0.9. Has been improved.
For example, in the circuits shown in FIGS. 10 and 11, the power factor PF is about 0.8, but the high power factor can be obtained in the embodiment in this way. As described above, the conventional high frequency inductor L10 is omitted, and the power factor improving capacitor C30 is added.
According to the explanation of FIG. 3, the conduction angle of the AC input current IAC when the AC input voltage VAC = 230 V and the load power Po = 150 W is substantially equal to the conduction angle of the AC input voltage VAC. It was something. However, as shown in FIG. 3, the waveform of the current Ic3 flowing in the power factor improving capacitor C30 in this case has a peak level that rises relatively steeply in the first half of the half cycle of the AC input voltage VAC. In the second half period, a waveform in which the peak level gradually decreases is obtained. Similarly, the current I1 flowing through the positive input terminal of the bridge rectifier circuit Di also has a waveform in which the peak level rises sharply in the first half period and gradually decreases in the second half period. As the current I1 has such a waveform, the waveform of the AC input current IAC is similar to the waveform of the current I1 and does not have a complete sine wave shape. Therefore, in this case, the conduction angle is almost maximized, but the value of the power factor PF is only about 0.9.

  Further, according to the characteristics shown in FIG. 5, the power factor PF has a result that its value improves as the AC input voltage VAC increases. For example, depending on the configuration of a power factor correction circuit having only one conventional power factor correction capacitor (power factor correction series resonance capacitor), the power factor decreases as the AC input voltage VAC increases. However, since the power factor improving capacitor C30 is added as in the embodiment, the opposite characteristics are obtained.

Also, the DC input voltage Ei tends to increase as the AC input voltage VAC increases as shown in FIG. On the other hand, as shown in FIG. 4, in this case, a flat characteristic is obtained with respect to the load fluctuation.
In this way, the level of the DC input voltage Ei changes with respect to the load fluctuation because the high-frequency inductor L10 is omitted, and only one series resonant circuit is formed on the primary side of the power supply circuit. According to what was said. That is, since the number of series resonance circuits is one and the resonance point of the primary side series resonance circuit is also one, the fluctuation of the DC input voltage Ei with respect to the load fluctuation, that is, the fluctuation of the primary side series resonance current is suppressed. It will be.

  Next, a configuration example of the switching power supply circuit according to the second embodiment will be described with reference to the circuit diagram of FIG. In the power supply circuit shown in FIG. 6, the power factor correction circuit 10 provided in the power supply circuit of the first embodiment shown in FIG. The configuration is applied in the case of a rectifier circuit. In this figure, the same parts as those in FIG.

In the power factor correction circuit 11 shown in this figure, a rectifier diode D1 (first switching diode), a rectifier diode D2 (second switching diode), and smoothing capacitors Ci1 and Ci2 are formed to form a voltage doubler rectifier circuit. (First smoothing capacitor, second smoothing capacitor).
For power factor improvement, a power factor improving capacitor C20, a power factor improving capacitor C30, and a filter capacitor CN are provided.
In this case, the rectifier diodes D1 and D2 have a function as a rectifier diode for obtaining a rectified output in the voltage doubler rectifier circuit and a function as a power factor improving switching element for switching the rectified current. In this respect, it is common to the circuit shown in FIG. 1, and this also has a common merit in that it is not necessary to separately configure a power factor improving switching diode and the number of parts can be reduced.
In this case, the rectifier diodes D1 and D2 also switch the rectified current in the switching cycle of the switching elements Q1 and Q2, so that a high-speed diode element is selected.

In FIG. 6, smoothing capacitors Ci1 and Ci2 are connected in series, and these connection points are connected to one end of a filter capacitor CN inserted in parallel to the line of the commercial AC power supply AC. It is connected to the negative electrode line of the AC power supply AC. The negative terminal side of the smoothing capacitor Ci2 is connected to the primary side ground.
The cathode of the rectifier diode D1 is connected to the positive terminal of the smoothing capacitor Ci1. Further, the anode of the rectifier diode D1 is connected to the cathode of the rectifier diode D2, and the anode of the rectifier diode D2 is connected to the primary side ground.

  Also in the power factor correction circuit 11, the primary side series resonance capacitor C1 is connected to the connection point between the power factor correction capacitor C20 and the power factor correction capacitor C30. In this case, the other end of the power factor improving capacitor C30 is connected to the connection point between the anode of the rectifying diode D1 and the cathode of the rectifying diode D2 on the power factor improving circuit 11 side. The other end of the power factor improving capacitor C20 is connected to a connection point between the positive terminal of the smoothing capacitor Ci1 and the drain of the switching element Q1.

  In this case, one end of the high-frequency inductor L10 is further connected to the connection point between the power factor improving capacitor C30 and the anode of the rectifier diode D1 and the cathode of the rectifier diode D2. The other end of the high-frequency inductor L10 is connected in series to the positive line of the commercial AC power supply AC by being connected to the other end of the filter capacitor CN described above.

The voltage doubler rectifier circuit formed in the power factor correction circuit 11 according to the above-described connection form has a high frequency inductor L10 → rectifier diode D1 during a period in which the commercial AC power supply AC (AC input voltage VAC) is one half cycle. → Rectified current flows through the path of the smoothing capacitor Ci1. In other words, the rectifier diode D1 rectifies the commercial AC power supply AC, and the smoothing capacitor Ci1 smoothes the rectified output, so that the rectified smoothing voltage corresponding to the commercial AC power supply AC is the same as the voltage across the smoothing capacitor Ci1. Generate.
Similarly, during a period in which the commercial AC power supply AC (AC input voltage VAC) is in the other half cycle, a rectified current flows through a path of the smoothing capacitor Ci 2 → the rectifier diode D 2 → the high frequency inductor L 10. That is, the rectifier diode D2 rectifies the commercial AC power supply AC, and the smoothing capacitor Ci2 smoothes the rectified output, so that the rectified smoothing voltage corresponding to the commercial AC power supply AC is the same as the voltage across the smoothing capacitor Ci2. Generate.
As a result, the rectified and smoothed voltage Ei having a level corresponding to twice the level of the commercial AC power supply AC is obtained as the voltage across the series connection circuit of the smoothing capacitors Ci1 to Ci2. The rectified and smoothed voltage Ei is supplied as a DC input voltage to the subsequent switching converter.

  Here, according to the above description, the power factor improving capacitor C30 in this case is connected to the connection point between the rectifier diodes D1 and D2, but according to the rectified current path, the power point improvement capacitor C30 is connected to this connection point. The rectified current flows in common every half cycle of the commercial AC power supply AC (AC input voltage VAC).

In the power factor correction circuit 11 having the above configuration, the primary side series resonance current from the primary side series resonance circuit flows to the smoothing capacitor Ci (Ci1, Ci2) via the power factor improvement capacitor C20 side, The feedback is made by the path flowing through the smoothing capacitor Ci through the power factor improving capacitor C30 side.
That is, the power factor improvement circuit 11 shown in FIG. 6 also achieves the power factor improvement operation by the magnetic coupling type power regeneration system.

  In this case, as described above, the high-frequency inductor L10 is inserted in series between the connection point of the rectifier diodes D1 and D2 and the end of the filter capacitor CN. This is shown in FIG. Unlike the case, if such a high-frequency inductor L10 is not inserted, in the case of the circuit of FIG. 6, a path through which the fed back primary side series resonance current flows to the smoothing capacitor Ci through the filter capacitor CN is formed. This is because the switching operation of the rectifier diodes D1 and D2 is stopped.

A waveform diagram corresponding to the power factor correction operation of the power factor correction circuit 11 according to the above description is shown in FIG.
The power supply circuit shown in FIG. 6 is different from the power supply circuit shown in FIG. 1 in that the AC input voltage VAC = 85V to 144V and the load power Po = 0W to 150W. In FIG. Operation at the time of AC input voltage VAC = 100V and load power Po = 150W is shown.
In FIG. 7, the switching elements Q1 and Q2 are also switched under the state where the AC input voltage VAC having the waveform shown here is input. Accordingly, an operation of regenerating and returning the switching output from the primary side series resonance circuit to the power factor correction circuit 11 is obtained.
Accordingly, the voltage V1 between the connection point of the rectifier diodes D1 and D2 and the power factor improving capacitor C30 and the primary side ground is also changed to an alternating waveform with a switching period as shown in FIG. The waveform is obtained in such a manner that the bottom level rises as the input voltage VAC level increases, and the peak level decreases as the VAC level decreases.

Here, in this case, as can be seen with reference to the waveform of the current Ic3 flowing through the power factor improving capacitor C30, it is understood that the primary side series resonance current component always flows in the path on the power factor improving capacitor C30 side. . Therefore, as described above, when the high-frequency inductor L10 is not inserted, the primary side series resonance current component flows into the smoothing capacitor Ci through the filter capacitor CN, and the rectifier diode D1 · It can be understood that the potential for switching the D2 cannot be obtained.
In this case, the voltage corresponding to the primary side series resonance current is always applied to the rectifier diodes D1 and D2 by the insertion of the high frequency inductor L10. Therefore, the rectifier diodes D1 and D2 are stationary in the half cycle of the AC input voltage VAC. The switching operation is performed (see the waveforms of the voltage V1 and the current I1).
In this way, the smoothing capacitors D1 and D2 are constantly switching in each half cycle, and in accordance with the increase of the positive / negative level in each half cycle of the AC input voltage VAC, the smoothing capacitor The amount of rectified current flowing through Ci is increased. This is indicated by the fact that the bottom level of the current I1 increases and the peak level decreases as the positive / negative level of the AC input voltage VAC increases.

As described above, the rectifier diodes D1 and D2 are constantly switched in the half cycle of the AC input voltage VAC, so that the conduction angle of the AC input current IAC is expanded to the maximum. Will be. As a result, it can be seen that the power factor is also improved in the circuit of FIG.
As shown in the figure, as the AC input current IAC in this case, the waveform in the period near the peak level is made to be substantially square. During this period, the bottom level of the current I1 as described above is set. This corresponds to a period during which the peak level rises and a period when the peak level falls. In other words, the waveform during this period is the bottom level of the current I1 in accordance with the increase in the positive / negative level of the AC input voltage VAC as described above, while the rectifier diodes D1 and D2 are constantly switching. Is obtained in response to an increase in peak level and a decrease in peak level.

8 and 9 show the change characteristics of the power factor PF and the rectified and smoothed voltage Ei (DC input voltage) for the power supply circuit shown in FIG.
FIG. 8 shows each characteristic with respect to fluctuations in load power Po = 150 W to 0 W when AC input voltage VAC = 100 V is constant, and FIG. 9 shows AC input voltage when load power Po = 150 W is constant. Each characteristic with respect to the fluctuation of VAC = 85V to 144V is shown.
In order to obtain the experimental results shown in these figures, components were selected as follows for the main part of the power supply circuit shown in FIG.

Insulating converter transformer PIT: EER-40 ferrite core, gap G = 1.6mm, primary winding N1 = 65T, secondary windings N2A, N2B = 2T, coupling coefficient k = 0.75
Primary side series resonant capacitor C1 = 0.033μF
Power factor improving capacitor C20 = 0.01 μF
Power factor improving capacitor C30 = 0.1 μF
Filter capacitor CN = 1μF
High frequency inductor L10 = 130μH

First, as shown in FIG. 8, the power factor PF with respect to the load fluctuation has a characteristic that its value increases as the load tends to be heavy.
According to the experiment, the power factor PF in this case has a power factor PF = 0.95 or more in the range of the load power Po = 150 W to 50 W when the AC input voltage VAC = 100V. In addition, when the AC input voltage VAC = 100 V, the result that the power factor PF = 0.75 or more was obtained for the load power Po = 150 W to 20 W.

Further, with respect to the fluctuation of the AC input voltage VAC, the result that the value of the power factor PF is improved with the increase is obtained. According to experiments, when the load power Po = 150 W and the AC input voltage VAC = 144 V, a high power factor value of power factor PF = 0.98 was obtained.
In this way, the high power factor is obtained in the circuit of FIG. 6 because the power factor improving capacitor C30 is added as in the case of the circuit of FIG. 1, and in this case, a high frequency inductor L10 is provided to provide the rectifier diode D1. , D2 is considered to be caused by the switching operation being steadily performed in each half cycle of the AC input voltage VAC.

The setting of the power factor in the circuit shown in FIG. 6 includes the setting of the capacitance of the primary side series resonance capacitor C1, the power factor improving capacitor C20, and the power factor improving capacitor C30, and the inductance of the high frequency inductor L10. Setting is required.
In the second embodiment, the combined capacitance (capacitance) of the primary side series resonant capacitor C1, the power factor improving capacitor C20, and the power factor improving capacitor C30 is set to C20 <C30, and the power factor is improved. The capacitance is set to be equal to the capacitance of the primary side series resonance capacitor C1 that should be set when the circuit 11 is not provided, and the inductance of the high frequency inductor L10 is set to about 120 μH to 130 μH.
In other words, in practice, as described above, the primary side series resonant capacitor C1 = 0.033 μF, the power factor improving capacitor C20 = 0.01 μF, and the power factor improving capacitor C30 = 0.1 μF. The combined capacitance is
{(C20 + C30) C1} / (C20 + C30) + C1
= {(0.01 + 0.1) 0.033} / (0.01 + 0.1) +0.033
Thus, by setting to about 0.025 μF and setting the high-frequency inductor L10 = 130 μH, the high power factor as described above can be set.

  Further, as shown in FIG. 9, the DC input voltage Ei tends to increase as the AC input voltage VAC increases. As shown in FIG. 8, the load fluctuation is not constant in this case, but the fluctuation width is relatively small.

Also in the power supply circuit of the second embodiment shown in FIG. 6, the insulating converter transformer PIT is similar to the configuration described in FIG. 2, and is superimposed on the secondary side DC output voltage Eo. Solves the ripple problem.
As in the case of the first embodiment, the power factor correction capacitor C30 is added to the power factor correction circuit, so that a high power factor can be set.
As the high frequency inductor L10 provided for the power factor correction circuit 11, a very slight inductance is set as exemplified above, and therefore the power choke coil PCH described above with reference to FIG. It will not be such a large one. Therefore, also in the second embodiment, the area weight of the power factor correction circuit 11 can be small and light.

Here, the present invention is not limited to the configuration of the power supply circuit described so far.
First, regarding the insulating converter transformer PIT, the structure may be changed as appropriate as long as the magnetic flux density is less than the required one, including the core type, for example.
For example, although the switching converter as an embodiment is based on a separately excited current resonance type converter, it may be configured to include a self-excited current resonance type converter. In this case, for example, a bipolar transistor can be selected as the switching element. Furthermore, the present invention can also be applied to a current resonance type converter in which four stone switching elements are full-bridge coupled. For example, as the primary side switching elements (Q1, Q2) of the switching converter, an element other than a MOS-FET may be employed as long as it is an element that can be used by another excitation type, such as an IGBT (Insulated Gate Bipolar Transistor). Absent. Also, the constants of the component elements described above may be changed according to actual conditions.
Further, the detailed configuration of the winding voltage detection type synchronous rectification circuit, which is a secondary-side double-wave rectification circuit, may be changed as appropriate.

1 is a circuit diagram illustrating a configuration example of a switching power supply circuit according to a first embodiment of the present invention. It is a figure which shows the structural example of the insulation converter transformer with which the power supply circuit of embodiment is equipped. It is a wave form diagram which shows operation | movement of the principal part corresponding to the power factor improvement operation | movement of the power supply circuit of 1st Embodiment. It is a figure which shows the power factor with respect to load fluctuation | variation in the power supply circuit of 1st Embodiment, and the characteristic of a rectification | straightening smoothing voltage (DC input voltage). It is a figure which shows the power factor with respect to the fluctuation | variation of alternating current input voltage in the power supply circuit of 1st Embodiment, and the characteristic of a rectification | straightening smoothing voltage (DC input voltage). It is a circuit diagram which shows the structural example of the switching power supply circuit as the 2nd Embodiment of this invention. It is a wave form diagram which shows operation | movement of the principal part corresponding to the power factor improvement operation | movement of the power supply circuit of 2nd Embodiment. It is a figure which shows the power factor with respect to load fluctuation | variation in the power supply circuit of 2nd Embodiment, and the characteristic of a rectification | straightening smoothing voltage (DC input voltage). It is a figure which shows the characteristic of the power factor with respect to the fluctuation | variation of alternating current input voltage in the power supply circuit of 2nd Embodiment, and the rectification | straightening smoothing voltage (DC input voltage). It is a circuit diagram which shows the structural example of the conventional switching power supply circuit. It is a circuit diagram which shows the other structural example of the conventional switching power supply circuit. It is a circuit diagram which shows the structural example by a choke input system as a conventional power factor improvement technique. It is a figure which shows the characteristic of the power factor with respect to load fluctuation | variation, and a rectification | straightening smoothing voltage (DC input voltage) in the switching power supply circuit which employ | adopts a choke input system.

Explanation of symbols

  1 control circuit, 2 oscillation / drive circuit, 10 power factor correction circuit, Di bridge rectifier circuit, Ci smoothing capacitor, Q1, Q2 switching element, PIT isolation converter transformer, C1 primary side series resonance capacitor, Cp partial resonance capacitor, N1 primary Winding, N2 (N2A, N2B) Secondary winding, CN filter capacitor, D1, D2, D3, D4 rectifier diode, C20, C30 power factor improving capacitor, L10 high frequency inductor, Q3-Q6 MOS-FET, Rg1, Rg2 gate resistance, Dg1, Dg2 Schottky diode, Ld inductor, Co (secondary side) smoothing capacitor

Claims (4)

Rectifying and smoothing means for generating a rectified and smoothed voltage according to the commercial AC power supply by a rectifying circuit that inputs a commercial AC power supply and performs a rectifying operation, and a smoothing capacitor that smoothes a rectified output by the rectifying circuit;
Switching means formed by including a switching element that performs switching by inputting the rectified and smoothed voltage as a DC input voltage;
Switching driving means for switching and driving the switching element;
At least a primary winding to which a switching output obtained by the switching operation of the switching means is supplied and a secondary winding in which an alternating voltage is excited by the switching output obtained by the primary winding are wound. An isolated converter transformer,
The primary side which is formed by at least the leakage inductance component of the primary winding of the insulating converter transformer and the capacitance of the primary side series resonant capacitor connected in series to the primary winding, and the operation of the switching means is a current resonance type A series resonant circuit;
A secondary side DC output voltage generating means configured to input an alternating voltage excited to the secondary winding of the insulating converter transformer and perform a rectification operation to generate a secondary side DC output voltage;
The switching drive means is controlled according to the level of the secondary side DC output voltage, and the switching frequency of the switching means is varied to perform constant voltage control on the secondary side DC output voltage. Constant voltage control means;
Power factor improving means for improving the power factor, and
The magnetic flux density of the insulating converter transformer is set to be equal to or less than a predetermined value so that the secondary side rectified current is in a continuous mode regardless of the fluctuation of the secondary side DC output voltage, and
The power factor improving means is
As a diode element forming the rectifying circuit in the rectifying and smoothing means, it is inserted in series for each rectifying current path formed every half cycle of the commercial AC power supply, and switches the rectifying current flowing through each rectifying current path. A first switching diode and a second switching diode configured as described above,
A first power factor improving capacitor having one end connected to the positive terminal of the smoothing capacitor in the rectifying and smoothing means;
A second power factor having one end connected to a connection point between the first switching diode and the second switching diode, in which a rectified current flows in common in each half cycle of the commercial AC power supply. An improvement capacitor,
A filter capacitor inserted in parallel with the line of the commercial AC power supply,
An end of the primary side series resonance circuit is connected to a connection point between the other end of the first power factor improving capacitor and the other end of the second power factor improving capacitor;
A switching power supply circuit. The rectifying and smoothing means is
A bridge rectifier circuit is provided as the rectifier circuit, and is an equal voltage rectifier circuit configured to obtain the rectified and smoothed voltage at a level corresponding to the same level as the level of the commercial AC power supply at both ends of the smoothing capacitor.
The power factor improving means is
Of the four diode elements constituting the bridge rectifier circuit, two diode elements inserted in series in a rectified current path formed in one half cycle of the commercial AC power supply are provided as the first switching diode. The second switching diode includes two diode elements inserted in series in a rectified current path formed in the other half cycle of the commercial AC power source,
One end of the second power factor improving capacitor is connected to the positive input terminal of the bridge rectifier circuit.
The switching power supply circuit according to claim 1. The rectifying and smoothing means is
As the rectifier circuit, a first diode element that rectifies the commercial AC power supply corresponding to one half cycle of the commercial AC power supply, and a first diode element that rectifies the commercial AC power supply corresponding to the other half cycle of the commercial AC power supply. Two diode elements, and
A first smoothing capacitor for smoothing the rectified output of the first diode element to generate a DC voltage; and a second smoothing capacitor for smoothing the rectified output of the second diode element to generate a DC voltage. A voltage doubler rectifier circuit configured to generate a voltage across the series connection of the first smoothing capacitor and the second smoothing capacitor as the rectified smoothing voltage,
The power factor improving means is
The first diode element is provided as the first switching diode, the second diode element is provided as the second switching diode, and
One end of the second power factor improving capacitor is connected to a connection point between the anode of the first switching diode and the cathode of the second switching diode;
Further, the rectifying current path of the rectifying / smoothing means formed corresponding to one half cycle of the commercial AC power supply, and the rectifying of the rectifying / smoothing means formed corresponding to the other half cycle of the commercial AC power supply. A high-frequency inductor is inserted into a common line in the current path.
The switching power supply circuit according to claim 1. As a rectifier circuit forming the secondary side DC output voltage generating means, a synchronous rectifier circuit by a winding voltage detection method is provided,
The switching power supply circuit according to claim 1.

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