JP2005304116A - Switching power supply - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide a switching power supply in which increase of ripple is suppressed even if an arrangement for improving power factor is included. <P>SOLUTION: In a complex resonance converter combining a half-bridge coupling current resonance converter and a partial resonance voltage circuit, a power factor improving circuit 10 of such an arrangement (power regeneration system) as a primary series resonance current obtained in a primary series resonance circuit is regenerated as power and fed back to a smoothing capacitor Ci improves power factor. Under such an arrangement, flux density of an insulating converter transformer PIT is set at a predetermined level or below so that the rectification operation current of a secondary both-wave rectification circuit has a continuous mode regardless of load variation thus suppressing increase in ripple of the secondary DC output voltage caused by power regeneration. Furthermore, lowering or variation of power factor is suppressed by providing a switching diode D2 in place of a high frequency inductor provided in the power factor improving circuit. <P>COPYRIGHT: (C)2006,JPO&NCIPI

Description

  The present invention relates to a switching power supply circuit provided as a power supply for various electronic devices.

The present applicant has previously proposed various power supply circuits having a resonance type converter on the primary side. Various power supply circuits configured with a power factor correction circuit for improving the power factor of the resonant converter have also been proposed.
6 and 7 are circuit diagrams each showing an example of a switching power supply circuit configured based on the invention previously filed by the present applicant.

  First, the power supply circuit of FIG. 6 is configured to include a power factor correction circuit 20 for a self-excited current resonance type switching converter. The switching converter in this power supply circuit is a switching converter that combines a current resonance type converter using a half-bridge coupling method and a partial voltage resonance circuit that performs voltage resonance only when the semiconductor switch (switching element) is turned off.

The power supply circuit shown in FIG. 6 includes a bridge rectifier circuit Di that performs full-wave rectification on the commercial AC power supply AC. In this case, the rectified output rectified by the bridge rectifier circuit Di is charged to the smoothing capacitor Ci via the power factor correction circuit 20, and both ends of the smoothing capacitor Ci correspond to a level equal to the AC input voltage VAC. The rectified and smoothed voltage Ei is obtained.
The power factor correction circuit 20 will be described later.

The power supply circuit includes a self-excited current resonance converter that uses a rectified and smoothed voltage Ei that is a voltage across the smoothing capacitor Ci as an operation power supply.
In this current resonance type converter, as shown in the figure, the switching elements Q1 and Q2 by two bipolar transistors are half-bridge coupled and then connected so as to be inserted between the positive side of the smoothing capacitor Ci and the primary side ground. Has been.
Starting resistors RS1 and RS2 are inserted between the collectors and bases of the switching elements Q1 and Q2, respectively. The resistors RB1 and RB2 connected to the bases of the switching elements Q1 and Q2 set the base current (drive current) of the switching elements Q1 and Q2.
Clamp diodes DD1 and DD2 are inserted between the bases and emitters of the switching elements Q1 and Q2, respectively. The clamp diodes DD1 and DD2 form a current path of a clamp current that flows through the base-emitter during the period when the switching elements Q1 and Q2 are turned off, respectively.
The resonance capacitors CB1 and CB2 together with drive windings NB1 and NB2 of the drive transformer PRT described below form a series resonance circuit (self-oscillation drive circuit) for self-oscillation, and switching elements Q1,. Determine the switching frequency of Q2.

The drive transformer PRT (Power Regulating Transformer) in this case is provided to drive the switching elements Q1 and Q2 and to perform constant voltage control by variably controlling the switching frequency. The drive transformer PRT in this figure is an orthogonal type saturable reactor in which the drive windings NB1 and NB2 are wound and the control winding NC is wound in a direction perpendicular to the respective windings. ing.
One end of the drive winding NB1 of the drive transformer PRT is connected to the base of the switching element Q1 via a series connection of a resistor RB1 and a resonance capacitor CB1. The other end side of the drive winding NB1 is a tap point that is a connection point with the end of the resonance current detection winding ND. The other end (tap point) of the drive winding NB1 is the switching element Q1. Connected to the emitter.
One end of the drive winding NB2 is grounded to the ground, and the other end is connected to the base of the switching element Q2 via a series connection of a resistor RB2 and a resonance capacitor CB2.
The drive winding NB1 and the drive winding NB2 are wound so that voltages having opposite polarities are generated.

An insulating converter transformer PIT (Power Isolation Transformer) transmits the switching outputs of the switching elements Q1 and Q2 to the secondary side.
One end of the primary winding N1 of the insulating converter transformer PIT is connected to the contact (switching output point) of the emitter of the switching element Q1 and the collector of the switching element Q2 via the resonance current detection winding ND, so that the switching output Will be obtained.
The other end of the primary winding N1 is connected to the cathode of the fast recovery diode D1 in the power factor correction circuit 20 through the series resonance capacitor C1.

  In this case, the series resonant capacitor C1 and the primary winding N1 are connected in series, but the capacitance of the series resonant capacitor C1 and the leakage inductance of the insulating converter transformer PIT including the primary winding N1 (series resonant winding) ( A primary side current resonance circuit for making the operation of the switching converter into a current resonance type is formed by the leakage inductance component L1.

A partial resonance capacitor Cp is connected in parallel with the collector-emitter of the switching element Q2.
By connecting the partial resonance capacitor Cp, the voltage resonance operation can be obtained only when the switching elements Q1 and Q2 are turned off by the capacitance of the partial resonance capacitor Cp and the leakage inductance component L1 of the primary winding N1. That is, a partial voltage resonance circuit is formed.

On the secondary side of the insulating converter transformer PIT in this figure, a center tap is provided for the secondary winding N2, and rectifier diodes DO1, DO2, DO3, DO4 and smoothing capacitors CO1, CO2 are connected as shown in the figure. As a result, two sets of both-wave rectifier circuits are provided, which include a set of [rectifier diodes DO1, DO2, smoothing capacitor CO1] and a set of [rectifier diodes DO3, DO4, smoothing capacitor CO2]. A double-wave rectifier circuit composed of [rectifier diodes DO1, DO2, smoothing capacitor CO1] generates a DC output voltage EO1, and a double-wave rectifier circuit composed of [rectifier diodes DO3, DO4, smoothing capacitor CO2] generates a DC output voltage EO2. To do.
In this case, the DC output voltage EO1 and the DC output voltage EO2 are also branched and input to the control circuit 1. In the control circuit 1, the DC output voltage EO1 is used as a detection voltage, and the DC output voltage EO2 is used as an operating power source.
Note that, for example, Schottky diodes are selected for the rectifier diodes DO1 and DO2 and the rectifier diodes DO3 and DO4 in accordance with the high-speed switching operation (rectification operation) corresponding to the switching cycle.

  The control circuit 1 performs constant voltage control by supplying, as a control current, a direct current whose level is varied according to the level of the secondary side direct-current output voltage E01, for example, to the control winding NC of the drive transformer PRT. .

As a switching operation of the power supply circuit having the above configuration, when commercial AC power is first turned on, for example, a starting current is supplied to the bases of the switching elements Q1 and Q2 via the starting resistors RS1 and RS2, for example. If the switching element Q1 is turned on first, the switching element Q2 is controlled to be turned off. Then, as an output of the switching element Q1, a resonance current flows from the resonance current detection winding ND → the primary winding N1 → the series resonance capacitor C1, but when the resonance current becomes zero, the switching element Q2 is turned on, and the switching element Q1 is Controlled to be off. Then, a resonant current in the opposite direction flows through the switching element Q2. Thereafter, a self-excited switching operation in which the switching elements Q1, Q2 are alternately turned on is started.
In this way, the switching elements Q1 and Q2 alternately open and close using the terminal voltage of the smoothing capacitor Ci as the operating power supply, thereby supplying a drive current close to the resonance current waveform to the primary winding N1 of the insulating converter transformer PIT. An alternating output is obtained at the next winding N2.

As described above, the control circuit 1 is determined by supplying, for example, a DC current whose level is variable according to the level of the DC output voltage E01 on the secondary side to the control winding NC of the drive transformer PRT as a control current. Perform voltage control.
That is, by passing a control current according to the level of the DC voltage output EO1 through the control winding NC, the inductances of the drive windings NB1 and NB2 are changed, thereby changing the inductance value forming the self-excited oscillation circuit. To be. If the inductance value is varied in this way, the oscillation frequency of the self-excited oscillation circuit changes, and as a result, the switching frequency is variably controlled. Thus, the drive current supplied to the primary winding N1 of the primary side series resonance circuit is controlled by variably controlling the switching frequency of the switching elements Q1, Q2 in accordance with the level of the DC output voltage EO1. By controlling the energy transmitted to the secondary side, constant voltage control of the secondary side DC output voltage is achieved.
Hereinafter, the constant voltage control method using the above method will be referred to as a “switching frequency control method”.

Next, the configuration of the power factor correction circuit 20 will be described.
The power factor correction circuit 20 employs a configuration as a power regeneration system using a magnetic coupling type.
In the power factor correction circuit 20, a filter choke coil LN, a fast recovery diode D1, and a high frequency inductor L10 are connected in series between the positive output terminal of the bridge rectifier circuit Di and the positive terminal of the smoothing capacitor Ci. .
The filter capacitor CN is inserted between the anode side of the fast recovery diode D1 and the positive terminal of the smoothing capacitor Ci, thereby forming a normal mode low-pass filter together with the filter choke coil LN.
Further, since the parallel resonant capacitor C20 is provided in parallel with the high frequency inductor L10, the parallel resonant capacitor C20 and the high frequency inductor L10 constitute a parallel resonant circuit. Thereby, it has the effect | action which suppresses the raise of the rectification | straightening smoothing voltage Ei when a load becomes light.

  The power factor correction circuit 20 is connected to the above-described primary-side current resonance circuit (N1, C1) with respect to the connection point between the cathode of the fast recovery diode D1 and the choke coil LS. The current / voltage by the switching output obtained in the resonance circuit is fed back.

In such a power factor correction circuit 20, the primary side series resonance current flowing in the primary side series resonance circuit (C1-N1 (L1)) is regenerated as electric power according to the switching output of the switching elements Q1, Q2, By feeding back to the smoothing capacitor Ci through the parallel connection of the inductance L10 // capacitor C20, the fast recovery diode D1 is switched when the positive / negative absolute value of the AC input voltage VAC is 1/2 or more. Will work.
As a result, the charging current to the smoothing capacitor Ci flows even during a period in which the rectified output voltage level is lower than the voltage across the smoothing capacitor Ci.
As a result, the average waveform of the AC input current approaches the waveform of the AC input voltage, and the conduction angle of the AC input current is expanded. As a result, the power factor is improved.

Next, FIG. 7 shows another configuration example of the switching power supply circuit configured based on the invention previously filed by the present applicant.
The power supply circuit shown in FIG. 7 is also provided with a current resonance type converter in which two switching elements are half-bridge coupled, and the drive system is a separate excitation type. Also in this case, the power factor improving circuit 21 for improving the power factor is provided.
Note that the same portions as those in FIG. 6 are denoted by the same reference numerals and description thereof is omitted.

In the primary side current resonance type converter shown in this figure, for example, a MOS-FET is selected as the two switching elements Q1 and Q2 connected by the half bridge coupling method.
Here, the drain of the switching element Q1 is connected to the line of the rectified smoothing voltage Ei, the source of the switching element Q1 and the drain of the switching element Q2 are connected, and the source of the switching element Q2 is connected to the primary side ground. The half-bridge connection is compatible with the excitation type.
These switching elements Q1 and Q2 are switched by the oscillation drive circuit 2 so that the on / off operation is alternately repeated, and the rectified and smoothed voltage Ei is intermittently used as a switching output.
In this case, clamp diodes DD1 and DD2 connected in the direction shown in the figure are provided between the drain and source of each switching element Q1 and Q2.

  In this case, one end of the primary winding N1 of the insulating converter transformer PIT is connected to the connection point (switching output point) of the source and drain of the switching elements Q1 and Q2, thereby connecting the primary winding N1 to the primary winding N1. In contrast, a switching output is supplied. The other end of the primary winding N1 is connected to the anode point of the fast recovery diode D1 of the power factor correction circuit 21 via the series resonant capacitor C1.

Also in this case, a current resonance circuit in which the switching power supply circuit is a current resonance type is formed by the capacitance of the series resonance capacitor C1 and the leakage inductance component L1 of the insulating converter transformer PIT including the primary winding N1.
A partial voltage resonance circuit is formed by the partial resonance capacitor Cp connected in parallel between the drain and source of the switching element Q2 and the leakage inductance component L1 of the primary winding N1.

In this case, the control circuit 1 outputs, for example, a control signal having a level corresponding to the fluctuation of the DC output voltage EO1 to the oscillation drive circuit 2. The oscillation drive circuit 2 varies the switching frequency by changing the frequency of the switching drive signal supplied from the oscillation drive circuit 2 to each gate of the switching elements Q1 and Q2 based on the control signal supplied from the control circuit 1. I have to. Thereby, the constant voltage control is performed as in the case of FIG.
The start circuit 3 is provided to start the oscillation drive circuit 2 by detecting the voltage or current obtained on the rectifying / smoothing line immediately after the power is turned on, and is additionally wound around the insulating converter transformer PIT. A low-level DC voltage obtained by rectifying N4 with a rectifier diode D30 and a smoothing capacitor C30 is used as an operation power supply.

The power factor improvement circuit 21 shown in this figure employs a configuration of a power factor improvement circuit as a power regeneration system using an electrostatic coupling type.
In the power factor correction circuit 21, a high frequency inductor L10 and a fast recovery type diode D1 are connected in series between the positive output terminal of the bridge rectifier circuit Di and the positive terminal of the smoothing capacitor Ci. Here, the filter capacitor CN is provided in parallel to the series connection circuit of the high frequency inductor L10 and the fast recovery type diode D1. Even in such a connection form, the filter capacitor CN forms a normal mode low-pass filter together with the filter high-frequency inductor L10.
The resonant capacitor C20 is provided in parallel with the fast recovery diode D1.
In addition, a current resonance circuit (N1, C1) is connected to the power factor correction circuit 21 at a connection point between the high frequency inductor L10 and the anode of the fast recovery diode D1.

Also in the power factor correction circuit 21 formed in this manner, the primary-side series resonance current is regenerated as power and fed back to the smoothing capacitor Ci through the capacitor C20, whereby the fast recovery diode D1 is The switching operation is performed when the positive and negative absolute values of the AC input voltage VAC are 1/2 or more.
As a result, the charging current to the smoothing capacitor Ci flows even during the period when the rectified output voltage level is lower than the voltage across the smoothing capacitor Ci, and the average waveform of the AC input current is the waveform of the AC input voltage. As a result, the conduction angle of the AC input current is expanded, so that the power factor is improved.

Here, the insulating converter transformer PIT provided in the power supply circuit shown in FIGS. 6 and 7 has the following structure, for example. First, as the core, an EE type core obtained by combining an E type core made of a ferrite material is provided. Then, after dividing the winding part on the primary side and the secondary side, the primary winding N1 and the secondary winding are wound around the central magnetic leg of the EE core.
In addition, a gap of about 1.0 mm is formed with respect to the central magnetic leg of the EE type core of the insulating converter transformer PIT, and the induced voltage per 1T (turn) of the secondary winding (N2). The number of turns of the secondary winding (N2) and the primary winding (N1) is set so that the level becomes 5V (5V / T).
As a result, a coupling coefficient of about 0.80 to 0.85 is obtained between the primary side winding (N1) and the secondary side winding (N2), and the required leakage inductance (L1). Is to be obtained.

  Further, as a technique for improving the power factor other than those shown in FIGS. 6 and 7, for example, as shown in FIG. 8, a configuration in which a power choke coil PCH is inserted into a commercial AC power supply line is also known. . Such power factor improvement by the power choke coil is also called a choke input method.

FIG. 9 shows the characteristics of the power supply circuit configured by inserting the power choke coil PCH into the commercial AC power supply line as shown in FIG. Each characteristic of the rate PF is shown. This figure also shows the characteristics of a power supply circuit having a configuration in which the power choke coil PCH is not inserted (power factor improvement is not performed) for comparison. In this case, the subsequent stage of the power supply circuit shown in FIG. 8 may have a circuit configuration in which the power factor correction circuits 20 and 21 are omitted from the power supply circuit of FIG. 6 or FIG.
In practice, the power choke coil PCH shown in FIG. 8 has a power factor PF of 0 at the maximum load power (in this case, Po = 125 W) as the load power Po, as in the power factor characteristic shown by the solid line in FIG. The value of the inductance Lc is set so as to be .75 or more. Thereby, the regulation value of the harmonic distortion regulation class D of the current home appliances / general-purpose electronic devices can be satisfied, for example. Note that 10 mH was selected as the actual value of the inductance Lc.

Japanese Patent Laid-Open No. 2003-189614 (FIGS. 6, 7, and 11)

However, as shown in FIGS. 6 and 7, the power supply circuit having the power factor correction circuit has the following problems.
In the power supply circuits shown in these figures, the magnetic feedback type and the electrostatic coupling type are adopted as the power feedback (regeneration) methods by the power factor correction circuits 20 and 21, respectively. The primary side series resonance current flowing through the power is regenerated and fed back to the smoothing capacitor Ci. This means that the rectification current path of the commercial AC power supply and the primary side series resonance circuit to which the switching output is supplied are connected to form a feedback path. For this reason, the current of the commercial alternating current cycle is superimposed on the primary side series resonance current flowing in the primary winding N1 of the insulating converter transformer PIT.
As a result, the ripple voltage of the commercial power supply cycle of the DC output voltages E01 and E02 on the secondary side increases from before the power factor improvement. For example, in the case of a circuit configuration in which the circuit portions of the power factor improvement circuits 20 and 21 are not provided in FIGS. 6 and 7, the power factor PF is about 0.55, but the power factor PF = When configured to obtain about 0.8, the ripple voltage increases 5 to 6 times.

  As a countermeasure, the capacitance of the smoothing capacitors C01 and C02 for smoothing the DC output voltage must be increased 5 to 6 times. That is, even if the gain of the control circuit 1 is improved as much as possible, in order to make it equal to the circuit before the power factor improvement, it is necessary to increase the capacitances of the smoothing capacitors C01 and C02 by 5 to 6 times. The cost will be increased, and practical application will be impractical. Since such a countermeasure is not practical, it is difficult to employ the circuits as shown in FIGS. 6 and 7 in a device that strictly requires, for example, the ripple voltage to be below a certain level.

Therefore, if the power factor is improved by the choke input method as shown in FIG. 8, the above-described ripple problem is solved.
However, the power choke coil PCH required in such a choke input system is problematic in that it increases in size and increases in weight, size, and cost as load power increases.
Furthermore, a place where there is no influence of the leakage magnetic flux must be selected as the arrangement position of the power choke coil PCH. Alternatively, since measures to prevent the influence of leakage magnetic flux are necessary, problems such as difficulty in layout design on the substrate and an increase in the substrate size and weight due to the need for a shield member arise. .
In other words, as a power factor improvement technology, the power regeneration method currently has a problem of increased ripple, but the choke input method has problems such as an increase in circuit weight, size and cost due to the power choke coil PCH. The situation of becoming. From this, it can be said that there is a demand for a configuration for improving the power factor that can solve the problem of ripple increase, and that can be reduced in size and weight.

Therefore, in the present invention, in view of the above problems, the switching power supply circuit is configured as follows.
That is, first, a rectifying / smoothing unit that generates a rectified and smoothed voltage by inputting a commercial AC power supply, a switching unit that includes a switching element that performs switching by inputting the rectified and smoothed voltage as a DC input voltage, and the switching Switching drive means for switching and driving the element.
Then, at least a primary winding to which a switching output obtained by the switching operation of the switching means is supplied and a secondary winding in which an alternating voltage is excited by the switching output obtained in the primary winding are wound. An insulating converter transformer formed at least, a leakage inductance component of a primary winding of the insulating converter transformer, and a capacitance of a primary side series resonant capacitor connected in series to the primary winding, A primary-side series resonance circuit whose operation is a current resonance type.
A secondary-side DC output voltage generator configured to generate a secondary-side DC output voltage by performing an rectification operation by inputting an alternating voltage excited in the secondary winding of the insulating converter transformer; The constant voltage control for the secondary side DC output voltage is performed by controlling the switching drive means according to the level of the secondary side DC output voltage and varying the switching frequency of the switching means. Constant voltage control means and power factor improvement means for improving the power factor.
Then, first, the magnetic flux density of the insulating converter transformer is set to be equal to or less than a predetermined value so that the secondary side rectified current is in a continuous mode regardless of the fluctuation of the secondary side DC output voltage. Yes.
Then, as the power factor improving means, first, the rectification current flowing between the rectification current path between the rectification output terminal of the rectification circuit and the positive terminal of the smoothing capacitor is inserted in series with the rectification current path. A series connection circuit is provided in which a first power factor improving switching element for switching current and a second power factor improving switching element are connected in series.
And a power factor improving series resonance capacitor connected in parallel to the first power factor improving switching element, and a filter capacitor connected in parallel to at least the series connection circuit.
In addition, an end of the primary side series resonance circuit is connected to a connection point between the first power factor improving switching element and the second power factor improving switching element. Is.

The switching power supply circuit of the present invention having the above configuration includes a current resonance type converter as the primary side switching converter. In addition, the power factor is improved using a power regeneration system.
Then, as described above, the magnetic flux density of the insulating converter transformer is set to be lower than the required value, so that the fluctuation of the secondary side DC output voltage, that is, the load fluctuation and the level fluctuation of the commercial AC power supply (AC input voltage). Regardless of this, the secondary side rectification operation is always in a continuous mode that does not produce a period in which the secondary rectification current becomes discontinuous.
In this way, if the magnetic flux density of the insulating converter transformer is kept below the required level and the secondary side rectification operation is set to the continuous mode, it is superimposed on the primary side series resonance circuit, which was supposed to occur when the power factor is improved by the power regeneration method. The ripple voltage of the commercial AC power supply cycle superimposed on the secondary side DC output voltage, which is generated along with the ripple of the commercial AC power supply cycle, is suppressed.

Further, according to the configuration of the power factor improving means according to the present invention, unlike the conventional electrostatic coupling type configuration, the high frequency inductor is omitted, and a power factor improving switching element using, for example, a diode element is provided.
Thus, by adopting a configuration in which the conventional high frequency inductor is omitted, the power factor can be set only by setting the capacitance of the power factor improving series resonant capacitor and the primary side series resonant capacitor. It is possible to set a higher power factor than the conventional configuration in which the setting of the inductance of the high-frequency inductor is also made.
Further, if the high-frequency inductor can be omitted, the power factor variation, which is supposed to be caused by the variation in inductance, can be reduced by the forward voltage drop variation not occurring in the current diode element.

As described above, the switching power supply circuit of the present invention has a configuration for improving the power factor by the power regeneration method, and the ripple voltage of the secondary side DC output voltage is suppressed. The capacitance of the secondary side rectifying and smoothing capacitor for generating the side DC output voltage can be within a practical range. That is, it becomes possible to realize and push forward the practical use of a power supply circuit having a power factor improving configuration based on a power regeneration system more easily than before.
This also eliminates the need for the choke input method as an alternative power factor improvement technique, but this has resulted in a significant reduction in circuit board size and weight as a power circuit having a power factor improvement function. Means that.

  Furthermore, according to the configuration of the power factor improving means of the present invention, a power factor improving switching element (diode element) is provided instead of the conventional high frequency inductor, so that a higher power factor can be obtained. Power factor variation can be further reduced.

Hereinafter, the best mode for carrying out the invention (hereinafter referred to as an embodiment) will be described.
FIG. 1 is a circuit diagram illustrating a configuration example of a switching power supply circuit according to an embodiment. The power supply circuit shown in this figure employs a configuration in which a partial voltage resonance circuit is combined with a current resonance type converter using a half-bridge coupling method by a separate excitation type as a basic configuration on the primary side.

In the power supply circuit shown in FIG. 1, first, a noise filter is formed by filter capacitors CL and CL and a common mode choke coil CMC with respect to the commercial AC power supply AC.
A full-wave rectifying / smoothing circuit including a bridge rectifier circuit Di and one smoothing capacitor Ci is connected to the commercial AC power supply AC that is a subsequent stage of the noise filter. However, in the present embodiment, the power factor correction circuit 10 is provided between the positive output line of the bridge rectifier circuit Di and the positive terminal of the smoothing capacitor Ci. The configuration and operation of the power factor correction circuit 10 will be described later.
When this full-wave rectifying / smoothing circuit inputs a commercial AC power supply AC and performs a full-wave rectifying operation, a rectified and smoothed voltage Ei (DC input voltage) is obtained at both ends of the smoothing capacitor Ci. In this case, the rectified and smoothed voltage Ei is at a level corresponding to an equal magnification of the AC input voltage VAC. In this case, a low speed recovery type is selected for the four rectifier diodes forming the bridge rectifier circuit Di.

  As shown in the figure, the current resonance type converter for switching (intermittently) by inputting the DC input voltage includes a switching circuit in which two switching elements Q1 and Q2 by MOS-FETs are connected by half bridge coupling. Damper diodes DD1 and DD2 are connected in parallel between the drains and sources of the switching elements Q1 and Q2. The anode and cathode of the damper diode DD1 are connected to the source and drain of the switching element Q1, respectively. Similarly, the anode and cathode of the damper diode DD2 are connected to the source and drain of the switching element Q2, respectively. The damper diodes DD1 and DD2 are body diodes provided in the switching elements Q1 and Q2, respectively.

  A partial resonance capacitor Cp is connected in parallel between the drain and source of the switching element Q2. A parallel resonance circuit (partial voltage resonance circuit) is formed by the capacitance of the partial resonance capacitor Cp and the leakage inductance L1 of the primary winding N1. A partial voltage resonance operation in which voltage resonance occurs only when the switching elements Q1, Q2 are turned off is obtained.

  In this power supply circuit, an oscillation / drive circuit 2 is provided to drive the switching elements Q1, Q2 in a switching manner. The oscillation / drive circuit 2 includes an oscillation circuit and a drive circuit, and for example, a general-purpose IC can be used. Then, a drive signal (gate voltage) having a required frequency is applied to the gates of the switching elements Q1 and Q2 by the oscillation circuit and the drive circuit in the oscillation / drive circuit 2. Thereby, the switching elements Q1 and Q2 perform the switching operation so as to be alternately turned on / off at a required switching frequency.

The insulating converter transformer PIT is provided to transmit the switching outputs of the switching elements Q1 and Q2 to the secondary side.
One end of the primary winding N1 of the insulation transformer PIT is connected to the cathode of the switching diode D2, the anode of the switching diode D1, and the capacitor in the power factor correction circuit 10 through a series connection of the primary side series resonant capacitor C1. Connected to the connection point with C20. The other end of the primary winding N1 is connected to a connection point (switching output point) between the source of the switching element Q1 and the drain of the switching element Q2, so that the switching output is transmitted. Yes.

  Here, the insulating converter transformer PIT generates a required leakage inductance L1 in the primary winding N1 of the insulating converter transformer PIT by a structure described later. The leakage inductance L1 and at least the capacitance of the primary side series resonance capacitor C1 form a primary side series resonance circuit. According to the connection mode described above, the switching outputs of the switching elements Q1, Q2 are transmitted to the primary side series resonance circuit. The primary side series resonance circuit performs a resonance operation by the transmitted switching output, thereby making the operation of the primary side switching converter a current resonance type.

According to the above description, the primary side switching converter shown in this figure has the operation as the current resonance type by the primary side series resonance circuit (L1-C1) and the part by the partial voltage resonance circuit (Cp // L1) described above. A voltage resonance operation is obtained.
That is, the power supply circuit shown in this figure has a configuration as a complex resonance type converter in which a resonance circuit for making the primary side switching converter a resonance type is combined with another resonance circuit.

  An alternating voltage corresponding to the switching output transmitted to the primary winding N1 is excited in the secondary winding of the insulating converter transformer PIT. In this case, as the secondary winding N2, secondary windings N2A and N2B which are divided into two parts by providing a center tap are provided. In this case, the secondary windings N2A and N2B have the same predetermined number of turns.

The secondary windings N2A and N2B are provided with a synchronous rectification circuit by double-wave rectification, which includes N-channel MOS-FETs Q3, Q4, Q5, and Q6 as rectification elements as shown in the figure.
For these MOS-FETs Q3 to Q6, for example, a low breakdown voltage trench structure is selected to obtain a low on-resistance.

  The center tap output of the secondary winding N2 of the insulating converter transformer PIT is connected to the connection point of the positive terminal of the smoothing capacitor Co via a series connection of the inductor Ld as shown.

One end of the secondary winding N2 (the end on the secondary winding N2B side) is connected to a connection point between the drain of the MOS-FET Q3 and the drain of the MOS-FET Q5. The connection points of the sources of these MOS-FETs Q3 and Q5 are connected to the secondary side ground.
Similarly, the other end of the secondary winding N2 (the end on the secondary winding N2A side) is connected to the connection point between the drain of the MOS-FET Q4 and the drain of the MOS-FET Q6. The connection point of the sources of the FETs Q4 and Q6 is connected to the secondary side ground.
Note that body diodes DD3, DD4, DD5, and DD6 are connected in parallel to the drain-source of the MOS-FETs Q3, Q4, Q5, and Q6, respectively.

  According to such a connection form, in the rectification current path including the secondary winding N2B, the parallel connection circuit of the MOS-FET Q3 // MOS-FET Q5 which is the rectification element is inserted in series. In the rectified current path including the secondary winding N2A, a parallel connection circuit of MOS-FET Q4 // MOS-FET Q6, which is also a rectifying element, is inserted in series.

In the synchronous rectifier circuit shown in this figure, the drive circuit for driving the MOS-FET Q3 and the MOS-FET Q5 includes the end of the secondary winding N2A on the side not center-tapped and the gates of the MOS-FETs Q3 and Q5. The gate resistor Rg1 is connected so as to be inserted in common.
Similarly, the driving circuit for driving the MOS-FETs Q4 and Q6 is inserted so as to be common between the end portion of the secondary winding N2B which is not center-tapped and the gates of the MOS-FETs Q4 and Q6. It is formed with a gate resistance Rg2.
That is, in this case, the MOS-FETs Q3 and Q5 are turned on (conductive) / off (non-conductive) at the same timing when the alternating voltage excited by the secondary winding N2A is detected by the gate resistor Rg1. The MOS-FETs Q4 and Q6 are also turned on (conducting) / off (nonconducting) at the same timing when the alternating voltage excited by the secondary winding N2B is detected by the gate resistor Rg2. It is what has been.
Also, the pair of MOS-FETs Q3 and Q5 is turned on / off according to the voltage at the end of the secondary winding N2A that is not center-tapped, and the pair of MOS-FETs Q4 and Q6 is the center tap of the secondary winding N2B. Turning on / off in accordance with the voltage at the end that is not performed means that each pair of MOS-FETs is turned on / off in accordance with an alternating voltage having a reverse polarity. That is, the MOS-FETs Q3 and Q5 and the MOS-FETs Q4 and Q6 are rectified so that they are alternately turned on / off according to the timing at which the alternating voltage excited by the secondary winding N2 is inverted. The operation (switching operation) is performed.

Here, when an ON voltage is applied to the gate of the MOS-FET, the drain-source is equivalent to a mere resistor, so that current flows in both directions. If this is to function as a secondary side rectifying element, it is necessary to pass a current only in the direction in which the positive terminals of the secondary side smoothing capacitors (smoothing capacitors Co1 to Co4) are charged. If a current flows in the opposite direction, a discharge current flows from the secondary side smoothing capacitor to the insulating converter transformer PIT side, and power cannot be effectively transmitted to the load side. Further, the MOS-FET generates heat and noise due to the reverse current, resulting in switching loss on the primary side.
Based on the detection of the voltage of the secondary winding, the drive circuit described above is such that only the current in the direction of charging the positive terminal of the secondary-side smoothing capacitor (that is, in this case, the source-to-drain direction) flows. This is a circuit for switching the MOS-FETs Q3 to Q6. In other words, the circuit configuration of the synchronous rectifier circuit in this case employs a configuration in which the MOS-FET is driven on / off in synchronization with the rectified current by the winding voltage detection method.

  In this case, for the gate resistor Rg1 that is to form the drive circuit system of the MOS-FETs Q3 and Q5 and the gate resistor Rg2 that is to form the drive circuit system of the MOS-FETs Q4 and Q6, A Schottky diode Dg1 and a Schottky diode Dg2 are connected in parallel in the direction shown in the figure. By inserting the Schottky diodes Dg1 and Dg2 in this way, the accumulated charges of the gate input capacitances of the MOS-FETs Q3, Q5, Q4, and Q6 correspond to the Schottky diodes Dg1 and Schottky corresponding to these turn-off times, respectively. A path for discharging via the key diode Dg2 is formed. As a result, these MOS-FETs are reliably turned off to obtain good switching characteristics.

Further, as described above, in the power supply circuit shown in FIG. 1, the inductor Ld is inserted in series between the center tap of the secondary winding N2 and the secondary side smoothing capacitor. That is, in this case, the inductor Ld is inserted in the secondary side rectified current path with respect to the line through which the rectified current flows in common during each period in which the secondary side alternating voltage is positive / negative.
By inserting the inductor Ld in this way, it is possible to suppress noise that was supposed to be generated in the secondary side DC output voltage Eo.
When a secondary rectifier circuit is provided with a synchronous rectifier circuit using a MOS-FET, high-frequency noise is easily superimposed on the secondary-side DC output voltage Eo due to the influence of switching noise or the like caused by the MOS-FET. Thus, by inserting the inductor Ld into the rectified current path in this way, the high frequency noise component is suppressed by smoothing with the impedance component.
Further, depending on the inductor Ld inserted in the rectified current path as described above, an effect of suppressing the generation of the reverse current that is supposed to be generated in the secondary side rectified current can be obtained as will be described later.

With the synchronous rectifier circuit having the circuit configuration described so far, an operation of charging a rectified current obtained by rectifying the secondary-side smoothing capacitor by double-wave rectification can be obtained.
That is, in one half cycle of the alternating voltage excited on the secondary side, the current flowing from the secondary winding N2B conducts through the parallel connection circuit of the MOS-FET Q3 // Q5 in the source-to-drain direction, and the smoothing capacitor Charged to Co. Further, in the other half cycle of the alternating voltage, the current flowing through the secondary winding N2A is conducted in the source-to-drain direction through the parallel connection circuit of the MOS-FET Q3 // Q5, and the smoothing capacitor Co is charged. As a result, a double-wave rectification operation in which the smoothing capacitor Co is charged in a period in which the alternating voltage is positive / negative can be obtained.
A secondary side DC output voltage Eo as shown in the figure is obtained as the voltage across the smoothing capacitor. The secondary side DC output voltage Eo is supplied to a load side (not shown) and is also branched and input as a detection voltage for the control circuit 1 described below.

In the present embodiment, the reason why the MOS-FETs as the rectifying elements are connected in parallel in the synchronous rectifier circuit as described above is as follows.
As an example here, as a load condition that the power supply circuit of the present embodiment should actually cope with, the load current varies from 30 A to 0 A with respect to the secondary side DC output voltage Eo = 5V. This is a so-called low voltage / large current load condition in which the fluctuation range is substantially wide. Therefore, if the load tends to be heavy, a considerably large current is also applied to the secondary side rectifier circuit. Flowing. Therefore, the MOS-FET that is a rectifier element is configured to have a plurality of parallel connections, thereby reducing the burden on the element when a large current flows and ensuring high reliability. .

The control circuit 1 supplies a detection output corresponding to the level change of the secondary side DC output voltage Eo to the oscillation / drive circuit 2. The oscillation / drive circuit 2 drives the switching elements Q1 and Q2 such that the switching frequency is varied according to the input detection output of the control circuit 1. By changing the switching frequency of the switching elements Q1 and Q2, the resonance impedance of the primary side series resonance circuit is changed and transmitted from the primary winding N1 of the insulating converter transformer PIT to the secondary winding N2 (N2A, N2B) side. Although the amount of electric power to be changed also changes, this operates so as to stabilize the level of the secondary side DC output voltage Eo.
For example, when the secondary-side DC output voltage Eo decreases due to a heavy load tendency, the switching frequency is controlled to be lowered. However, this reduces the resonant impedance. The secondary side DC output voltage Eo is raised. On the other hand, when the secondary side DC output voltage Eo rises due to a light load tendency, the resonance impedance is increased by controlling the switching frequency to be increased, and the secondary side The DC output voltage Eo is reduced.

Next, the configuration of the power factor correction circuit 10 will be described.
As described above, the power factor correction circuit 10 is provided so as to be inserted into the rectification current path in the rectification smoothing circuit for obtaining the DC input voltage (Ei) from the commercial AC power supply AC. As the power regeneration system, a power factor correction circuit belonging to the electrostatic coupling type is adopted.

In the power factor correction circuit 10, first, the anode of the switching diode D2 (second power factor improvement switching element) is connected to the positive output terminal (rectification output terminal) of the bridge rectification circuit Di. The cathode of the switching diode D2 is connected to the anode of the switching diode D1 (first power factor improving switching element), and the cathode of the switching diode D1 is connected to the positive terminal of the smoothing capacitor Ci. In other words, a switching diode D2 (anode → cathode) -switching diode D1 (anode → cathode) series connection circuit is inserted between the positive electrode output terminal (rectifier output terminal) of the bridge rectifier circuit Di and the positive electrode terminal of the smoothing capacitor Ci. Is done.
In this case, a high speed type (high speed recovery type) is selected for each of the switching diodes D1 and D2.

A filter capacitor CN is connected in parallel to the series connection circuit of the switching diode D2 and the switching diode D1. The filter capacitor CN is provided to suppress normal mode noise.
Further, a power factor improving series resonance capacitor C20 is connected in parallel to the switching diode D1.

  In the power factor correction circuit 10 having such a configuration, as described above, with respect to the connection point between the cathode of the switching diode D2, the anode of the switching diode D1 and the power factor improving series resonance capacitor C20. The primary side series resonance circuit (L1-C1) is connected.

According to such a connection configuration, the power factor improving series resonance capacitor C20 is connected in series to the primary winding N1 of the insulating converter transformer PIT through the series connection of the primary side series resonance capacitor C1. Will be.
According to this, when the primary side series resonant circuit is viewed from the switching output point on the primary side to the positive terminal of the smoothing capacitor Ci, this series resonant capacitor C20 for power factor correction is connected to the primary side series resonant capacitor C1 together with the primary side series resonant capacitor C1. It can be seen that it forms the capacitance of the resonant circuit.

  In the power factor correction circuit 10 having the above configuration, the switching output (primary side series resonance current) obtained in the primary side series resonance circuit is regenerated as power, and the switching diode D1 // the power factor improvement series resonance capacitor C20 is connected in parallel. An operation of returning to the smoothing capacitor Ci through the connection is obtained. In this case, since the capacitance (capacitance) of the power factor improving series resonance capacitor C20 is interposed between the smoothing capacitor Ci and the primary side series resonance circuit, power regeneration is performed by electrostatic coupling.

The waveform diagram of FIG. 3 shows the operation of the power factor correction circuit 10 by the commercial AC power supply cycle.
First, as the voltage V2 generated between the positive electrode output terminal of the bridge rectifier circuit Di and the primary side ground, a waveform in which the timing when the peak level is synchronized with the AC input voltage VAC is obtained as illustrated. The voltage V2 can be regarded as the level of the rectified and smoothed voltage Ei on which the ripple component of the commercial AC power supply period is superimposed.
And since the power regeneration is performed from the primary side series resonance circuit to the power factor improvement circuit 10 side as described above, the connection point between the primary side series resonance capacitor C1 and the power factor improvement circuit 10 side As shown in the figure, the voltage V1 between the primary side ground and the voltage V2 is obtained as a waveform in which an alternating voltage component due to the switching period is superimposed on the voltage V2. The peak level of the voltage V1 changes at a substantially constant level as shown in the figure.

The fast recovery type switching diode D1 performs a switching operation by applying an alternating voltage component in the voltage V1, for example, when the positive / negative absolute value of the AC input voltage VAC is about ½ or more of its peak value. The rectified current is made intermittent.
In this way, the switching diode D1 is switched so that the rectified current is intermittent, whereby the current flowing through the switching diode D2 inserted in series between the switching diode D1 and the positive output terminal of the bridge rectifier circuit Di. As shown in the figure, Id is obtained as an alternating waveform in which the vicinity of the peak value becomes a substantially spire-shaped envelope.
Thus, from the fact that an alternating waveform is obtained as the current Id, it can be seen that the switching diode D2 intermittently rectifies the current together with the switching diode D1.

The rectified current that flows when the switching diode D1 and the switching diode D2 perform the switching operation in this way flows into the smoothing capacitor Ci via the parallel connection circuit of the switching diode D1 // the power factor improving series resonance capacitor C20. .
In such a rectified current conduction period (that is, a period in which the illustrated current Id flows along with the switching operation of D1 and D2), the rectified output voltage level of the bridge rectifier circuit Di is inherently at both ends of the smoothing capacitor Ci. The period of expansion of the AC input current IAC shown in FIG. 3 also substantially coincides with the period of conduction of the rectified current. That is, the conduction angle of the AC input current IAC is larger than that without the power factor correction circuit, and the waveform of the AC input current IAC approaches the waveform of the AC input voltage VAC. Yes. That is, the power factor is improved.

In this case, as the current Ic flowing through the power factor improving series resonance capacitor C20, an alternating waveform according to the switching period is obtained as shown, and a waveform corresponding to the primary side series resonance current is obtained. Understand. In addition, the positive / negative amplitude level of the current Ic is decreased during the period in which the current Id flows.
From such a waveform of the current Ic, the primary side series resonance current flowing into the power factor correction circuit 10 is equal to the switching diode D1 and the power during the period when the rectified current flows along with the switching operation by the switching diodes D1 and D2. It can be understood that only the power factor improving series resonance capacitor C20 is flowing during the period in which the rectified current does not flow through both the ratio improving series resonance capacitor C20.

Here, as understood from the above description, the power factor correction circuit 10 according to the embodiment is a conventional power factor correction circuit 21 having the same electrostatic coupling type as shown in FIG. As compared with the above, a configuration in which the high-frequency inductor L10 is replaced with a switching diode D2 is adopted.
In this way, the power factor correction circuit 10 provided with the switching diode D2 in place of the conventional high-frequency inductor L10 has the following advantages.

First, by providing the switching diode D2 in place of the high-frequency inductor L10 as described above, a higher power factor can be set.
That is, in the conventional electrostatic coupling type configuration, the power factor is set by setting the inductance of the high-frequency inductor L10, and the capacitances of the power factor improving series resonance capacitor C20 and the primary side series resonance capacitor C1. In order to obtain a higher power factor, the inductance value of the high-frequency inductor L10 is set to be larger.
However, as a high value is set as the inductance of the high-frequency inductor L10, the copper loss tends to increase, and the power loss at this portion also increases. For this reason, in the configuration including the conventional high-frequency inductor L10, it is difficult to set a high power factor.
On the other hand, in the configuration of the embodiment in which the high frequency inductor L10 is omitted as described above, it is not necessary to set the inductance for setting the power factor, and the power loss as described above does not occur. The rate can be set.

Further, as understood from the above description, in the embodiment, the power factor can be set only by setting the capacitances of the power factor improving series resonance capacitor C20 and the primary side series resonance capacitor C1. It is.
For example, depending on the setting of the capacitance of the power factor improving series resonance capacitor C20, the impedance component for the primary side series resonance current fed back to the power factor improving circuit 10 can be set. The power level to be fed back to 10 can be set. That is, it is possible to set the peak level of the voltage V1 shown in FIG. Since the power level to be fed back can be set in this way, the power factor to be obtained can be set.
In this way, if the power factor can be set only by setting the capacitances of the power factor improving series resonant capacitor C20 and the primary side series resonant capacitor C1, in addition to these settings, the inductance of the high frequency inductor L10 is also set. Compared with the conventional case that must be done, circuit design can be facilitated.

Furthermore, if the power factor can be set only by setting the capacitances of the power factor improving series resonance capacitor C20 and the primary side series resonance capacitor C1, variations in power factor values can be suppressed.
That is, in the past, setting of the power factor required setting of the inductance of the high-frequency inductor L10 as described above. Therefore, the variation in the value of the power factor is due to the power factor improving series resonance capacitor C20 and the primary side series. In addition to the variation in the capacitance of the resonant capacitor C1, such a variation in the inductance of the high-frequency inductor L10 is also included.
On the other hand, since there is no variation in the forward voltage drop VF as the switching diode D2 in the circuit of the embodiment, the variation in the power factor is the capacitance of the series resonance capacitor C20 for power factor improvement and the capacitance of the primary side series resonance capacitor C1. It depends only on the variation.
For example, in this case, the variation in capacitance of the power factor improving series resonant capacitor C20 and the primary side series resonant capacitor C1 is about ± 3% in both the circuit of FIG. 7 and the circuit of the embodiment, and the high frequency inductor L10 in FIG. Since the inductance variation is about ± 5%, in the case of the circuit of the embodiment, theoretically, the power factor variation can be suppressed by about ± 2%.

By the way, in the configuration of the switching converter as in the power supply circuit shown in FIGS. 6 and 7, the secondary side DC output voltage is stabilized by the variable control of the switching frequency of the primary side switching element. Has been. In the case of adopting such a configuration, for example, in a state where the load tends to be light, stabilization is achieved by controlling the switching frequency to be high. In this state, the secondary-side rectifier circuit operates in a so-called continuous mode in which the period during which the secondary-side rectified current flows through the secondary-side smoothing capacitor is continuous and there is no period of pause.
On the other hand, if the primary side switching frequency is controlled to decrease as the secondary side DC output voltage drops due to heavy load, the secondary side rectification is performed with respect to the secondary side smoothing capacitor. A transition is made to a so-called discontinuous mode in which a current stops flowing and a current discontinuous period occurs. That is, there is a state in which the discontinuous mode is set according to the load variation as the secondary-side double-wave rectification operation.
The secondary side DC output voltage varies depending on the commercial AC power supply AC (AC input voltage VAC), and a constant voltage control operation corresponding to this is also performed, so that the secondary side DC output voltage also depends on the level of the AC input voltage VAC. There will be a state of discontinuous mode.

In the configuration in which the power factor is improved by the power regeneration method (magnetic coupling method, electrostatic coupling type) as in the power supply circuit shown in FIG. 6 and FIG. 7, the ripple on the commercial AC power supply cycle is included in the primary side series resonance current. It has been described that, by superimposing, the ripple voltage of the commercial AC power source period superimposed on the secondary side DC output voltage is significantly increased as compared with the case where the power factor correction circuit is not provided.
As described above, this is mainly caused by the secondary side rectification operation being in the discontinuous mode due to, for example, load fluctuations and fluctuations in the AC input voltage VAC. In other words, the cause is eliminated if the continuous mode is maintained as the secondary side rectification operation regardless of the load fluctuation and the fluctuation of the AC input voltage VAC. This means that an increase in the ripple voltage of the commercial AC power supply cycle superimposed on the side DC output voltage can be effectively suppressed.

Therefore, in the present embodiment, the power supply circuit shown in FIG. 1 has the following configuration.
FIG. 2 is a cross-sectional view showing a structural example of an insulating converter transformer PIT included in the power supply circuit of FIG.
As shown in this figure, the insulating converter transformer PIT includes an EE type core (EE-shaped core) in which E-type cores CR1 and CR2 made of a ferrite material are combined so that their magnetic legs face each other.
And the bobbin B formed with the shape which divided | segmented so that it might mutually become independent about the winding part of a primary side and a secondary side, for example with a resin etc. is provided. The primary winding N1 is wound around one winding portion of the bobbin B. Further, the secondary winding N2 (N2A, N2B) is wound around the other winding portion. By attaching the bobbin B on which the primary side winding and the secondary side winding are wound in this way to the EE type cores (CR1, CR2), the primary side winding and the secondary side winding are different from each other. By the winding area, the center magnetic leg of the EE core is wound. In this way, the structure of the insulating converter transformer PIT as a whole is obtained. In this case, for example, EER-40 is selected on the assumption that the EE type core corresponds to a load condition as described later and a rated level condition of the AC input voltage.

  In addition, a gap G having a gap length of about 1.6 mm is formed on the central magnetic leg of the EE type core as shown in the figure. Thereby, as the coupling coefficient k, for example, a loosely coupled state with k = 0.75 or less is obtained. That is, as a conventional example, it is in a more loosely coupled state than the insulating converter transformer PIT of the power supply circuit shown in FIGS. Note that k = 0.75 was set as the actual coupling coefficient k. The gap G can be formed by making the central magnetic legs of the E-type cores CR1 and CR2 shorter than the two outer magnetic legs.

  Further, the induced voltage level per 1T (turn) of the secondary winding is also lower than that of the power supply circuit shown in FIGS. 6 and 7, so that the primary winding N1 and the secondary windings N2A and N2B Set the number of windings (turns). For example, by setting the primary winding N1 = 65T and the secondary winding N2A = N2B = 2T, the induced voltage level per 1T (turn) of the secondary winding is 2.5 V / T or less.

By setting the structure of the insulating converter transformer PIT and the number of turns of the primary winding N1 and the secondary winding (N2A, N2B), the magnetic flux density in the core of the insulating converter transformer PIT in this case is reduced. Thus, the leakage inductance in the insulating converter transformer PIT increases as compared with the power supply circuits shown in FIGS.
If the magnetic flux density generated in the core of the insulating converter transformer PIT can be reduced to a required level or less in this way, the electric power excited on the secondary side also changes, thereby causing a heavy load and a low AC input voltage. Even in this condition, the secondary side rectification operation can be set to the continuous mode.

  In this way, the continuous mode is obtained even under heavy load and low AC input voltage states, regardless of changes in the secondary side DC output voltage Eo due to load fluctuations, AC input voltage fluctuations, etc. This means that the secondary side rectification operation is performed in the continuous mode. As a result, in the case of adopting the configuration including the power factor correction circuit based on the power regeneration system as in the present embodiment, it is assumed that the secondary side DC output voltage Eo is superposed by the ripple superposed on the primary side series resonance current. In addition, an increase in the ripple of the commercial AC power supply cycle is greatly suppressed.

  Note that the ripple of the commercial AC power supply cycle, which was supposed to be superimposed on the secondary side DC output voltage Eo in this way, can be suppressed because the secondary side rectification operation becomes a continuous mode as described above. This is because the peak level of the secondary side rectified current is suppressed. In addition, as described above with reference to FIG. 2, the magnetic flux density of the insulating converter transformer is reduced to a required level or less, so that the power transmission from the primary side to the secondary side also changes, and the secondary side accordingly. This is also due to the fact that the influence of the ripple on the commercial AC power supply cycle generated in the primary side series resonance current is reduced.

  As a result, the level of the ripple voltage of the commercial AC power supply cycle superimposed on the secondary side DC output voltage also decreases, and it is not necessary to increase the capacitance of the secondary side smoothing capacitor. That is, the practical use of a switching power supply circuit including a power factor correction circuit based on a power regeneration system can be easily realized. For example, in the power supply circuit shown in FIGS. 6 and 7, the ripple voltage has increased 5 to 6 times as compared with the case where the power factor correction circuit is not provided. It is suppressed to the extent.

In addition, by adopting a power regenerative power factor correction circuit in a switching power supply circuit in this way, a power choke coil is inserted into a commercial AC power supply line as a means for improving the power factor. There is no need to do.
As mentioned earlier, the power choke coil is considerably large among power circuit components, and the influence of leakage magnetic flux cannot be ignored, so it is possible to omit such a power choke coil. For example, it is possible to solve problems such as an increase in the size of the power circuit board, an increase in weight, and a problem in layout design on the board.

As a specific example, when the power choke coil PCH is inserted instead of the power factor correction circuits 20 and 21 based on the configuration of the switching converter shown in FIGS. 6 and 7, for example, the weight of the power choke coil PCH Is about 153 g, the occupied volume is 32.4 cubic centimeters, and the mounting area on the printed circuit board is about 10.8 square centimeters.
On the other hand, in this embodiment, as a specific example of the components constituting the power factor correction circuit 10, both the filter capacitor CN and the power factor improving series resonance capacitor C20 may be a withstand voltage product of 200V. In the embodiment, an inductor element as the high-frequency inductor L10 may not be provided, and the switching diode D2 and the switching diode D1 are relatively small and light as elements. Therefore, in the case of the power supply circuit shown in FIG. 1, each component forming the power factor correction circuit 10 can be made small and light, and the power factor correction circuit 10 as a whole can also have the weight of the power choke coil PCH described above. Occupied volume and mounting area are significantly below.

  In addition, as described above, in the embodiment, an inductor element is not required for the power factor correction circuit 10, so that it is not necessary to design the arrangement on the substrate in consideration of the influence of leakage magnetic flux, and the magnetic There is no need to apply a shield. This also facilitates reduction in size and weight of the power circuit board.

  Here, the power supply circuit of the present embodiment shown in FIG. 1 includes a synchronous rectifier circuit using a low-on-resistance MOS-FET as a rectifier element as a rectifier circuit on the secondary side. This is closely related to the fact that the secondary-side double-wave rectification operation is set to the continuous mode regardless of load fluctuations and AC input voltage VAC level fluctuations as described. This point will be described.

Here, the description starts with the power supply circuit shown in FIGS.
As described above, in the power supply circuit adopting the switching frequency control method based on the configuration shown in FIGS. 6 and 7, for example, when the load tends to be heavy or the AC input voltage VAC tends to decrease. The secondary side rectified current flowing through the secondary-side double-wave rectifier circuit becomes a discontinuous mode in which the current does not flow continuously.
Such a discontinuous mode state can be said to be a state in which the secondary side rectified current flows in a period shorter than the period in which the primary side series resonance current flows. And since the rectified current flows in a short period in this way, the peak level of the rectified current at this time becomes relatively high, and accordingly, the conduction loss of each rectifier diode on the secondary side is relatively large. turn into.
In the circuits shown in FIG. 6 and FIG. 7, a corresponding power loss occurs on the secondary side due to the conduction loss of the rectifier diode due to such a discontinuous mode.

  Therefore, as one of the techniques for reducing the secondary-side power loss due to the conduction loss of such a rectifier diode, a synchronous circuit using a low-on-resistance MOS-FET as a rectifier for the secondary-side double-wave rectifier circuit. It is known as a rectifier circuit. For example, a Schottky diode is selected as the secondary rectifier diode of the power supply circuit shown in FIGS. 6 and 7, but the on-resistance is much smaller in, for example, a trench-structure MOS-FET. Therefore, by making the secondary side rectifier circuit a synchronous rectifier circuit, it is possible to reduce conduction loss in the rectifier element and reduce power loss on the secondary side.

  As such a synchronous rectifier circuit, for example, a resistance element for detecting an alternating voltage obtained at the secondary winding N2 (secondary winding N2A, N2B) of the insulating converter transformer PIT is provided, and the detected voltage serves as a rectifier element. The MOS-FET is turned on / off. This is also called a winding voltage detection method.

However, in the discontinuous mode state, even after the charging current to the smoothing capacitor becomes 0 level, the primary side series resonance current having the same polarity flows in the primary winding N1 during the discontinuous period. The polarity of the induced voltage of the winding N2 is not reversed, and during this period, the MOS-FET is not turned off and is kept on.
Since the MOS-FET is turned on even after the charging current to the smoothing capacitor becomes 0 level in this way, a reverse current flows as a rectified current during this period, and the invalidity due to the reverse current is lost. Electricity is generated.
Therefore, in the case of a synchronous rectifier circuit that employs a winding voltage detection method, the conduction loss in the rectifier element is reduced, but the generation of reactive power due to such reverse current effectively increases the overall power conversion efficiency. It is difficult to plan.

  Therefore, as a technique for solving the problem of generation of reactive power due to the reverse rectified current as described above, a synchronous rectifier circuit based on a rectified current detection method is known. This rectified current detection method is a technique for turning off the MOS-FET before the rectified current charged in the secondary-side smoothing capacitor Co becomes 0 level.

As a circuit configuration for this purpose, for example, a current (rectified current) flowing through the secondary winding N2 is detected by a current transformer or the like. The current detected by the current transformer is output as a voltage (detection voltage), and this detection voltage is compared with a predetermined reference voltage by a comparator.
Here, when the rectified current starts flowing so as to charge the smoothing capacitor Co, the rectified current is detected by the current transformer, and a detection voltage corresponding to the rectified current level is input to the comparator. The comparator compares the reference voltage with the detection voltage, and outputs an H level when the detection voltage exceeds the reference voltage, for example. This H level output is applied as an on-voltage from the buffer to the gate of the MOS-FET, which is a rectifying element, to turn on the MOS-FET. As a result, the rectified current flows in the source-to-drain direction of the MOS-FET.
When the level of the rectified current decreases with the passage of time and the detection voltage, which is the output of the current transformer, becomes lower than the reference voltage accordingly, the comparator inverts the output. By outputting this inverted output through the buffer, the gate capacitance of the MOS-FET which is a rectifying element is discharged, and the MOS-FET is turned off.

  With such an operation, the MOS-FET which is a rectifying element is turned off at a timing before the rectified current becomes 0 level. As a result, like the synchronous rectification circuit using the winding voltage detection method, the reverse current does not flow through the MOS-FET in the period in which the rectification current is discontinuous, and no reactive power is generated. Conversion efficiency is increased.

However, as can be seen from the above description, in such a synchronous rectification circuit of the rectification current detection system, at least one set of current transformers and one output of the current transformer correspond to one MOS-FET, and the MOS-FET Requires a relatively complicated drive circuit system for driving the drive. As a result, the circuit configuration becomes complicated, resulting in inconveniences such as reduced manufacturing efficiency, increased costs, and increased circuit board size.
In particular, when the configuration of a current resonance type switching converter is based on the power supply circuit shown in FIGS. 6 and 7, for example, it is necessary to use a double-wave rectifier circuit as a synchronous rectifier circuit. Therefore, two sets of the above-described current transformer and driving circuit system are required corresponding to each half-wave period, and the above-described problem is further increased.

Considering the adoption of a synchronous rectifier circuit on the premise that there is a state of discontinuous mode as the secondary side rectification operation in this way, the merits of the winding voltage detection method and the rectification current detection method are Must be in a trade-off relationship. That is, the winding voltage detection method is disadvantageous in terms of power conversion efficiency, but the circuit configuration is simplified. In contrast, the rectified current detection method is advantageous in terms of power conversion efficiency because reactive power is not generated, but the circuit configuration is complicated.
In other words, as long as there is a condition for the discontinuous mode as the secondary side rectification operation, for example, even if an attempt is made to adopt a synchronous rectification circuit in consideration of power conversion efficiency, a rectification current detection method must be adopted. As a result, the circuit configuration becomes complicated.

On the other hand, in the present embodiment, the configuration of the insulating converter transformer PIT described above allows the secondary-side double-wave rectification operation to be performed in the continuous mode regardless of load fluctuations and AC input voltage fluctuations. It is said.
As described above, the synchronous rectifier circuit provided in the power supply circuit shown in FIG. 1 employs the winding voltage detection method. However, since the continuous mode is always obtained as described above, There is no reactive power. In other words, the present embodiment includes a winding voltage detection type synchronous rectifier circuit, thereby suppressing an increase in circuit scale as a simple circuit configuration and further avoiding an increase in cost, while reducing current consumption. This effectively solves the problem of reduction in power conversion efficiency caused by reactive power in continuous periods.

In summary, the power supply circuit according to the present embodiment firstly sets the insulating converter transformer PIT in a loosely coupled state with respect to the problem of an increase in ripple when a power factor improvement circuit using a power regeneration system is provided. The problem is solved by setting the secondary-side rectification operation to the discontinuous mode.
Furthermore, the secondary-side double-wave rectification operation is set to the discontinuous mode, so that reactive power during the current discontinuity period does not occur in the secondary-side rectifier circuit. A synchronous rectifier circuit is provided. This effectively reduces power loss in the secondary side rectifier circuit by adding a small number of components.

  In the present embodiment, the inductor Ld is inserted into the secondary side rectified current path. However, when the inductance value is appropriately set, the impedance component of the inductor Ld causes the rectified current. An operation of suppressing the reverse current component is obtained. That is, the insertion of the inductor Ld further enhances the suppression of the reverse current during the current discontinuous period.

For reference, the power factor PF and the change characteristics of the rectified smoothing voltage Ei (DC input voltage) are shown in FIGS. 4 and 5 as a result obtained by actually conducting experiments on the power supply circuit shown in FIG. I will show you.
In these figures, FIG. 4 shows each characteristic with respect to fluctuations in the load power Po = 150 W to 0 W when the AC input voltage VAC = 230 V is constant, and FIG. 5 shows the load power Po = 150 W constant. In this case, each characteristic with respect to fluctuation of the AC input voltage VAC = 170V to 264V is shown.
In order to obtain the experimental results shown in these figures, components were selected as follows for the main part of the power supply circuit shown in FIG.

Insulating converter transformer PIT: EER-40 ferrite core, gap G = 1.6mm, primary winding N1 = 65T, secondary windings N2A, N2B = 2T, coupling coefficient k = 0.75
Primary side series resonant capacitor C1 = 0.022μF
Series resonant capacitor for power factor improvement C20 = 0.047μF
Filter capacitor CN = 1μF

First, for the power factor PF, the characteristics shown in FIGS. 4 and 5 are obtained. In particular, in FIG. 4, the power factor PF with respect to the load fluctuation has a characteristic that its value increases as the load tends to be heavy. When the maximum load power Po = 150 W, the power factor PF is about 0.9. Has been improved.
For example, in the circuits shown in FIGS. 6 and 7, the power factor PF is about 0.8, but the high power factor can be obtained in the embodiment in this way. As described above, the switching diode D2 is provided in place of the conventional high-frequency inductor L10.

Also, the DC input voltage Ei tends to increase as the AC input voltage VAC increases as shown in FIG. On the other hand, the DC input voltage Ei in this case has a flat characteristic with respect to load fluctuations as shown in FIG.
The reason why the level of the DC input voltage Ei changes with respect to the load fluctuation in this manner is that the switching diode D2 is provided in place of the high frequency inductor L10. That is, by omitting the high-frequency inductor L10 in this way, only one series resonant circuit is formed on the primary side of the power supply circuit. In this way, the number of series resonance circuits is one, and the resonance point of the primary side series resonance circuit is also one, so that the fluctuation of the DC input voltage Ei with respect to the load fluctuation, that is, the fluctuation of the primary side series resonance current is suppressed. It will be.

The present invention is not limited to the configuration of the power supply circuit described so far.
First, regarding the insulating converter transformer PIT, the structure may be appropriately changed as long as the magnetic flux density is less than the required one, including the core type, for example.
In addition, for example, the switching converter according to each of the above embodiments is based on a separately excited current resonance converter, but may be configured to include a self-excited current resonance converter. In this case, for example, a bipolar transistor can be selected as the switching element. Furthermore, the present invention can also be applied to a current resonance type converter in which four stone switching elements are full-bridge coupled. For example, as the primary side switching elements (Q1, Q2) of the switching converter, an element other than a MOS-FET may be employed as long as it is an element that can be used by another excitation type, such as an IGBT (Insulated Gate Bipolar Transistor). Absent. Also, the constants of the component elements described above may be changed according to actual conditions.
Further, the detailed configuration of the winding voltage detection type synchronous rectification circuit, which is a secondary-side double-wave rectification circuit, may be changed as appropriate.

It is a circuit diagram which shows the structural example of the switching power supply circuit as embodiment of this invention. It is a figure which shows the structural example of the insulation converter transformer with which the power supply circuit of embodiment is equipped. It is a wave form diagram which shows operation | movement of the principal part corresponding to the power factor improvement operation | movement of the power supply circuit of embodiment. It is a figure which shows the power factor with respect to the load fluctuation | variation in the power supply circuit of embodiment, and the characteristic of a rectification | straightening smoothing voltage (DC input voltage). It is a figure which shows the characteristic of the power factor with respect to the fluctuation | variation of alternating current input voltage, and the rectification | straightening smoothing voltage (direct current input voltage) in the power supply circuit of embodiment. It is a circuit diagram which shows the structural example of the conventional switching power supply circuit. It is a circuit diagram which shows the other structural example of the conventional switching power supply circuit. It is a circuit diagram which shows the structural example by a choke input system as a conventional power factor improvement technique. It is a figure which shows the characteristic of the power factor with respect to load fluctuation | variation, and a rectification | straightening smoothing voltage (DC input voltage) in the switching power supply circuit which employ | adopts a choke input system.

Explanation of symbols

  1 control circuit, 2 oscillation / drive circuit, 10 power factor correction circuit, Di bridge rectifier circuit, Ci smoothing capacitor, Q1, Q2 switching element, PIT isolation converter transformer, C1 primary side series resonance capacitor, Cp partial resonance capacitor, N1 primary Winding, N2 (N2A, N2B) Secondary winding, CN filter capacitor, D1, D2 switching diode, C20 power factor improving series resonance capacitor, Q3-Q6 MOS-FET, Rg1, Rg2 gate resistance, Dg1, Dg2 shot Key diode, Ld inductor, Co (secondary side) smoothing capacitor

Claims (2)

Rectifying and smoothing means for inputting a commercial AC power supply and generating a rectified and smoothed voltage;
Switching means formed by including a switching element that performs switching by inputting the rectified and smoothed voltage as a DC input voltage;
Switching driving means for switching and driving the switching element;
At least a primary winding to which a switching output obtained by the switching operation of the switching means is supplied and a secondary winding in which an alternating voltage is excited by the switching output obtained by the primary winding are wound. An isolated converter transformer,
The primary side which is formed by at least the leakage inductance component of the primary winding of the insulating converter transformer and the capacitance of the primary side series resonance capacitor connected in series to the primary winding, and the operation of the switching means is a current resonance type A series resonant circuit;
A secondary side DC output voltage generating means configured to input an alternating voltage excited to the secondary winding of the insulating converter transformer and perform a rectification operation to generate a secondary side DC output voltage;
The switching drive means is controlled according to the level of the secondary side DC output voltage, and the switching frequency of the switching means is varied to perform constant voltage control on the secondary side DC output voltage. Constant voltage control means;
Power factor improving means for improving the power factor, and
The magnetic flux density of the insulating converter transformer is set to be equal to or less than a predetermined value so that the secondary side rectified current is in a continuous mode regardless of the fluctuation of the secondary side DC output voltage, and
The power factor improving means is
In the rectifying / smoothing means, a first power factor improving first inserted in series with respect to the rectified current path between the rectified output terminal of the rectifier circuit and the positive terminal of the smoothing capacitor and switching the rectified current flowing through the rectified current path. A series connection circuit in which the switching element and the second power factor improving switching element are connected in series;
A power factor improving series resonance capacitor connected in parallel to the first power factor improving switching element;
A filter capacitor connected in parallel to at least the series connection circuit,
An end of the primary side series resonance circuit is connected to a connection point between the first power factor improving switching element and the second power factor improving switching element.
A switching power supply circuit. As a rectifier circuit forming the secondary side DC output voltage generating means, a synchronous rectifier circuit by a winding voltage detection method is provided,
The switching power supply circuit according to claim 1.

Images