JP2005210866A - Motor control device - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide a motor control device which performs power running without use of a microcomputer and has high rectifying efficiency. <P>SOLUTION: An arm drive means 30 receives a power running or rectifying mode command, selects and outputs an arm power running driving signal or an arm rectification driving signal to an arm switching element. A phase correction/ driving signal distribution means 31 advances a phase of a magnetic pole position detection signal of an AC motor in accordance with the rotational speed of the AC motor. An arm rectification detection device 32 compares the magnitude of a high-potential terminal potential for a main power supply with that of an output terminal potential and, if the output terminal potential is larger, outputs an upper arm rectification driving signal to an upper arm drive means and, if the output terminal potential is smaller than a low-potential terminal potential for the main power supply, outputs a lower arm rectification driving signal to a lower arm drive means. <P>COPYRIGHT: (C)2005,JPO&NCIPI

Description

  The present invention relates to a motor control device, and more particularly to a motor control device suitable for use in a vehicle AC motor.

  As a conventional power running drive control method for an AC motor, for example, as described in Japanese Patent Application Laid-Open No. 2001-45789, a method of switching a rectangular wave drive signal in accordance with the rotational speed is known. The energization width of the rectangular wave drive signal is switched based on information on the rotational speed and motor efficiency stored in advance.

JP 2001-45789 A

  In order to obtain the maximum torque during powering driving, it is necessary to advance the phase of the applied voltage command of each phase switching element with respect to the phase of each phase induced voltage of the armature coil. This advance angle varies depending on the rotational speed and the interlinkage magnetic flux generated in the exciting coil. Therefore, in the power running drive system described in Japanese Patent Laid-Open No. 2001-45789, a microcomputer is required because the energization width is switched according to the rotational speed based on the information on the rotational speed and motor efficiency stored in advance. Therefore, there is a problem that the cost becomes high.

  Recently, a method of restarting an engine after an idle stop using an alternator (generator) as an electric motor has been studied. At this time, since the engine is in a warm-up state, complicated control is not necessary as compared with cold start, and thus complicated control by a microcomputer or the like is not necessary. From this point of view, the use of a microcomputer increases the cost.

  Conventionally, the rectification of the generated current by the alternator has a problem that the efficiency is low because the diode rectification is performed.

  An object of the present invention is to provide a motor control device that performs power running control without using a microcomputer and has high rectification efficiency.

(1) In order to achieve the above object, the present invention selects the upper arm power running drive signal or the upper arm commutation drive signal in response to the power running or commutation mode command, and drives the upper arm power running to the control terminal of the upper arm switching element. An upper arm driving means for outputting a signal or an upper arm rectification drive signal, and selecting a lower arm powering drive signal or a lower arm rectification drive signal in response to a power running or rectification mode command, and lower arm power running to a control terminal of the lower arm switching element Lower arm drive means that outputs drive signal or lower arm rectification drive signal and magnetic pole position detection signal of AC motor are input, and its phase is advanced according to the number of rotations of AC motor, and upper arm power running drive signal and lower arm power running Distribute to drive signals, output upper arm powering drive signal to the upper arm driving means, and output lower arm powering drive signal to the lower arm driving means. The upper arm that compares the phase correction / drive signal distribution means with the high potential terminal potential for the main power supply and the output terminal potential and outputs the upper arm rectification drive signal to the upper arm drive means when the output terminal potential is large Rectification detection means, lower arm rectification detection means for comparing the magnitude of the output terminal potential and the low potential terminal potential for main power supply and outputting a lower arm rectification drive signal to the lower arm drive means when the output terminal potential is small; Is provided.
With this configuration, power running control can be performed without using a microcomputer, and rectification efficiency can be improved.

  (2) In the above (1), preferably, the phase correction / drive signal distribution means inputs a magnetic pole position detection signal of an AC motor and converts a rotation frequency of the AC motor into a DC voltage; A constant current source whose current value changes according to the output voltage of the frequency-voltage conversion circuit, a triangular wave generating means for generating a triangular wave by a constant current supplied from the constant current source, and the constant current to the triangular wave generating means A capacitor charging switch for discharging the current of the source, a capacitor discharging switch for sucking the current of the constant current source from the triangular wave generating means, a triangular wave output from the triangular wave generating means and a reference voltage, and a voltage A voltage comparator for generating a pulse, and an upper arm powering drive signal based on the potential of the main power supply low potential terminal and the potential of the main power supply low potential terminal. A drive signal distribution circuit that distributes to the lower arm powering drive signal for output to the lower arm drive means, and converts the reference potential of the upper arm powering drive signal from the potential of the low potential terminal for the main power source to the potential of the output terminal And a level shift up circuit for outputting to the upper arm driving means.

  (3) In the above (1), preferably, the upper arm rectification detecting means inputs a high potential terminal potential for main power supply and an output terminal potential, and outputs a negative voltage to the output terminal when the output terminal potential is large. The negative voltage detection means for outputting a positive voltage to the output terminal when the output terminal potential is small, and the amplification means for amplifying the voltage level when the output of the negative voltage detection means is a negative voltage, The lower arm rectification detection means inputs the output terminal potential and the main power supply low potential terminal potential, outputs a negative voltage to the main power supply low potential terminal when the output terminal potential is small, and when the output terminal potential is large It comprises negative voltage detection means for outputting a positive voltage to the output terminal, and amplification means for amplifying the voltage level when the output of the negative voltage detection means is negative voltage.

  (4) In the above (1), preferably, an upper arm switching element in which the drain is connected to the high potential terminal for main power supply, the source is connected to the output terminal, the drain is connected to the output terminal, and the source is And a lower arm switching element connected to the low potential terminal for the main power source.

  According to the present invention, power running control can be performed without using a microcomputer, and rectification efficiency can be improved.

Hereinafter, the configuration of a motor control device according to an embodiment of the present invention will be described with reference to FIGS.
First, the configuration of a motor control system using the motor control device according to the present embodiment will be described with reference to FIG.
FIG. 1 is a block diagram showing a configuration of a motor control system using a motor control device according to an embodiment of the present invention.

  The motor control device 3 according to the present embodiment includes a power module 10 and a power module control circuit 12A. The DC voltage of the battery 5 is converted into an AC voltage by the power module 10 and supplied to the armature coil 16 of the stator of the AC motor 9. The U-phase, V-phase, and W-phase magnetic pole positions are detected by the U-phase magnetic pole position detection means 13U, the V-phase magnetic pole position detection means 13V, and the W-phase magnetic pole position detection means 13W, respectively. The exciting current flowing through the exciting coil 14 of the rotor of the AC motor 9 is controlled by the exciting current control circuit 15. The exciting current control circuit 15 is controlled by an external controller (high-order control device). Further, a mode switching signal for switching to the power running mode or the power generation mode is supplied from the external controller (higher level control device) to the power module control circuit 12A.

In the power module 10, UP is the U-phase upper arm MOSFET, DUP is the UP diode, VP is the V-phase upper MOSFET, DVP is the VP parasitic diode, WP is the W-phase upper arm MOSFET, and DWP is the WP parasitic diode. is there. The drains of the MOSFETs are commonly connected to the positive electrode of the battery 5. Further, UN is a U-phase lower arm MOSFET, DUN is an UN parasitic diode, VN is a V-phase lower arm MOSFET, DVN is a VN parasitic diode, WN is a W-phase lower arm MOSFET, and DWN is a WN parasitic diode. The source of the MOSFET is commonly connected to the negative electrode of the battery and grounded. The source of the U-phase upper arm MOSFET (UP) and the drain of the U-phase lower arm MOSFET (UN) are connected in common and are also connected to the U terminal of the AC motor 2. The source of the V-phase upper arm MOSFET (VP) and the drain of the V-phase lower arm MOSFET (VN) are connected in common and are also connected to the V terminal of the AC motor 2. The source of the W-phase upper arm MOSFET (WP) and the drain of the W-phase lower arm MOSFET (WN) are connected in common and are also connected to the W terminal of the AC motor 2.
In the power module control circuit 12A, the U-phase upper arm drive signal switching circuit 30a switches the power-running drive signal 34a or the rectifying drive signal 33a by the mode switching signal 18 to the gate of the U-phase upper arm MOSFET (UP). Supply. The U-phase lower arm drive signal switching circuit 30b switches the power-running drive signal 34b or the rectification drive signal 33b by the mode switching signal 18 and supplies it to the gate of the U-phase lower arm MOSFET (UN).

  The U-phase upper arm rectification detection / drive circuit 31a compares the magnitude relationship between the positive electrode voltage VB of the battery and the voltage VU of the U terminal, and generates a positive voltage pulse when VU> VB. The U-phase lower arm rectification detection / drive circuit 31b compares the voltage VU at the U terminal with the negative voltage VG (0 V) of the battery, and generates a positive voltage pulse when VU <VG.

  The U-phase advance / drive signal distribution circuit 32 generates a voltage pulse having a phase advanced with respect to the phase of the output voltage pulse signal hu of the U-phase magnetic pole position detecting means 13U during power running. The advance operation method of the U-phase advance / drive signal distribution circuit 32 will be described later with reference to FIG. 2, but the output voltage pulse of the W-phase magnetic pole position detection means 13 W according to the rotation speed of the armature coil 16. By delaying the phase of the signal hw, a voltage pulse having a phase that is apparently advanced with respect to the phase of the output voltage pulse signal hu is generated.

  The V-phase upper arm drive signal switching circuit 40a switches the power running drive signal 44a or the rectified drive signal 43a by the mode switching signal 18 and supplies it to the gate of the V phase upper arm MOSFET (VP). The V-phase lower arm drive signal switching circuit 40b switches the power-running drive signal 44b or the rectification drive signal 43b by the mode switching signal 18 and supplies it to the gate of the V-phase lower arm MOSFET (VN).

  The V-phase upper arm rectification detection / drive circuit 41a compares the magnitude relationship between the positive electrode voltage VB of the battery and the voltage VV of the V terminal, and generates a positive voltage pulse when VV> VB. The V-phase lower arm rectification detection / drive circuit 41b compares the magnitude of the voltage VV at the V terminal and the negative voltage VG (0V) of the battery, and generates a positive voltage pulse when VV <VG.

  The V-phase advance / drive signal distribution circuit 42 generates a voltage pulse having a phase advanced with respect to the phase of the output voltage pulse hv of the V-phase magnetic pole position detecting means 13V during power running.

  The W-phase upper arm drive signal switching circuit 50a switches the power-running drive signal 54a or the rectifying drive signal 53a by the mode switching signal 18 and supplies it to the gate of the W-phase upper arm MOSFET (WP). The W-phase lower arm drive signal switching circuit 50b switches the power-running drive signal 54b or the rectification drive signal 53b by the mode switching signal 18 and supplies it to the gate of the W-phase lower arm MOSFET (WN).

  The W-phase upper arm rectification detection / drive circuit 51a compares the magnitude relationship between the positive electrode voltage VB of the battery and the voltage VW of the W terminal, and generates a positive voltage pulse when VW> VB. The W-phase lower arm rectification detection / drive circuit 51b compares the magnitude of the voltage VW at the W terminal and the negative voltage VG (0 V) of the battery, and generates a positive voltage pulse when VW <VG.

The W-phase advance / drive signal distribution circuit 52 generates a voltage pulse having a phase advanced with respect to the phase of the output voltage pulse signal hw of the W-phase magnetic pole position detecting means 13W during power running.
Next, the circuit operation of this embodiment will be described. In the present embodiment, the operations of the U phase, the V phase, and the W phase are the same, so only the U phase operation will be described. There are two types of functions of the control device of the present embodiment: power running drive control and rectification drive control of the AC generator / alternator 2.

  First, the power running drive control operation will be described. When a signal (mode switching signal 18) for starting the power running control operation from the external controller enters the U-phase upper arm drive signal switching circuit 30a, the U-phase upper arm drive signal switching circuit 30a selects the power running drive signal 34a, and the U phase The same signal as the power running drive signal 34a is output to the gate of the upper arm MOSFET (UP). The low level of the power running drive signal 34a is the same potential as the U terminal, and the high level is a potential equal to or higher than the threshold value of the U-phase upper arm MOSFET (UP). Similarly, when the power running control operation start signal (mode switching signal 18) enters the U-phase lower arm drive signal switching circuit 30b, the U-phase lower arm drive signal switching circuit 30b selects the power running drive signal 34b, and the U phase The same signal as the power running drive signal 34b is output to the gate of the lower arm MOSFET (UN). The low level of the power running drive signal 34b is the same potential as the negative voltage VG (0V) of the battery, and the high level is a potential equal to or higher than the threshold value of the U-phase lower arm MOSFET (UN). To prevent the U-phase upper arm MOSFET (UP) and the U-phase lower arm MOSFET (UN) from being turned on simultaneously, when the power running drive signal 34a is at the high level, the power running drive signal 34b is at the low level, and conversely the power running drive signal. When 34a is at a low level, the power running drive signal 34b is at a high level. Depending on the input capacitance Ciss and gate resistance Rg of the U-phase upper arm MOSFET (UP) and U-phase lower arm MOSFET (UN), the dead time (power running drive signal 34a) is greater than the time constant (product of Ciss and Rg). And a period during which the power running drive signal 34b is simultaneously at the low level) are set as the power running drive signals 34a and 34b.

The U-phase upper arm MOSFET (UP) and the U-phase lower arm MOSFET (UN) start switching by the power running drive signals 34a and 34b, current flows through the armature coil 16, and the rotor begins to rotate. At the same time, each phase magnetic pole position detecting means 13U, 13V, 13W detects the magnetic pole position and outputs voltage pulse signals hu, hv, hw, respectively. The phase relationship of each signal varies depending on the number of magnetic poles of the rotor and the mechanical angle of the magnetic pole position detection means. In this embodiment, the high level and low level of the voltage pulse signals hu, hv, hw are inverted every 180 degrees in electrical angle. doing. The voltage pulse signal hw is input to the U-phase advance / drive signal distribution circuit 32, and the phase of the voltage pulse signal hw is apparently delayed by delaying the phase of the voltage pulse signal hw in accordance with the rotational speed of the armature coil 3. On the other hand, a voltage pulse having an advanced phase is generated, and the voltage pulse is distributed to the upper arm power running drive signal 34a and the lower arm power running drive signal 34b. The detailed configuration of the advance / drive signal distribution circuit 32 will be described later with reference to FIG. The above is the U-phase power running drive control operation, and the same operation is also performed in the V-phase and W-phase.
Next, the rectification drive control operation will be described. When the mode switching signal 18 enters the U-phase upper arm drive signal switching circuit 30a, the U-phase upper arm drive signal switching circuit 30a selects the rectification drive signal 33a and rectifies the gate of the U-phase upper arm MOSFET (UP). The same signal as the drive signal 33a is output. The low level of the rectification drive signal 33a is the same potential as the U terminal, and the high level is a potential equal to or higher than the threshold value of the U-phase upper arm MOSFET (UP). Similarly, when the rectification control operation start signal (mode switching signal 18) enters the U-phase lower arm drive signal switching circuit 30b, the U-phase lower arm drive signal switching circuit 30b selects the rectification drive signal 33b, The same signal as the rectification drive signal 33b is output to the gate of the arm MOSFET (UN). The low level of the power running drive signal 33b is the same potential as the negative voltage VG (0V) of the battery, and the high level is a potential equal to or higher than the threshold value of the U-phase lower arm MOSFET (UN).

  The U terminal, V terminal, and W terminal generate positive induced voltages VU, VV, and VW by the rotation of the rotor. When the positive voltage VB of the battery and the voltage VU of the U terminal are input to the U-phase upper arm rectification detection / drive circuit 31a, the magnitude relationship between the voltage VB and the voltage VU is compared. When VU> VB, the potential VU of the U terminal is compared. A positive voltage is generated with reference to. The potential difference of the positive voltage is a voltage Vra that is equal to or higher than the threshold value Vth of the U-phase upper arm MOSFET (UP). Further, when VU ≦ VB, the U-phase upper arm rectification detection / drive circuit 31a outputs substantially the same potential as the U terminal. In this way, the drive signal 33a at the time of rectification, which is the output of the U-phase upper arm rectification detection / drive circuit 31a, satisfies the voltage Vra (High level) and VU ≦ VB when VU> VB with reference to the potential of the U terminal. When it becomes a voltage pulse of almost 0V (Low level). Further, the U-phase lower arm rectification detection / drive circuit 31b has substantially the same operation.

When the voltage VU of the U terminal and the negative voltage VG of the battery are input to the U-phase lower arm rectification detection / drive circuit 31b, the magnitude relationship between the voltage VU and the voltage VG is compared. When VG> VU, VG (0 V) is obtained. Generate a positive voltage with reference. The potential difference of the positive voltage is a voltage Vrb that is equal to or higher than the threshold value Vth of the U-phase lower arm MOSFET (UN). When VG ≦ VU, the U-phase lower arm rectification detection / drive circuit 31b outputs substantially the same potential as VG (0 V). In this way, the drive signal 33b at the time of rectification, which is the output of the U-phase lower arm rectification detection / drive circuit 31b, is based on VG (0V) when VG> VU, and when VG> VU, VG ≦ VU. The voltage pulse is almost 0V (Low level). The drive signals 33a and 33b at the time of rectification are respectively supplied to the U-phase upper arm MOSFET (UP) and the U-phase lower arm MOSFET via the U-phase upper arm rectification detection / drive circuit 31a and the U-phase lower arm rectification detection / drive circuit 31b. The gate of (UN) is entered, and the U-phase upper arm MOSFET (UP) and the U-phase lower arm MOSFET (UN) are switched. The above is the drive control operation at the time of U-phase MOS rectification, and the same operation is performed for the V-phase and the W-phase.
In this way, the phase of the magnetic pole position voltage pulse can be advanced during powering, and each phase switching element is switched by the voltage pulse, so that the phase of each phase current can be advanced relative to the phase of each phase induced voltage. . As a result, high torque can be obtained even at high rotation. Further, rectification efficiency can be improved by MOS rectification.

Next, the configuration and operation of the advance / drive signal distribution circuit 32 used in the motor control apparatus according to the present embodiment will be described with reference to FIGS. For convenience of explanation, the operation of the U-phase advance / drive signal distribution circuit 32 will be described as an example. However, the configuration of the V-phase advance / drive signal distribution circuit 42 and the W-phase advance / drive signal distribution circuit 52 and The operation is the same.
First, the configuration of the advance / drive signal distribution circuit 32 used in the motor control apparatus according to the present embodiment will be described with reference to FIG.
FIG. 2 is a block diagram showing the configuration of the advance / drive signal distribution circuit used in the motor control apparatus according to the embodiment of the present invention.

  The frequency-voltage (f-V) converter 209 includes a one-shot multivibrator 201 and an integrator 202. The one-shot multivibrator 201 detects the rising edge of the voltage pulse signal hw output from the magnetic pole position detection means 13W, and outputs a voltage pulse having a predetermined pulse width TW1. If the pulse width output by the magnetic pole position detection means 13W is TW2, there is a relationship of TW1 <TW2. The integrator 202 receives the output voltage pulse of the one-shot multivibrator 201, integrates it with time, and outputs a DC voltage VDC. The frequency-voltage (f-V) converter 209 generates a voltage signal VDC proportional to the frequency of the voltage pulse signal hw output from the magnetic pole position detection means 13W, that is, the rotational speed of the AC motor 9.

  The capacity charging / discharging current source 210 includes a capacity charging current source 203a, a capacity discharging current source 203b, a capacity charging switch 204a, a capacity discharging switch 204b, and an inverter 205. The capacity charging current source 203a discharges a direct current from a power supply VCC for operating the U phase advance angle / drive signal distribution circuit in accordance with the voltage value of the direct current voltage VDC. The capacitive discharge current source 203b sucks a direct current according to the voltage value of the direct current voltage VDC. The capacity charging switch 204a receives the voltage pulse signal hw and switches the output current of the capacity charging current source 203a. The inverter 205 receives the voltage pulse signal hw and outputs the inverted voltage pulse signal. The capacitive discharge switch 204b receives the inverted voltage pulse signal 205a output from the inverter 205 and switches the output current of the capacitive discharge current source 203b.

  The capacitor C integrates the direct current to generate a triangular wave. The inclination of the triangular wave changes according to the voltage value of the DC voltage VDC, that is, the rotational speed of the AC motor. The higher the rotational speed of the AC motor, the greater the inclination of the triangular wave. The comparator 206 inputs a triangular wave to the non-inverting input side, inputs a reference voltage (VCC / 2) to the inverting input side, compares these voltage values, and outputs an output voltage pulse hu '.

  The upper and lower arm drive signal distribution circuit 207 distributes the voltage pulse hu 'to the upper arm drive signal and the lower arm drive signal, and outputs an upper arm drive signal 207a and a lower arm power running drive signal 34b. The level shift up circuit 208 is for converting the reference voltage level of the upper arm drive signal 207a from VG (0 V) to VU. The output voltage of the level shift up circuit 208 is the upper arm power running drive signal 34a. The upper arm power running drive signal 34a is the voltage pulse signal described in FIG.

  The one-shot multivibrator 201 is a general circuit and can be constituted by a general-purpose IC or a bipolar transistor. The integrator 202 may use an operational amplifier circuit. The upper and lower arm drive signal distribution circuit 207 can be formed by a general logic gate. The circuit of the level shift up circuit 208 can also be easily created with a switching element and a resistor.

  Next, the configuration and operation of the advance angle circuit in the advance angle / drive signal distribution circuit 32 will be described.

When the voltage pulse hw enters the one-shot multivibrator 201, the one-shot multivibrator 201 detects the rise of the voltage pulse hw, and the period of the pulse width TW1 (seconds) (TW1 <TW2) from the rise time. It is assumed that only a voltage pulse that becomes a high level is generated. When the frequency f (Hz) (> 0) of the voltage pulse hw, the pulse width TW2 (seconds), and duty 50%, TW2 = 1 / (2f), and the frequency of the output voltage pulse of the one-shot multivibrator 201 is The frequency f remains the same as the frequency of the voltage pulse hw. Accordingly, the one-shot multivibrator 201 has a function of converting the voltage pulse hw into a voltage pulse having a frequency f and a pulse width TW1. The one-shot multivibrator is known as a general general-purpose analog IC. Generally, the pulse width TW1 can be arbitrarily set by a time constant of an external capacitor and a resistor. When this voltage pulse enters the integrator 202, it is integrated over time and converted to a DC voltage VDC. There is a proportional relationship between the frequency f and the DC voltage V.

VDC = kf (k> 0) (1)
= VH ・ TW1 / T
= VH ・ TW1 ・ f

And VH is a high level voltage of the voltage pulse hw. The frequency-voltage converter 209 composed of the one-shot multivibrator 201 and the integrator 202 may be a general general-purpose analog IC.

The output VDC of the integrator enters the capacity charging current source 203a and the capacity discharging current source 203b and is converted into a constant current IDC. The relationship between VDC and IDC will be described later with reference to FIG. When the capacitor charging switch 204a is turned on by the voltage pulse signal hw, the current IDC is charged to the capacitor C, and when the capacitor charging switch 204b is turned on by the inverted voltage pulse signal 205a of the voltage pulse signal hw, the capacitor C is discharged by the current IDC. . Let Td (seconds) be the time from the start of charging to the capacitor C until the voltage reaches Vcc / 2.

IDC = C · Vcc / (2Td) (2)

The relationship holds. Since the capacitor C is repeatedly charged and discharged in this manner, a triangular wave is generated and enters the non-inverting input side of the comparator 206. The comparator 206 compares the reference voltage Vcc / 2 on the inverting input side with the triangular wave voltage, and outputs a voltage Vcc (high level) when the voltage on the non-inverting input side is larger than Vcc / 2. When the voltage is less than or equal to Vcc / 2, a voltage of 0 V (Low level) is output. Therefore, the output voltage pulse hu ′ of the comparator 206 is delayed by Td with respect to the phase of the voltage pulse hw. When the delay time Td is converted into an electrical angle θd (degrees),

θd = 360 · Td · f (3)

It becomes. The phase of the voltage pulse hu ′ is delayed by θd (degrees) with respect to the phase of hw, but is advanced by θc (degrees) with respect to the phase of hu. Since the phase of the voltage pulse hw is 120 degrees ahead of the phase of hu,

θc = θd−120 (4)
It becomes. When Expression (4) is transformed using Expression (1), Expression (2), and Expression (3),

θc = {360 · C · Vcc · f / (2 IDC)} − 120 (5)

It becomes.

Here, the relationship between the rotation frequency and the advance angle by the advance / drive signal distribution circuit 32 used in the motor control apparatus according to the present embodiment will be described with reference to FIG.
FIG. 3 is an explanatory diagram of the relationship between the rotation frequency and the advance angle by the advance / drive signal distribution circuit used in the motor control device according to the embodiment of the present invention.

As shown in FIG. 3, between θc and f,

θc = −Kf (K> 0, f1> 0) (6)

If there is a relationship, from Equation (5), the relationship between IDC and f is

IDC = 180 · C · Vcc · f / (120-Kf) (7)

It becomes. However, Kf <120.

Next, the relationship between the rotation frequency and the constant current source current value by the advance / drive signal distribution circuit 32 used in the motor control apparatus according to the present embodiment will be described with reference to FIG.
FIG. 4 is an explanatory diagram of a rotation frequency and a constant current source current value by an advance / drive signal distribution circuit used in the motor control device according to the embodiment of the present invention.

  FIG. 4 is a plot of the relationship of equation (7). It is defined by 0 <f <120 / K and becomes a downwardly convex curve. Therefore, if the constant current sources 203a and 203b satisfying the relationship of the equation (7) are designed, the relationship of the equation (6) always holds, and the phase θc advances as the frequency f increases.

  By configuring the advance / drive signal distribution circuit 32 as described above, the phase of the magnetic pole position voltage pulse can be advanced during powering, and each phase switching element is switched by the voltage pulse. The phase of each phase current can be advanced. As a result, high torque can be obtained even at high rotation.

Next, the configuration of the constant current source 203 (203a, 203b) used in the advance / drive signal distribution circuit 32 of the motor control device according to the present embodiment will be described with reference to FIG.
FIG. 5 is a circuit diagram showing a configuration of a constant current source used in the advance / drive signal distribution circuit of the motor control device according to the embodiment of the present invention.

  Since the current characteristics shown in FIG. 4 are similar to the voltage-current characteristics of the diode, the constant current sources 203a and 203b according to the present embodiment are made using the exponential function characteristics of the diode. The configuration shown in FIG. 5 is only one example of a constant current source circuit corresponding to Equation (7), and other circuit means may be used.

  A DC voltage VDC is input to the voltage input terminal 501. The constant current source 203 includes an operational amplifier 502, a first pnp transistor 503, an npn transistor 504, a first resistor 505, a diode 506, a second resistor 507, a second pnp transistor 508, And an output terminal 509 for outputting a direct current.

  The operational amplifier 502 operates with the power source Vcc, the non-inverting input side is connected to the voltage input terminal 501, the inverting input side is connected to the emitter of the npn transistor 504, and the output is connected to the base of the npn transistor 504. One terminal of the first resistor 505 is connected to the emitter of the npn transistor 504, and the other terminal is connected to the anode of the diode 506 and one terminal of the second resistor. The cathode of the diode 506 is connected to the other terminal of the second resistor 507 and is grounded at the same time. The first pnp transistor and the second pnp transistor are current mirror circuits.

  Next, the operation of the constant current source 203 will be described. When the DC voltage VDC is input to the input terminal 501, the operational amplifier 502 operates so that the voltage on the non-inverting input side becomes VDC, so that the emitter terminal voltage of the npn transistor also becomes VDC, and the first resistor 505 and the second resistor. A current I flows through 507.

  When the resistance values of the first resistor 505 and the second resistor 507 are R1 and R2, respectively, and the on-voltage of the diode is Vd, the current flowing through the first resistor when I × R2 <Vd is I1 = VDC / ( R1 + R2). Further, since the operational amplifier 502 discharges current to the base of the npn transistor 504, the npn transistor 504 operates. If the current amplification factor hFE of the npn transistor 504 is sufficiently large (for example, 300 or more), the collector current is almost the same as the emitter current, so the collector current is I1. The current I1 is discharged from the output terminal 509 via the current mirror circuit.

Further, since the diode 506 is turned on when I1 × R2 ≧ Vd, the current flowing through the first resistor 505 is I2 = (VDC−Vd) / R2. Similarly, the current I2 is discharged from the output terminal 509 via the current mirror circuit. Therefore, when VDC <Vd × (R1 + R2) / R2, the current I1 = VDC / (R1 + R2) is discharged from the output terminal 509, and when VDC ≧ Vd × (R1 + R2) / R2, the current I2 = ( VDC-Vd) / R2 is discharged. When these equations are erased from VDC using equation (1),
When f <Vd × (R1 + R2) / (kR2),

IDC = kf / (R1 + R2) (8)

When f ≧ Vd × (R1 + R2) / (kR2),

IDC = (kf−Vd) / R2 (9)

It becomes. Here, a curve close to Equation (7) can be realized by adjusting R1 and R2. Note that k = VH · TW1 from the equation (1). Here, VH is a high level voltage of the voltage pulse hw.

Next, the configuration of the capacity charging switch 204a and the capacity discharging switch 204b used in the advance / drive signal distribution circuit 32 of the motor control device according to the present embodiment will be described with reference to FIG.
FIG. 6 is a circuit diagram showing a configuration of a capacity charging switch and a capacity discharging switch used in the advance / drive signal distribution circuit of the motor control device according to the embodiment of the present invention. The configuration shown in FIG. 6 is only one example of a capacity charging switch and a capacity discharging switch, and other circuit means may be used.

  The switch circuit 204 includes a constant current input terminal 601, a first npn transistor 602, a second npn transistor 603, a first pnp transistor 604, a second pnp transistor 605, and a third npn transistor 606. A third pnp transistor 607, a fourth pnp transistor 608, a fourth npn transistor 609, a fifth npn transistor 630, a first pMOS transistor 631, and a first nMOS transistor 632. Composed. Reference numeral 631a is a gate terminal of the first pMOS transistor 531, reference numeral 632a is a gate terminal of the first nMOS transistor 632, and reference numeral 633 is a constant current output terminal.

  The constant current input terminal 601 is connected to the output terminal 509 in FIG. 5, and the gate terminals 631a and 632a are both connected to the output of the magnetic pole position detecting means hw in FIG. The first npn transistor 602, the second npn transistor 603, and the third npn transistor 606 constitute a current mirror circuit, and the same current as the current IDC input from the constant current input terminal 601 is supplied to the second npn transistor 603. Inhale from the collector terminal. Similarly, the same current as the current IDC input from the constant current input terminal 601 is also drawn from the collector terminal of the third npn transistor 606.

  The first pnp transistor 604 and the second pnp transistor 605 constitute a current mirror circuit, and further, the emitter of the first pnp transistor 604 and the source of the first pMOS transistor 631 are connected, and the first pnp transistor 604 The collector and the drain of the first pMOS transistor 631 are connected. The first pMOS transistor 631 turns on and off the current IDC of the current mirror circuit. The current mirror circuit and the first pMOS transistor 631 realize the function of a charging constant current switch.

  When the voltage pulse hw of the magnetic pole position detecting means hw is at the high level, the gate and source potential difference of the first pMOS transistor 631 becomes almost zero, and the first pMOS transistor 631 is turned off. At this time, if a current IDC flows through the collector of the first pnp transistor 604, Vd (about 0.7 V) is generated between the emitter and base of the first pnp transistor 604. As a result, Vd (approximately 0.7 V) is also applied between the emitter and base of the second pnp transistor 605, the second pnp transistor 605 is turned on, and a current IDC flows through the collector of the second pnp transistor 605. .

  On the other hand, when the voltage pulse hw of the magnetic pole position detection means hw is at a low level, the gate and source potential difference of the first pMOS transistor 631 becomes large, and when the threshold voltage of the first pMOS transistor 631 is exceeded, The pMOS transistor 631 is turned on. At this time, even if the current IDC flows through the collector of the first pnp transistor 604, only approximately zero volts is generated between the emitter and the base of the first pnp transistor 604. As a result, the voltage between the emitter and base of the second pnp transistor 605 is also almost zero volts, the second pnp transistor 605 is turned off, and no current flows through the collector.

  In this way, the current mirror circuit is turned on / off by the voltage pulse hw, and the current IDC is discharged or stopped to the constant current output terminal.

  The third pnp transistor 607 and the fourth pnp transistor 608 constitute a current mirror circuit, and discharge the same current as the current IDC sucked from the collector of the third npn transistor 607 from the collector of the fourth pnp transistor 608. The purpose of this current mirror circuit is to produce a discharge current IDC used for the discharge constant current switch of the next stage. The fourth npn transistor 609 and the fifth npn transistor 630 constitute a current mirror circuit, and further, the emitter of the fourth npn transistor 609 and the source of the first nMOS transistor 632 are connected, and the fourth npn transistor 609 The collector and the drain of the first nMOS transistor 632 are connected. The first nMOS transistor 632 turns on / off the current IDC of the current mirror circuit. The current mirror circuit and the first nMOS transistor 632 realize the function of a constant current switch for discharge.

  When the voltage pulse hw of the magnetic pole position detection means hw is at the high level, the gate and source potential difference of the first nMOS transistor 632 increases, and when the threshold voltage of the first nMOS transistor 632 is exceeded, the first pMOS transistor 631 is turned on. At this time, even if the current IDC flows through the collector of the fourth npn transistor 609, only zero volts is generated between the emitter and base of the fourth npn transistor 609. As a result, the voltage between the emitter and base of the fifth npn transistor 630 is also almost zero volts, and the fifth npn transistor 630 is turned off and no current flows through the collector.

  On the other hand, when the voltage pulse hw of the magnetic pole position detection means hw is at the low level, the gate and source potential difference of the first nMOS transistor 632 becomes substantially zero, and the first nMOS transistor 632 is turned off. At this time, if a current IDC flows through the collector of the fourth npn transistor 609, Vd (about 0.7 V) is generated between the emitter and base of the fourth npn transistor 609. As a result, Vd (about 0.7 V) is also applied between the emitter and base of the fifth npn transistor 630, the fifth npn transistor 630 is turned on, and a current IDC flows through the collector of the fifth npn transistor 630.

  In this way, the current mirror circuit is turned on / off by the voltage pulse hw, and the current IDC is sucked in or stopped from the constant current output terminal.

Next, the operation of the advance / drive signal distribution circuit 32 of the motor control device according to the present embodiment will be described with reference to FIGS.
7 and 8 are waveform diagrams showing the operation of the advance / drive signal distribution circuit of the motor control device according to the embodiment of the present invention. FIG. 7 shows the advance operation when the frequency f is small (f = f1). FIG. 8 shows the advance operation when the frequency f is large (f = f2).

  First, the advance operation when the frequency f is small (f = f1) will be described with reference to FIG. 7A shows the waveform of the voltage pulse hu, FIG. 7B shows the waveform of the voltage pulse hw and the triangular wave, and FIG. 7C shows the waveform of the output voltage pulse hu ′ of the comparator 206. Yes.

  The phase of the voltage pulse hw shown in FIG. 7B is advanced by 120 degrees with respect to the phase of the voltage pulse hu shown in FIG. The triangular wave shown in FIG. 7B is a voltage pulse that enters the non-inverting input side of the comparator 206 in FIG. The comparator 206 outputs a voltage pulse hu ′ as shown in FIG. 7C compared with the reference voltage Vcc / 2 that enters the inverting input side.

  The phase of the voltage pulse hu ′ shown in FIG. 7C is advanced by θc1 with respect to the phase of the voltage pulse hu shown in FIG. The advance angle θc1 at the frequency f1 is −Kf1. Here, K is IDC1 = 180 · C · Vcc · f1 / (120−Kf1) from Equation (7), IDC1 = kf1 / (R1 + R2) from Equation (8), and k = VH · TW1 from Equation (1). It is required that IDC1 is deleted from. Here, the advance angle θc1 in the case of the constant current circuit shown in FIG. 5 is obtained, but the equation is different when another circuit means is used. Moreover, Formula (8) was applied as f1 was f1 <Vdx (R1 + R2) / (kR2).

  Next, the advance operation when the frequency f is large (f = f2) will be described with reference to FIG. 8A shows the waveform of the voltage pulse hu, FIG. 8B shows the waveform of the voltage pulse hw and the triangular wave, and FIG. 8C shows the waveform of the output voltage pulse hu ′ of the comparator 206. Yes. For comparison with the case of FIG. 7, an example of a frequency twice the frequency of FIG. 7 is shown.

  The phase of the voltage pulse hw shown in FIG. 8B is advanced by 120 degrees with respect to the phase of the voltage pulse hu shown in FIG. The triangular wave shown in FIG. 8B is a voltage pulse that enters the non-inverting input side of the comparator 206 in FIG. The comparator 206 outputs a voltage pulse hu 'shown in FIG. 8C in comparison with the reference voltage Vcc / 2 that enters the inverting input side.

  As shown in FIG. 4, when the frequency is large, increasing the current value IDC of the constant current source makes the gradient of the triangular wave steep, and the advance angle θc becomes larger than that in FIG. The phase of the voltage pulse hu ′ is advanced by θc2 with respect to the phase of the voltage pulse hu ′. The advance angle θc2 at the frequency f2 is −Kf2. Here, K is IDC2 = 180 · C · Vcc · f2 / (120−Kf2) from Equation (7), IDC2 = (kf2−Vd) / R2 from Equation (9), and k = VH · from Equation (1). It is required to erase IDC2 from TW1. Here, the advance angle θc2 in the case of the constant current circuit shown in FIG. 5 is obtained, but the equation is different when another circuit means is used. Further, Expression (9) is applied as f2 is f2 ≧ Vd × (R1 + R2) / (kR2). However, when f2 <Vd × (R1 + R2) / (kR2), Expression (8) needs to be applied.

  As described above, the relationship between the frequency and the advance angle as shown in FIG. 3 can be realized.

Next, the gate drive signals of the U, V, and W phase MOSFETs during powering drive control when the motor control apparatus according to the present embodiment is used will be described with reference to FIG.
FIG. 9 is a waveform diagram of gate drive signals of U, V, and W phase MOSFETs during powering drive control when the motor control device according to the embodiment of the present invention is used.

  As shown in FIG. 9B, the magnetic pole position detection voltages hu, hv, hw of each phase are generated with respect to the induced voltage of each phase shown in FIG. 9A. The phases of these voltage pulses are shifted by 120 degrees. Voltage pulses hup and hun shown in FIG. 9C are drive signals for the U-phase upper arm MOSFET and the lower arm MOSFET, respectively. When each voltage pulse is at high level, the MOSFET is turned on, and when it is at low level, the MOSFET is turned off. When the voltage pulse hup is at high level, the voltage pulse hun is at low level, and when the voltage pulse hup is at low level, the voltage pulse hun is at high level. In addition, a period in which each signal is simultaneously turned off (dead time) is set so that the upper arm MOSFET and the lower arm MOSFET are not turned on at the same time. The phase of the voltage pulse hup is advanced by θc with respect to the phase of the voltage pulse hu. Similarly, voltage pulses hvp and hvn are drive signals for the V-phase upper arm MOSFET and lower arm MOSFET, respectively, and voltage pulses hwp and hwn are drive signals for the W-phase upper arm MOSFET and lower arm MOSFET, respectively.

Next, the configuration of the rectification detection / drive circuit 31 (31a, 31b) used in the motor control device according to the present embodiment will be described with reference to FIG.
FIG. 10 is a block diagram showing a configuration of a commutation detection / drive circuit used in the motor control device according to the embodiment of the present invention. Although the basic configuration and operation of the U-phase upper arm rectification detection / drive circuit 31a and the U-phase lower arm rectification detection / drive circuit 31b shown in FIG. 1 will be described here, the V-phase and W-phase rectification detection / drive circuit are described. The same applies to.

  The rectification detection / drive circuit 31a includes a U-phase upper arm negative voltage detection circuit 1001a and a U-phase upper arm voltage amplification circuit 1002a. The rectification detection / drive circuit 31b includes a U-phase lower arm negative voltage detection circuit 1001b and a U-phase lower arm voltage amplification circuit 1002b. Reference symbol UVp is an upper arm power source, and reference symbol LVp is a lower arm power source. The upper arm power supply UVp outputs a predetermined voltage with the U terminal as a reference. Further, the lower arm power supply LVp outputs a predetermined voltage with reference to the G terminal. Other reference numerals are the same as those in FIG.

  The U-phase upper arm negative voltage detection circuit 1001a compares the magnitudes of the b terminal voltage VB and the U terminal voltage VU, and outputs the calculation result of (VB−VU). The U-phase upper arm negative voltage detection circuit 1001a outputs a positive voltage to the U terminal when VU ≦ VB, and outputs a negative voltage to the U terminal when VU> VB. The U-phase upper arm negative voltage detection circuit 1001a can be realized by a subtracter using an operational amplifier, for example.

  U-phase upper arm voltage amplification circuit 1002a amplifies the negative voltage portion of the output of U-phase upper arm negative voltage detection circuit 1001a to the threshold voltage of U-phase upper arm MOSFET. The U-phase upper arm voltage amplifier circuit 1002a can also be realized by a half-wave rectifier circuit and an amplifier circuit using an operational amplifier.

  On the other hand, the U-phase lower arm negative voltage detection circuit 1001b compares the G terminal voltage VG and the U terminal voltage VU, and outputs the calculation result of (VU−VG). The U-phase lower arm negative voltage detection circuit 1001b outputs a positive voltage to the G terminal when VG ≦ VU, and outputs a negative voltage to the G terminal when VG> VU. The U-phase lower arm negative voltage detection circuit 1001b can be realized by a subtracter using an operational amplifier, for example. U-phase lower arm voltage amplification circuit 1002b amplifies the negative voltage portion of the output of U-phase lower arm negative voltage detection circuit 1001b to the threshold voltage of U-phase lower arm MOSFET. The U-phase lower arm voltage amplifier circuit 1002b can also be realized by a half-wave rectifier circuit and an amplifier circuit using an operational amplifier.

  By configuring as described above, the rectification timing is detected from the U terminal voltage VU and the U-phase upper arm MOSFET and the U-phase lower arm MOSFET are turned on and off to enable the MOS rectification operation. Can be reduced.

Here, the operation of the rectification detection / drive circuit 31 used in the motor control apparatus according to the present embodiment will be described with reference to FIG.
FIG. 11 is a waveform diagram of the commutation detection / drive circuit used in the motor control device according to the embodiment of the present invention.

  In FIG. 11, VB is a B terminal voltage, VU is a U terminal voltage, VG is a G terminal voltage, FIG. 11 (A) shows an induced voltage, and FIG. 11 (B) is an output of the upper arm negative voltage detection circuit 1001a. 11C shows the output of the upper arm amplifier circuit 1002a, FIG. 11D shows the output of the lower arm negative voltage detection circuit 1001b, and FIG. 11E shows the output of the lower arm amplifier circuit 1002b. Is shown.

  As shown in FIG. 11B, the upper arm negative voltage circuit 1001a outputs the voltage value of (VB−VU) based on VU. When VU ≦ VB, a positive voltage is output with reference to VU, and when VU> VB, a negative voltage is output with reference to VU. As shown in FIG. 11C, the upper arm amplifier circuit 1002a amplifies the voltage level of the negative voltage section by half-wave rectifying this negative voltage circuit output.

  As shown in FIG. 11D, the lower arm negative voltage circuit 1001b outputs the voltage value of (VU−VG) based on VG. When VG ≦ VU, a positive voltage is output with reference to VU, and when VG> VU, a negative voltage is output with reference to VU. As shown in FIG. 11E, the upper arm amplifier circuit 1002b amplifies the voltage level of the negative voltage section by half-wave rectifying the negative voltage circuit output.

Next, the gate drive signals of the U, V, and W phase MOSFETs during the rectifying drive control of the motor control device according to the present embodiment will be described with reference to FIG.
FIG. 12 is a waveform diagram of gate drive signals of U, V, and W phase MOSFETs during rectification drive control of the motor control device according to the embodiment of the present invention.

  The voltage pulse RUp in FIG. 12 (B1) is a drive signal for the U-phase upper arm MOSFET, and the voltage pulse RUn in FIG. 12 (B2) is a drive signal for the U-phase lower arm MOSFET. When each voltage pulse is at high level, the MOSFET is turned on, and when it is at low level, the MOSFET is turned off. When the voltage pulse RUp is at a high level, RUn is at a low level, and when RUp is at a low level, RUn is at a high level. In addition, a period in which each signal is simultaneously turned off (dead time) is set so that the upper arm MOSFET and the lower arm MOSFET are not turned on at the same time.

  When the U terminal voltage VU rises above the B terminal voltage VB, RUp goes high, and when the U terminal voltage VU falls below the b terminal voltage VB, RUp goes low. When the U terminal voltage VU falls below the G terminal voltage VG, RUn goes high, and when the U terminal voltage VU rises above the G terminal voltage VG, RUn goes low. The V-phase and W-phase operate similarly. The voltage pulse RVp in FIG. 12 (B3) is a drive signal for the V-phase upper arm MOSFET, and the voltage pulse RVn in FIG. 12 (B4) is a drive signal for the V-phase lower arm MOSFET, as shown in FIG. 12 (B5). Voltage pulse RWp is a drive signal for the W-phase upper arm MOSFET, and voltage pulse RWn in FIG. 12 (B6) is a drive signal for the W-phase lower arm MOSFET.

By configuring as described above, the rectification timing is detected from each terminal voltage VU, VV, VW, and the MOSFETs of the upper and lower arms of each phase are turned on and off to enable the MOS rectification operation. Loss can be reduced.

It is a block diagram which shows the structure of the motor control system using the motor control apparatus by one Embodiment of this invention. It is a block diagram which shows the structure of the advance / driving signal distribution circuit used for the motor control apparatus by one Embodiment of this invention. It is explanatory drawing of the relationship between the rotation frequency by the advance angle and drive signal distribution circuit used for the motor control apparatus by one Embodiment of this invention, and an advance angle. It is explanatory drawing of the rotation frequency and constant current source current value by the advance / drive signal distribution circuit used for the motor control apparatus by one Embodiment of this invention. It is a circuit diagram which shows the structure of the constant current source used for the advance / drive signal distribution circuit of the motor control apparatus by one Embodiment of this invention. It is a circuit diagram which shows the structure of the switch for capacity | capacitance charge used for the advance / drive signal distribution circuit of the motor control apparatus by one Embodiment of this invention, and the switch for capacity | capacitance discharge. It is a wave form diagram which shows operation | movement of the advance / driving signal distribution circuit of the motor control apparatus by one Embodiment of this invention. It is a wave form diagram which shows operation | movement of the advance / driving signal distribution circuit of the motor control apparatus by one Embodiment of this invention. It is a wave form diagram of the gate drive signal of U, V, and W phase MOSFET at the time of powering drive control when using the motor control device by one embodiment of the present invention. It is a block diagram which shows the structure of the commutation detection and drive circuit used for the motor control apparatus by one Embodiment of this invention. It is a wave form diagram of a rectification detection and drive circuit used for a motor control device by one embodiment of the present invention. It is a wave form diagram of a gate drive signal of U, V, and W phase MOSFET at the time of rectification drive control of a motor controller by one embodiment of the present invention.

Explanation of symbols

DESCRIPTION OF SYMBOLS 3 ... Motor control apparatus 9 ... AC motor 10 ... Power module 12A ... Power module control circuit 13 ... Magnetic pole position detection means 15 ... Excitation current control circuit 30, 40, 50 ... Drive signal switching circuit 31, 41, 51 ... Rectification detection Drive circuit 32, 42, 52... Advance / drive signal distribution circuit

Claims (4)

Upper arm driving means for selecting an upper arm powering driving signal or upper arm rectifying driving signal in response to a powering or rectifying mode command and outputting the upper arm powering driving signal or upper arm rectifying driving signal to the control terminal of the upper arm switching element; ,
Lower arm driving means for selecting a lower arm powering driving signal or a lower arm rectifying driving signal in response to a powering or rectifying mode command and outputting the lower arm powering driving signal or the lower arm rectifying driving signal to the control terminal of the lower arm switching element; ,
Inputs the magnetic pole position detection signal of the AC motor, advances the phase according to the rotational speed of the AC motor, distributes it to the upper arm powering drive signal and the lower arm powering drive signal, and the upper arm powering drive signal to the upper arm driving means And a phase correction / drive signal distribution means for outputting a lower arm powering drive signal to the lower arm drive means,
An upper arm rectification detection means for comparing the magnitude of the high potential terminal potential for the main power supply and the output terminal potential and outputting an upper arm rectification drive signal to the upper arm drive means when the output terminal potential is large;
Lower arm rectification detection means for comparing the magnitude of the output terminal potential with the low potential terminal potential for main power supply and outputting the lower arm rectification drive signal to the lower arm drive means when the output terminal potential is small. A motor control device. The motor control device according to claim 1,
The phase correction / drive signal distribution means includes:
A frequency-voltage conversion circuit that inputs a magnetic pole position detection signal of an AC motor and converts the rotation frequency of the AC motor into a DC voltage;
A constant current source whose current value changes according to the output voltage of the frequency-voltage conversion circuit;
A triangular wave generating means for generating a triangular wave by a constant current supplied from the constant current source;
A capacitor charging switch for discharging the current of the constant current source to the triangular wave generating means;
A capacitive discharge switch for drawing the current of the constant current source from the triangular wave generating means;
A voltage comparator that generates a voltage pulse by comparing the triangular wave output from the triangular wave generating means with a reference voltage;
This voltage pulse is distributed to the upper arm powering drive signal based on the potential of the main power supply low potential terminal and the lower arm powering drive signal for output to the lower arm driving means based on the potential of the main power supply low potential terminal. Driving signal distribution circuit to
A motor control device comprising a level shift-up circuit for converting the reference potential of the upper arm powering drive signal from the potential of the low potential terminal for main power supply to the potential of the output terminal and outputting the same to the upper arm drive means . The motor control device according to claim 1,
The upper arm rectification detecting means is
Negative voltage that outputs high voltage terminal potential and output terminal potential for main power supply, outputs negative voltage to output terminal when output terminal potential is large, and outputs positive voltage to output terminal when output terminal potential is small Voltage detection means;
The negative voltage detection means comprises an amplification means for amplifying the voltage level when the output of the negative voltage detection means is a negative voltage,
The lower arm rectification detecting means is
When the output terminal potential and the main power supply low potential terminal potential are input and the output terminal potential is small, a negative voltage is output to the main power supply low potential terminal, and when the output terminal potential is large, the positive voltage is applied to the output terminal. Negative voltage detection means for outputting
A motor control device comprising amplification means for amplifying a voltage level when the output of the negative voltage detection means is a negative voltage. The motor control device according to claim 1, further comprising:
An upper arm switching element having a drain connected to the high potential terminal for main power supply and a source connected to the output terminal;
A motor control apparatus comprising: a lower arm switching element having a drain connected to an output terminal and a source connected to a low potential terminal for main power supply.

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