JP2005130675A - Inverter control method and polyphase current supply circuit - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide a control method for suppressing resonance of an input current. <P>SOLUTION: A speed control operating section 63 creates a first current command value i<SB>m</SB><SP>*</SP>based on the difference between the angular rotational speed ω<SB>m</SB>of a motor and its command value ω<SB>m</SB><SP>*</SP>. A current command value modulating section 641 creates a second current command value i<SB>T</SB><SP>*</SP>by modulating the first current command value i<SB>m</SB><SP>*</SP>based on a modulation factor r taking a value ¾sinθ<SB>in</SB>¾, where θ<SB>in</SB>is the phase angle of an input voltage v<SB>in</SB>. A correction current command value i<SB>corr</SB><SP>*</SP>for preventing the absolute value ¾i<SB>in</SB>¾ of input current from rising is added to the second current command value i<SB>T</SB><SP>*</SP>thus creating a driving current command value i<SB>dq</SB><SP>*</SP>. A current command value compensating section 643 updates the driving current command value i<SB>dq</SB><SP>*</SP>by adding a compensation signal i<SB>comp</SB><SP>*</SP>based on the difference between the absolute command value ¾i<SB>in</SB><SP>*</SP>¾ of input current and the absolute value ¾i<SB>in</SB>¾ of input current to the driving current command value i<SB>dq</SB><SP>*</SP>. <P>COPYRIGHT: (C)2005,JPO&NCIPI

Description

  The present invention relates to inverter technology.

  For example, Patent Document 1 and Non-Patent Document 1 disclose a technique for obtaining a multiphase AC current from a single-phase AC power source without significantly reducing the capacity of the smoothing capacitor and using a power factor improving reactor. In this technology, a DC voltage applied to an inverter that outputs a multi-phase AC current is obtained by pulsating load power in synchronization with the phase of an input voltage supplied from a single-phase AC power source. It pulsates greatly at a frequency almost twice that of.

  By such pulsation, the conduction width of the input current supplied from the single-phase AC power supply can be expanded, and the power factor is improved. On the other hand, even if the DC voltage applied to the inverter has such pulsation, a multiphase AC current can be output by appropriately controlling the switching in the inverter. Such switching control is referred to herein as single-phase capacitorless inverter control.

  In the single-phase capacitorless inverter control, the smoothing capacitor can be reduced in size and a reactor is not required, so that the entire circuit including the rectifier circuit and the inverter is reduced in size, resulting in cost reduction.

  However, if the capacity of the smoothing capacitor is small, the resonance frequency with the reactor of the AC power supply becomes higher than the power supply frequency, and the harmonic component of the input current appears prominently. Therefore, in order to suppress this, Patent Document 1 and Non-Patent Document 1 propose power control that feeds back an input current and makes it a sine wave.

JP 2002-51589 A Jin Haga, Kazuo Saito, Isao Takahashi “Electrolytic Capacitor-less High Power Factor Inverter Control Method for Single-Phase Diode Rectifier Circuit”, Proceedings of National Conference of the Institute of Electrical Engineers of Japan 4-069 (March 2003), page 99

  However, when feedback control is realized by digital control using a microcomputer or the like, the control band is limited depending on the control cycle, and suppression of high-frequency resonance is not easy. This resonance tends to become more prominent as the load becomes lighter.

  An object of the present invention is to provide a technique for easily reducing such resonance.

An inverter control method according to the present invention receives a diode group (2) that performs full-wave rectification by receiving an input voltage (v _in ) and an input current (i _in ) from an AC power supply (1), and receives the output of the diode group. Multiphase drive using a smoothing circuit (3) having a capacitor (31) and an inverter (4) that receives the output of the smoothing circuit and outputs a multiphase alternating current (i _u , i _v , i _w ) This is a method of controlling the switching of the inverter when driving the section (5).

In the first mode, (a) the rotational angular velocity (ω _m ) of the multiphase drive unit, the command value (ω _m ^* ) of the rotational angular velocity, and the phase angle (θ _in ) of the input voltage (v _in ) ) Based on the drive current command value (i _dq ^* ) of the multi-phase drive unit, and (b) the inverter switching operation command value (T _u , Generating T _v , T _w ). However, the step (a) includes (a-1) a first current command value (based on a difference between the rotational angular velocity (ω _m ) of the multi-phase drive unit and the rotational angular velocity command value (ω _m ^* ). generating a i _m ^*), (a a-2) the first current command value, the modulation coefficient changes at half the period of the cycle of the input voltage _{(r) (| sinθ in |} ; sin 2 θ _in ) to generate a second current command value (i _T ^* ), and (a-3) a correction current command value (i _corr ^* ) that prevents the absolute value of the input current from rising. Generating a command value ( _idq ^* ) of the drive current in addition to the second current command value.

A second aspect of the inverter control method according to the present invention is the first aspect, wherein the correction current command value (i _corr ^* ) has a waveform corresponding to a half cycle of a sine wave, and the absolute value of the input current. Fall from the rising edge of (| i _in |).

A third aspect of the inverter control method according to the present invention is the first aspect, wherein the correction current command value (i _corr ^* ) has a waveform corresponding to a half cycle of a sine wave, and the absolute value of the input current. 2. The inverter control method according to claim 1, wherein the inverter starts up after the half cycle has elapsed since the rise of (| i _in |).

A fourth aspect of the inverter control method according to the present invention is the first to third aspects, wherein the step (a) includes (a-4) a command value (| i _in ) of the absolute value of the input current. ^* |) And a compensation signal (i _comp ^* ) based on the difference between the absolute value (| i _in |) of the input current and the command value (i _dq ^* ) of the drive current are added. The method further includes a step of updating the command value ( _idq ^* ).

A fifth aspect of the inverter control method according to the present invention is the fourth aspect, wherein the step (a-4) includes (a-4-1) a sine value (sinθ _in ) of the phase angle of the input voltage. ) And a value obtained by averaging the product of the input current (i _in ) as a first value (i _in1 ), (a-4-2) the modulation coefficient (r) is Multiplying the value of 1 to obtain a command value (| i _in ^* |) of the absolute value of the input current.

A multiphase current supply circuit according to the present invention includes a diode group (2) that receives an input voltage (v _in ) and an input current (i _in ) from an AC power source (1) and performs full-wave rectification, and an output of the diode group And a smoothing circuit (3) having a capacitor (31) for receiving the output, and receiving the output of the smoothing circuit, performing switching control and supplying a multiphase AC current (i _u , i _v , i _w ) to the multiphase drive unit. It is a multiphase current supply circuit provided with the inverter (4) to output.

In the first mode, the first current command value (i _m ^* ) is calculated based on the difference between the rotational angular velocity (ω _m ) of the multiphase drive unit and the rotational angular velocity command value (ω _m ^* ). The generated speed control calculation unit (63) and the first current command value are based on a modulation coefficient (r) (| sinθ _in |; sin ² θ _in ) that changes in a half cycle of the input voltage. A current command value modulation unit (641) that generates a second current command value (i _T ^* ) by modulating the correction current command value (i _corr ^* ) that prevents the absolute value of the input current from rising. A drive current command value generation unit (642) that generates the drive current command value ( _idq ^* ) in addition to the current command value of 2, and a command value for the switching operation of the inverter based on the drive current command value Command value signal generating means (60) for generating (T _u , T _v , T _w ).

A second aspect of the multiphase current supply circuit according to the present invention is the first aspect, wherein the correction current command value (i _corr ^* ) exhibits a waveform corresponding to a half cycle of a sine wave, The absolute value (| i _in |) falls from the rising edge.

A third aspect of the multiphase current supply circuit according to the present invention is the first aspect, wherein the correction current command value (i _corr ^* ) exhibits a waveform corresponding to a half cycle of a sine wave, It rises after the half period has elapsed since the rise of the absolute value (| i _in |).

A fourth aspect of the multiphase current supply circuit according to the present invention is the first to third aspects, wherein the command value (| i _in ^* |) of the absolute value of the input current and the absolute value of the input current are (| i _in |) a compensation signal (i _comp ^*) based on the difference between the is added to the command value of the driving current (i _dq ^*), and updates the command value of the driving current (i _dq ^*) A drive current command value update unit (643) is further provided.

A fifth aspect of the multiphase current supply circuit according to the present invention is the fourth aspect, wherein the drive current command value update unit (644) includes a sine value (sinθ _in ) of a phase angle of the input voltage. An averaging unit (643a) that obtains twice the value obtained by averaging the product of the input current (i _in ) as a first value (i _in1 ), and multiplying the modulation coefficient by the first value A multiplier (643b) for obtaining a command value (| i _in ^* |) of an absolute value of the input current.

  According to the first aspect of the inverter control method and the first aspect of the multiphase current supply circuit according to the present invention, the resonance of the input current can be suppressed by preventing the rising of the input current by the correction current command value. it can.

  According to the second aspect of the inverter control method and the second aspect of the multiphase current supply circuit according to the present invention, since the absolute value of the input current can be suppressed from rising sharply, the effect of suppressing harmonics is high. .

  According to the third aspect of the inverter control method and the third aspect of the multiphase current supply circuit according to the present invention, it is possible to employ the regeneration regardless of the operation state because the regeneration of electric power is not required.

  According to the fourth aspect of the inverter control method and the fourth aspect of the multiphase current supply circuit according to the present invention, the followability of the absolute value of the input current with respect to the command value of the absolute value of the input current is improved, and the resonance Suppress.

  According to the fifth aspect of the inverter control method and the fifth aspect of the multiphase current supply circuit according to the present invention, the command value of the absolute value of the input current can be obtained.

  FIG. 1 is a circuit diagram illustrating a drive device according to an embodiment of the invention. The drive device includes a motor 5 that is a multiphase drive unit, and a multiphase current supply circuit that supplies a multiphase current thereto.

The multiphase current supply circuit includes a diode bridge 2, a smoothing circuit 3, an inverter 4, and a control circuit 6. Power single-phase AC is connected to the diode bridge 2, the input current i _in of the input voltage v _in and single-phase AC of the single-phase AC is supplied. The power source 1 is shown as a series connection of an ideal AC voltage source 11 and an inductive reactor 12.

Diode bridge 2 has a function of performing full-wave rectification, the input voltage v _in to a full-wave rectified input to the smoothing circuit 3. The smoothing circuit 3 has a capacitor 31. Specifically, the output of the diode bridge 2 is received between both ends of the capacitor 31, and the rectified voltage (voltage between both ends) v _dc generated at both ends of the capacitor 31 is output to the inverter 4.

The inverter 4 receives the output of the smoothing circuit 3, and the current i _inv is input to the inverter 4 from one end of the capacitor 31 (the cathode side of the diode constituting the diode bridge 2). Capacitance of the capacitor 31, the voltage across v _dc is set to pulsate at a frequency twice the frequency of the input voltage v _in. For example, the capacity of the capacitor 31 is set to about several tens of μF.

The inverter 4 supplies three-phase currents i _u , i _v , i _w to the motor 5. The currents i _u , i _v , and i _w correspond to the U phase, the V phase, and the W phase, respectively. The inverter 4 is connected to three transistors (upper arm side transistors) each having a collector connected to one end of the capacitor 31 and to the other end of the capacitor 31 (the anode side of the diode constituting the diode bridge 2). And three transistors (lower arm side transistors) having emitters.

Each of the upper arm side transistors is paired with each of the lower arm side transistors for each phase. The emitter of the upper arm side transistor forming the pair and the collector of the lower arm side transistor are connected in common, and currents i _u , i _v , i _w are output from the connection point. On / off switching of each of the upper arm side transistor and the lower arm side transistor is controlled based on the command value signals T _u , T _v , T _w of the switching operation from the control circuit 6. The command value signals T _u , T _v and T _w correspond to the U phase, V phase and W phase, respectively.

  In order to flow the regenerative current from the motor 5, a free wheel diode having an anode connected to the emitter and a cathode connected to the collector is provided for each of the upper arm side transistor and the lower arm side transistor. ing.

The control circuit 6 inputs the currents i _u , i _v , i _w , the rotation angle (mechanical angle) θ _m of the rotor of the motor 5, the phase angle θ _{in of the} input voltage, and the voltage v _dc across the capacitor 31. These quantities can be detected using known techniques. The control circuit 6 also receives a rotational angular velocity (mechanical angular velocity) command value ω _m ^* and a current phase command value β ^* which are the speeds of the motor 5. Based on these values, command value signals T _u , T _v , T _w are generated based on calculations described later.

  FIG. 2 is a block diagram illustrating the configuration of the control circuit 6. The control circuit 6 includes a position / speed calculator 61, a dq coordinate converter 62, a speed control calculator 63, a command value current calculator 64, a dq axis current controller 65, a PWM (Pulse Width Modulation) calculator 66, A PWM timer unit 67 is provided, and each has a function of executing the following calculation.

Based on the mechanical angle θ _m of the rotor of the motor 5, the position / speed calculating unit 61 and the rotational angle (electrical angle) θ _e of the rotor of the motor 5 and the rotational angular speed (the angular speed of the electrical angle) that is the speed of the motor 5 ) Ω _e and rotational angular velocity (mechanical angular velocity) ω _m are obtained and output. The dq coordinate converter 62 obtains a so-called d-axis current i _d and q-axis current i _q from the currents i _u , i _v , i _w and the electrical angle θ _{e of the} motor 5 based on the equation (1).

The speed control calculation unit 63 includes a subtracter 63a and a PI calculation unit 63b. The subtractor 63a takes the difference between the command value ω _m ^* of the mechanical angle of the motor 5 and the angular speed ω _{m of the} mechanical angle, and the difference is subjected to proportional / integral calculation (PI calculation) by the PI calculation unit 63b. ^* the first current command value i _m is output.

The command value current calculation unit 64 includes a current command value modulation unit 641, a resonance suppression correction unit 642, a current command value compensation unit 643, and a dq current command value generation unit 644. The current command value modulation section 641 and modulated using a modulation coefficient r to be described later ^* first current command value i _m, and generates a ^* second current command value i _T. The resonance suppression correction unit 642 generates a drive current command value i _dq ^* by adding a correction current command value i _corr ^* described later to the second current command value i _T ^* . In this sense, the resonance suppression correction unit 642 can also be grasped as drive current command value generation means. Based on the difference between the absolute value | i _in | of the input current i _in and the command value | i _in ^* |, the current command value compensation unit 643 adds a compensation current command value i _comp ^* described in detail later. To update the drive current command value i _dq ^* . In this sense, the current command value compensator 643 can be grasped as a drive current command value updating unit that updates the command value of the drive current.

The dq current command value generation unit 644 outputs the d-axis current command value i _d ^* and the q-axis current i _q ^* based on the equation (2) using the drive current command value i _dq ^* and the current phase command value β ^*. To do. That sine value of a predetermined value with respect to the drive current command value ^{_{i dq * (-β *) (}} -sinβ *) and the cosine value (cos .beta ^*) and multiplied by the respective d-axis current command value i _d ^* and the q-axis A current command value i _q ^* is generated.

The dq-axis current control unit 65 inputs the d-axis current i _d and the q-axis current i _q , the d-axis current command value i _d ^* and the q-axis current i _q ^* , and the angular velocity ω _{e of the} electrical angle, and the equation (3) The d-axis voltage command value v _d ^* and the q-axis voltage command value v _q ^* are output based on In Equation (3), K _d and K _q are proportional gains of d-axis and q-axis, L _d and L _q are motor inductances of d-axis and q-axis, respectively, and φ _a is a motor back electromotive voltage. It is a constant.

The rotation angle (electrical angle) θ _e of the rotor, the d-axis voltage command value v _d ^*, and the q-axis voltage command value v _q ^* are input to the PWM calculation unit 66, and each phase voltage is calculated based on Equation (4). Command values v _u ^* , v _v ^* , v _w ^* are generated.

Further, the both-end voltage v _dc is also input to the PWM calculation unit 66, and using this and each phase voltage command value v _u ^* , v _v ^* , v _w ^* , based on the equation (5), The on-time τ _j (j = u, v, w) of the upper arm side transistor is obtained. However, in the formula (5), a carrier cycle Tc is introduced. When the on-time τ _j exceeds the carrier period Tc, the value is forcibly set to Tc, and when the on-time τ _j is less than 0, the value is forcibly set to 0.

The PWM timer unit 67 stores on-time τ _u , τ _v , τ _w for each carrier period Tc, and controls signals T _u , T _v , T _w for turning on / off each phase transistor in response to the stored time. Is supplied to the inverter 4.

A group of the position / speed calculation unit 61, the dq coordinate conversion unit 62, the dq axis current control unit 65, the PWM calculation unit 66, and the PWM timer unit 67 includes a d axis current command value i _d ^* and a q axis current command value i. command value signal based on the _q ^* T _u, T _v, can be grasped as a command value signal generating means 60 for generating a T _w.

The current command value modulation unit 641 includes a modulation coefficient generation unit 641a and a multiplier 641b. Modulation coefficient generating unit 641a inputs the phase angle theta _in the voltage v _in to generate a modulation coefficient r. The multiplier 641b multiplies the first current command value i _m ^* by the modulation coefficient r to generate a second current command value i _T ^* .

If the ideal input current i _in0 is realized, the harmonic component of the input current i _in can be significantly suppressed. In this case, the input power P _in can be expressed by Equation (6).

Since the input power P _in such is proportional to the square of the sine wave phase theta _in the input voltage v _in, it varies at a frequency twice the frequency of the input voltage v _in.

Further, while the power P _m of the motor 5 as shown in equation (7) is the product of the rotational angular frequency omega _m and the torque tau _m of the motor 5, the efficiency eta ₄ of the inverter 4 to the input power P _in It can also be grasped as electric power multiplied by the efficiency η _{5 of the} motor 5.

The inertia occurs in the motor 5, the rotational angular frequency omega _m is kept substantially constant. Thus, varying at twice the frequency of the input voltage v _in the torque tau _m of the motor, from the viewpoint of controlling to the ideal input current i _in0 the input current i _in. Such control is introduced in, for example, Patent Document 1 described above.

The torque tau _m of the motor 5, when the phase beta of the current flowing through the motor (this command value is the current phase command value beta ^* at a) are constant, the drive current i _dq as shown in equation (8) ( This command value is proportional to the drive current command value i _dq ^* ).

Thus varying at twice the frequency of the input voltage v _in the drive current command value i _dq ^* is desirable from the viewpoint of control to the ideal input current i _in0 the input current i _in. Therefore adopting non-negative coefficients vary at twice the frequency of the voltage v _in the modulation coefficient r.

FIG. 3 is a block diagram illustrating the configuration of the modulation coefficient generator 641a. The modulation coefficient generation unit 641a includes a sine value generation unit 6410 and an absolute value generation unit 6411. The sine value generation unit 6410 inputs the phase angle θ _in to generate the sine value sin θ _in , and in response to this, the absolute value generation unit 6411 generates | sin θ _in | as the modulation coefficient r.

In addition, instead of the absolute value generation unit 6411, a function of generating the square of the input may be used. In this case, sin ² θ _in is adopted as the modulation coefficient r.

In an ideal case, the second current command value i _T ^* may be adopted as the drive current command value i _dq ^* . However, in order to suppress the resonance, it is necessary to generate performed by driving current command value i _dq ^* corrected to ^* the second current command value i _T.

Returning to FIG. 2, the resonance suppression correction unit 642 includes an absolute value generation unit 642a, a resonance suppression pattern superimposing unit 642b, and an adder 642c. The absolute value generation unit 642a generates an absolute value | i _in | of the input current i _in , and the resonance suppression pattern superimposing unit 642b generates a correction current command value i _corr ^* in response to the absolute value | i _in |. The adder 642c produces the drive current command value i _dq ^* by adding the correction current command value i _corr ^* in ^* the second current command value i _T.

FIG. 4 is a block diagram illustrating the configuration of the resonance suppression pattern superimposing unit 642b. The resonance suppression pattern superimposing unit 642b includes a resonance phase calculating unit 6421 and a pattern generating unit 6422. The resonance phase calculation unit 6421 generates the resonance phase θ _r based on the absolute value | i _in | of the input current i _in obtained from the absolute value generation unit 642a. The pattern generation unit 6422 generates a corrected current command value i _corr ^* based on the resonance phase θ _r .

5 to 7 are graphs for explaining the behavior of the corrected current command value i _corr ^* and the effect thereof. Figure 5 is a case of adopting the second current command value i _T ^* as the drive current command value i _dq ^* without adding the correction current command value i _corr ^*, 6 as the correction current command value i _corr ^* FIG. 7 shows a case where the resonance phase θr is effective between 0 and π (rad), and FIG. 7 shows a case where the resonance phase θr is effective between π and 2π (rad) as the corrected current command value i _corr ^*. Show.

The rise of the absolute value | i _in | of the input current i _in is adopted as the reference (0 rad) of the resonance phase θ _r . When the forward current flows through the diodes constituting the diode bridge 2, the voltage applied to the diode is negligible very small compared to the input voltage v _in and the voltage across v _dc, the input voltage v _in voltage across It is also possible to grasp the time when v _dc is _exceeded as a reference for the resonance phase θ _r .

The correction current command value i _corr ^* takes an effective value, for example, within a predetermined period 2Δ from the resonance phase θ _r = 0. 6 and 7, the period during which the correction current command value i _corr ^* takes an effective value is Δ, and in FIGS. 6 and 7, θ _r = 0 and π are adopted as the starting points of the period Δ, respectively. The case where it is done is illustrated.

The corrected current command value i _corr ^* has a waveform obtained by extracting the period Δ from the sine waveform of the period 2Δ, and the period 2Δ is given by 2π (LC) ^1/2 . Here, the value L can adopt the inductance of the inductive reactor 12, and the value C can adopt the capacitance value of the capacitor 31, respectively. The amplitude of the correction current command value i _corr ^*, the capacitance C of the capacitor 31, using the effective value V _in and the angular frequency omega _in the input voltage v _in, the ^{_{K r = 3 1/2 · ω in}} CV in Can be adopted. Therefore, the corrected current command value i _corr ^* is expressed by equation (9).

However, t indicates time, and the reference (value 0) is adopted as the rise of the absolute value | i _in | of the input current i _in . In FIG. 6, i _corr ^* = 0 except for t = 0 to π (LC) ^{1/2 and} in FIG. 7 except for t = π (LC) ^{1/2 to} 2π (LC) ^1/2. Become.

Note that the period Δ may be obtained experimentally. For example, it can be obtained from the oscillation of the absolute value | i _in | of the input current shown in FIG. Further, the amplitude of the corrected current command value i _corr ^* may be obtained experimentally.

When the corrected current command value i _corr ^* takes an effective value during the period Δ after the absolute value | i _in | of the input current i _in rises (resonance phase θ _r is 0 to π (rad)) In FIG. 6, the absolute value | i _in | of the input current i _in can be prevented from rising sharply, so that the effect of suppressing harmonics is high. However, since electric power is regenerated from the load side to the inverter side, it may not be easily realized depending on the operating state, for example, the magnitude of the rotational speed.

The corrected current command value i _corr ^* during the period Δ (resonance phase θ _r is between π and 2π (rad)) after the period Δ has elapsed since the absolute value | i _in | of the input current i _in has risen ^. Takes an effective value (FIG. 7), it is not possible to suppress the steep rise of the absolute value | i _in | of the input current i _in , but the load power is increased, so power regeneration is required. Therefore, it can be adopted regardless of the operating state.

Of course, in the present invention, if the correction current command value i _corr ^* prevents the rise of the absolute value of the input current, the period for taking the effective value is not limited, and the waveform is also limited to a sine wave. It is not a thing. The corrected current command value i _corr ^* expressed by the equation (9) is only a preferable example.

FIG. 8 is a flowchart illustrating the operation of the control circuit 6. First the speed control calculation unit 63 in step S11, the first current command value i _m ^* is generated. In step S12, the current command value modulation unit 641 generates a second current command value i _T ^* . Next, in step S13, the correction current command value i _corr ^* is added to the second current command value i _T ^* by the resonance suppression correction unit 642 to generate a drive current command value i _dq ^* . Next, in step S14, the current command value compensation unit 643 adds a compensation current command value i _comp ^* described later, and updates the drive current command value i _dq ^* . Next, in step S15, current control is performed by the dq current command value generation unit 644 and the dq axis current control unit 65, and in step S16, the command value signals T _u , T _v , T _w is generated.

FIG. 9 is a flowchart illustrating details of step S13. Step S13 has steps S130 to S139. In step S13, first, in step S130, the second current command value i _T ^* is adopted as the drive current command value i _dq ^* . Next phase theta _in the input voltage v _in to determine whether it is zero in step S131. Of course, _{in order} to determine whether or not the phase θ _in is zero, in practice, it is desirable to adopt that the phase θ _in is smaller than a predetermined positive value. If the phase θ _in is zero, the process proceeds to step S132. As the temporary value of the step S132 in the resonance phase theta _r, substituting the above values theta 2 [pi. Also, the value 1 is given to the flag flg. Thereafter, the process proceeds to step S133. If the phase θ _in is other than zero, the process proceeds to step S133 without going through step S132 (that is, without forcibly giving the value 1 to the flag flg).

In step S133, it is determined whether the value of the flag flg is 1 and the absolute value | i _in | of the input current i _in is non-zero. That is, it is determined whether the rising of the absolute value | i _in | of the input current i _in has been detected after the phase θ _in becomes zero. Of course, _{in order} to determine whether the absolute value | i _in | of the input current i _in is non-zero, in practice, it is desirable to adopt that the absolute value | i _in | is larger than a predetermined positive value.

If the rising edge of the absolute value | i _in | of the input current i _in is detected, the determination result in step S133 is Yes, the process proceeds to step S134, and the initial value zero is set in the resonance phase θr. The value of the flag flg is set to zero.

After step S134 is executed, the process proceeds to step S135. Also, if the determination result in step S133 is No, the process proceeds to step S135. In step S135, the resonance phase θ _r is increased by an increment of 2πf _r / f _c . Here, the frequencies f _r and f _c are a resonance frequency and a carrier frequency, respectively, and are represented by 1 / {2π (LC) ^1/2 } and 1 / T _c . Of course, it is determined experimentally as with period Δ for frequency f _r.

  After step S135 is executed, the method shown in FIG. 6 and the method shown in FIG. 7 are selected in step S136. In FIG. 9, symbols (i) and (ii) indicate branches when the method shown in FIG. 6 and the method shown in FIG. 7 are employed.

When the method shown in FIG. 6 is adopted, as described above, an effective value is taken as the correction current command value i _corr ^{* when the} resonance phase θ _r is 0 to π (rad). Therefore, the process proceeds to step S137, and if the resonance phase θ _r is 0 to π, the process proceeds to step S138, and the drive current is incremented by an amount corresponding to the correction current command value i _corr ^* shown in the equation (9). The command value i _dq ^* is increased. In other cases, the process of step S13 ends. Similarly, when the method shown in FIG. 7 is adopted, an effective value is taken as the correction current command value i _corr ^* between the resonance phase θr of π and 2π (rad) as described above. Therefore, the process proceeds to step S139, and if the resonance phase θ _r is π to 2π, the process proceeds to step S138.

If the rising of the absolute value | i _in | of the input current i _in has not yet occurred _in step S133, step S132 is executed and step S134 is not executed, so that the phase θ _r has the value Θ. It is 2π or more. Therefore, in either determination of steps S137 and S139, the process does not proceed to step S138, and therefore, the addition of the correction current command value i _corr ^* is not substantially performed.

Also, once the same is true if the corrected current command value i _corr ^* phase theta _r continues to increase by the step S135 even after taking the effective value. This continues from step S133 through step S134 until the phase θr is set to zero, and the correction current command value i _corr ^* is not substantially added.

  Of course, if one of the method shown in FIG. 6 and the method shown in FIG. 7 is decided, step S136 is unnecessary, and therefore either one of steps S137 and S139 is unnecessary. It is.

Returning to FIG. 2, the configuration and operation of the current command value compensator 643 will be described. The current command value compensation unit 643 updates the input current compensation unit 643d that generates the compensation current command value i _comp ^* and the drive current command value i _dq ^* obtained as an output of the resonance suppression correction unit 642 for update. And an adder 643e.

In an ideal case, the second current command value i _T ^* may be adopted as the drive current command value i _dq ^* . However, even if the drive current command value i _dq ^* is set in this way, the efficiency η ₄ , η ₅ (see equation (7)) varies depending on the variation of the operating point of the inverter 4 and the motor 5, the input current i _in, depending on the charging current to the capacitor 31 is distorted from the ideal input current i _in0.

Therefore, the compensation current command value i _comp ^* is added to update the drive current command value i _dq ^* . Since the input power P _in is ideally proportional to the drive current i _dq , when the input current i _in is larger than the ideal input current i _in0 , the compensation current command value i _comp ^* is reduced, When the input current i _in is smaller than the ideal input current i _in0 , the compensation current command value i _comp ^* is increased.

The current command value compensation unit 643 generates a compensation current command value i _comp ^* based on the difference between the absolute value | i _in | of the input current i _in and its command value | i _in ^* | The output is provided to the input current compensator 643d. The subtractor 643c subtracts the absolute value | i _in | of the input current i _in from the command value | i _in ^* |, and the input current compensator 642e performs PI control, for example, proportional control on the result, and a compensation current command The value i _comp ^* is generated. The absolute value | i _in | of the input current i _in may be separately generated in the current command value compensation unit 643, but the output from the absolute value generation unit 642a included in the resonance suppression correction unit 642 may be used or The value generation unit 642a may be shared by the resonance suppression correction unit 642 and the current command value compensation unit 643.

The current command value compensation unit 643 includes an averaging unit 643a and a multiplier 643b in order to generate a command value | i _in ^* | of the absolute value of the input current i _in . The averaging unit 643a averages the input current i _in to generate the amplitude i _in1 of the fundamental frequency component. The multiplier 643b is a product of the modulation coefficient r obtained from the modulation coefficient generation unit 641a included in the current command value modulation unit 641 and the amplitude i _in1 of the fundamental frequency component, and the command value | i _in ^* of the absolute value of the input current i _in | Is generated. The modulation coefficient generation unit 641a may be provided separately in the current command value compensation unit 643, or may be shared between the current command value modulation unit 641 and the current command value compensation unit 643.

FIG. 10 is a block diagram illustrating the configuration of the averaging unit 643a. The averaging unit 643a includes a sine value generation unit 6430, a multiplier 6431, and an integration calculation unit 6432. The amplitude i _in1 of the fundamental frequency component is calculated by equation (10).

Sine value generation section 6430 generates a sine value sin [theta _in the phase angle theta _in the input voltage v _in, the multiplier 6431 takes the product of the input current i _in and the sine value sin [theta _in, integration calculation portion 6432 the product The result obtained by integrating the phase angle θ _{in from the} interval 0 to 2π is divided by π to obtain the amplitude i _in1 of the fundamental frequency component.

As described above, to update the drive current command value i _dq ^* by adding a compensation current command value i _comp ^*, by performing the switching control of the inverter 4 based on this command value i of the input current i _in The followability of the input current i _in to _in ^* can be improved, and a control system that does not resonate easily can be constructed.

It is a circuit diagram which illustrates the drive device concerning an embodiment of the invention. 3 is a block diagram illustrating a configuration of a control circuit 6. FIG. It is a block diagram which illustrates the composition of modulation coefficient generating part 641a. It is a block diagram which illustrates the composition of resonance suppression pattern superposition part 642b. It is a graph explaining the behavior and effect of the correction current command value i _corr ^* . It is a graph explaining the behavior and effect of the correction current command value i _corr ^* . It is a graph explaining the behavior and effect of the correction current command value i _corr ^* . 3 is a flowchart illustrating an operation of a control circuit 6. 3 is a flowchart illustrating an operation of a control circuit 6. It is a block diagram which illustrates the composition of averaging part 643a.

Explanation of symbols

DESCRIPTION OF SYMBOLS 1 AC power supply 2 Diode bridge 3 Smoothing circuit 31 Capacitor 4 Inverter 5 Motor 6 Control circuit 60 Command value signal generation means 641 Current command value modulation | alteration part 642 Resonance suppression correction | amendment part (drive current command value generation means)
643 Current command value compensator (drive current command value updating means)
643a Averaging unit 643b Multiplier

Claims (10)

A diode group (2) that receives the input voltage (v _in ) and the input current (i _in ) from the AC power source (1) and performs full-wave rectification;
A smoothing circuit (3) having a capacitor (31) for receiving the output of the diode group;
Switching of the inverter when driving the multiphase drive unit (5) using the inverter (4) that receives the output of the smoothing circuit and outputs a multiphase alternating current (i _u , i _v , i _w ) A method of controlling
(A) Based on the rotational angular velocity (ω _m ) of the multi-phase drive unit, the command value (ω _m ^* ) of the rotational angular velocity, and the phase angle (θ _in ) of the input voltage, the drive current of the multi-phase drive unit Generating a command value (i _dq ^* ) of
(B) generating a command value (T _u , T _v , T _w ) for switching operation of the inverter based on the command value of the drive current;
The step (a) includes: (a-1) a first current command value (i _m ) based on a difference between the rotational angular velocity (ω _m ) of the multiphase drive unit and the rotational angular velocity command value (ω _m ^* ). ^* ) Step to generate,
(A-2) The first current command value is modulated based on a modulation coefficient (r) (| sinθ _in |; sin ² θ _in ) that changes in a half cycle of the input voltage, Generating a current command value (i _T ^* ) of
(A-3) A step of generating a command value (i _dq ^* ) of the drive current by adding a correction current command value (i _corr ^* ) that prevents the absolute value of the input current to rise to the second current command value An inverter control method comprising: The inverter control method according to claim 1, wherein the correction current command value (i _corr ^* ) has a waveform corresponding to a half cycle of a sine wave, and falls from the time when the absolute value (| i _in |) of the input current rises. The correction current command value (i _corr ^* ) has a waveform corresponding to a half cycle of a sine wave, and rises after the half cycle has elapsed since the rise of the absolute value (| i _in |) of the input current. The inverter control method according to Item 1. The step (a)
(A-4) The compensation signal (i _comp ^* ) based on the difference between the command value (| i _in ^* |) of the absolute value of the input current and the absolute value (| i _in |) of the input current is driven. 4. The inverter control method according to claim 1, further comprising a step of updating the command value (i _dq ^* ) of the driving current by adding to the command value ( _{id q} ^* ) of the current. 5. The step (a-4) includes
(A-4-1) The value obtained by averaging the product of the sine value (sinθ _in ) of the phase angle of the input voltage and the input current (i _in ) is obtained as the first value (i _in1 ). Steps,
(A-4-2) multiplying the modulation coefficient (r) by the first value to obtain a command value (| i _in ^* |) of the absolute value of the input current. Inverter control method. A diode group (2) that receives the input voltage (v _in ) and the input current (i _in ) from the AC power source (1) and performs full-wave rectification;
A smoothing circuit (3) having a capacitor (31) for receiving the output of the diode group;
A multi-phase current supply circuit comprising an inverter (4) that receives the output of the smoothing circuit, performs switching control, and outputs a multi-phase alternating current (i _u , i _v , i _w ) to a multi-phase drive unit; There,
A speed control calculation unit (63) that generates a first current command value (i _m ^* ) based on a difference between a rotation angular velocity (ω _m ) of the multiphase drive unit and a command value (ω _m ^* ) of the rotation angular velocity. )When,
The first current command value is modulated on the basis of a modulation coefficient (r) (| sinθ _in |; sin ² θ _in ) that changes in a half cycle of the input voltage, and a second current command value ( current command value modulation section (641) for generating i _T ^* ),
A drive current command value generation unit that generates a command value (i _dq ^* ) of the drive current by adding a correction current command value (i _corr ^* ) that prevents the absolute value of the input current to rise to the second current command value (642),
A multi-phase current supply circuit comprising command value signal generating means (60) for generating command values (T _u , T _v , T _w ) for switching operation of the inverter based on the command value of the drive current. The multiphase current supply according to claim 6, wherein the correction current command value (i _corr ^* ) has a waveform corresponding to a half cycle of a sine wave, and falls from the rise of the absolute value (| i _in |) of the input current. circuit. The correction current command value (i _corr ^* ) has a waveform corresponding to a half cycle of a sine wave, and rises after the half cycle has elapsed since the rise of the absolute value (| i _in |) of the input current. Item 7. The multiphase current supply circuit according to Item 6. A compensation signal (i _comp ^* ) based on the difference between the absolute value of the input current (| i _in ^* |) and the absolute value of the input current (| i _in |) i _dq ^*) to be added, the drive current command value and updates the command value (i _dq ^*) of the driving current update unit (643)
The multiphase current supply circuit according to claim 6, further comprising: The drive current command value update unit (643)
An averaging unit (643a) for obtaining twice the value obtained by averaging the product of the sine value (sinθ _in ) of the phase angle of the input voltage and the input current (i _in ) as the first value (i _in1 ); ,
The multiphase current supply circuit according to claim 9, further comprising a multiplier (643b) that multiplies the modulation coefficient by the first value to obtain a command value (| i _in ^* |) of an absolute value of the input current.

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