JP2005046000A - Synchronous motor drive method, compressor drive methods, and their arrangements - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To realize torque control for reducing a low speed vibration with a pulsating load of periodicity, in a condition of maximum efficiency further with a practical constitution. <P>SOLUTION: The torque control includes inverter control means 8,10 for controlling an inverter 5, in order to redundantly adding a fluctuation quantity with respect to both of the amplitude and phase of a current waveform or a voltage waveform when the torque control suppressing a speed fluctuation during a single rotation is conducted by a synchronous motor 6 controlled by the inverter 5, with respect to the load having periodic torque fluctuations. <P>COPYRIGHT: (C)2005,JPO&NCIPI

Description

  The present invention relates to a synchronous motor driving method, a compressor driving method, and these devices, and more specifically, a synchronous motor driving method and device for driving a synchronous motor such as a brushless DC motor using an inverter, and the like. The present invention relates to a method and apparatus for driving a compressor by a driven synchronous motor.

  2. Description of the Related Art Conventionally, a torque control technique for reducing vibrations associated with fluctuations in rotational speed during one rotation of a one-cylinder compressor by controlling the input voltage or input current of an inverter / motor (see Patent Document 1).

  Further, as a drive source, a brushless DC motor that has a position detection mechanism in advance and is easier to control torque than an AC motor is exclusively used.

  In particular, a brushless DC motor having a surface magnet structure in which a permanent magnet of a rotor is mounted on the surface, when performing torque control, controls to reduce the d-axis current that does not contribute to motor torque generation, that is, the current phase is the motor speed electromotive voltage. A method for controlling the phase to be in phase with the current phase (current phase = 0) is known as a driving method that does not cause a decrease in efficiency, and is often used because of simple control.

  On the other hand, a brushless DC motor with an embedded magnet structure in which a permanent magnet of a rotor is embedded can output two generated torques, that is, a magnet torque and a reluctance torque at the same time, and optimize two torque distributions according to the load torque, Compared to a brushless DC motor with a surface magnet structure, it can be controlled to achieve maximum torque with the minimum current (hereinafter referred to as maximum torque control). The application to the air conditioner etc. which require is progressing.

A maximum torque control method for a brushless DC motor having an embedded magnet structure is shown in Non-Patent Document 1, and it is known that the dq axis current may be controlled based on a relational expression determined by an electric constant of the motor. It has been.
See Japanese Examined Patent Publication No. 6-42789 "Control method suitable for PM motor with embedded magnet structure", Morimoto et al., IEEJ Semiconductor Power Conversion Study Materials SPC-92-5

 However, when the maximum torque control method is combined with torque control, the following problems occur.

  (1) Model errors due to motor temperature and magnetic saturation occur, and the maximum torque condition is not constantly maintained. Then, problems associated with motor model errors (specifically, winding resistance with changes in temperature, change in speed electromotive force constant, change in inductance value of each dq axis due to magnetic saturation and change in speed electromotive force constant) In order to solve this, it is necessary to actually measure changes in various parameters due to the temperature and magnetic saturation and to add them to the calculation, but this is extremely difficult in practice.

  (2) In combination with torque control that cancels even higher-order components that are less affected by vibration, power consumption is more than necessary.

  (3) The peak current increases due to the torque control, exceeds the limit of the inverter current, and needs to be shifted from the operating point of the maximum torque control.

  The present invention has been made in view of the above-mentioned problems, and a synchronous motor driving method capable of realizing torque control for reducing low-speed vibration with a periodic pulsating load in a maximum efficiency state and a practical configuration. It is an object of the present invention to provide a compressor driving method and these apparatuses.

  The synchronous motor driving method according to claim 1 is a current waveform or a voltage waveform when a torque having a periodic torque fluctuation is controlled by a synchronous motor controlled by an inverter to suppress a speed fluctuation during one rotation. This is a method of superimposing the fluctuation amount on both the amplitude and the phase.

  The synchronous motor driving method according to claim 2 is a method of controlling the amount of phase fluctuation based on the amount of amplitude fluctuation controlled by the output of the torque control unit.

  A synchronous motor driving method according to a third aspect is a method of controlling the amplitude fluctuation amount based on the phase fluctuation amount controlled by the output of the torque control unit.

  A synchronous motor driving method according to a fourth aspect of the present invention is a method of controlling the amount of variation in amplitude based on the output of the torque control unit and controlling the amount of variation in phase based on the detected amount related to efficiency.

  The synchronous motor driving method according to claim 5 is a method of controlling the amount of phase fluctuation based on the output of the torque control unit and controlling the amount of amplitude fluctuation based on the detected amount related to the efficiency.

  A synchronous motor driving method according to a sixth aspect of the present invention employs a method corresponding to a fundamental wave and a lower order harmonic as the fluctuation amount.

  A synchronous motor driving method according to a seventh aspect of the present invention employs a method corresponding to a fundamental wave as the fluctuation amount.

  The synchronous motor driving method according to claim 8 is a method of superimposing a third-order harmonic on the amplitude fluctuation amount.

  The synchronous motor driving method according to claim 9 is the same as the first neutral point voltage obtained by the resistor having one end connected to the output terminal of each phase of the inverter and the other end connected to each other. This is a method of detecting the magnetic pole position of the rotor of the synchronous motor by integrating the difference with the second neutral point voltage obtained by connecting one end of the stator winding of each phase of the motor to each other.

  A compressor driving method according to a tenth aspect is a method of driving a one-cylinder compressor by a synchronous motor driven by the synchronous motor driving method according to any one of the first to ninth aspects.

  The synchronous motor drive device according to claim 11 is a current waveform or a voltage waveform when torque control for suppressing speed variation during one rotation is performed on a load having periodic torque variation by a synchronous motor controlled by an inverter. Inverter control means for controlling the inverter so as to superimpose the fluctuation amount on both the amplitude and the phase is included.

  The synchronous motor drive apparatus according to claim 12 employs, as the inverter control means, one that controls the phase fluctuation amount based on the amplitude fluctuation amount controlled by the output of the torque control unit.

  The synchronous motor drive apparatus according to claim 13 employs, as the inverter control means, one that controls the amount of variation in amplitude based on the amount of variation in phase controlled by the output of the torque control unit.

  The synchronous motor drive device according to claim 14 employs, as the inverter control means, one that controls the amount of amplitude fluctuation based on the output of the torque control unit and controls the amount of phase fluctuation based on the detected amount related to the efficiency. Is.

  The synchronous motor drive apparatus according to claim 15 employs, as the inverter control means, one that controls the amount of variation in phase based on the output of the torque control unit, and controls the amount of variation in amplitude based on the detected amount related to efficiency. Is.

  The synchronous motor drive device according to claim 16 employs, as the inverter control means, one that uses the fluctuation corresponding to the fundamental wave and the lower harmonic.

  The synchronous motor drive device according to claim 17 employs, as the inverter control means, one that uses the fluctuation corresponding to the fundamental wave.

  The synchronous motor drive apparatus according to claim 18 employs an inverter control means that superimposes a third harmonic on the amplitude variation.

  The synchronous motor drive device according to claim 19 is provided with a resistor having one end connected to an output terminal of each phase of the inverter and the other end connected to each other to obtain a first neutral point voltage; A stator winding of each phase of the synchronous motor, one end of which is connected to each other to obtain a neutral point voltage, and an integrating means for integrating the difference between the first neutral point voltage and the second neutral point voltage; And magnetic pole position detecting means for detecting the magnetic pole position of the rotor of the synchronous motor based on the integration signal.

  A compressor driving device according to a twentieth aspect drives the one-cylinder compressor by a synchronous motor driven by the synchronous motor driving device according to any one of the eleventh to nineteenth aspects.

  The invention according to claim 1 has a specific effect that it is possible to realize torque control for reducing low-speed vibration with a periodic pulsating load in a state of maximum efficiency and a practical configuration.

  The invention of claim 2 has the same effect as that of claim 1.

  The invention of claim 3 has the same effect as that of claim 1.

  The invention of claim 4 can realize control including iron loss which has not been considered in the maximum torque control method, and has the same effect as that of claim 1.

  The invention of claim 5 can realize control including iron loss which has not been taken into account in the maximum torque control method, and has the same effect as that of claim 1.

  The invention of claim 6 has the same effect as any one of claims 1 to 5.

  The invention of claim 7 has the same effect as any one of claims 1 to 5.

  The invention of claim 8 has the same effect as any of claims 1 to 5.

  The invention according to claim 9 has the same effects as any one of claims 1 to 8.

  The invention of claim 10 has a specific effect that energy saving and cost reduction can be realized.

  The invention according to claim 11 has a specific effect that the torque control for reducing the low-speed vibration can be realized with the periodic pulsating load in the maximum efficiency state and the practical configuration.

  The invention of claim 12 has the same effect as that of claim 11.

  The invention of claim 13 has the same effect as that of claim 11.

  The invention of claim 14 can achieve control including iron loss that has not been considered in the maximum torque control method, and has the same effect as that of claim 11.

  The invention of claim 15 can realize control including iron loss that has not been taken into account in the maximum torque control method, and has the same effect as that of claim 11.

  The invention of claim 16 has the same effects as any of claims 11 to 15.

  The invention according to claim 17 has the same effects as any of claims 11 to 15.

  The invention of claim 18 has the same effect as that of any of claims 11 to 15.

  The invention of claim 19 has the same effects as any of claims 11 to 18.

  The invention of claim 20 has a specific effect that energy saving and cost reduction can be realized.

  According to the synchronous motor driving method of claim 1, when performing torque control for suppressing speed fluctuation during one rotation by a synchronous motor controlled by an inverter, for a load having periodic torque fluctuation, Since the fluctuation amount is superimposed on both the amplitude and phase of the voltage waveform, the torque control to reduce the low-speed vibration is realized with the periodic pulsating load in the maximum efficiency state and the practical configuration. be able to. Further details will be described.

  If the idea of the maximum torque control method of the brushless DC motor having the embedded magnet structure described above is applied to a periodic load that pulsates during one rotation, as shown in FIG. It can be seen that the current phase may be changed during one rotation. That is, it can be seen that a complicated calculation based on the model of the maximum torque control method can be replaced with a simple waveform control amount. Therefore, the maximum torque control can be accurately realized by appropriately correcting the direct current component and the fluctuation component of each of the current amplitude and current phase shown in FIG. 1 for each operating condition. Of course, instead of changing the current amplitude and current phase in one rotation, the voltage amplitude and voltage phase may be changed in one rotation.

  According to the synchronous motor driving method of the second aspect, the phase variation amount is controlled based on the amplitude variation amount controlled by the output of the torque control unit, so that the same effect as in the first aspect is achieved. Can do.

  According to the synchronous motor driving method of the third aspect, the amplitude fluctuation amount is controlled based on the phase fluctuation amount controlled by the output of the torque control unit, and therefore the same effect as in the first aspect is achieved. Can do.

  According to the synchronous motor driving method of claim 4, the amplitude fluctuation amount is controlled based on the output of the torque control unit, and the phase fluctuation amount is controlled based on the detection amount related to the efficiency. In addition to realizing control including iron loss that has not been taken into account in the control method, the same action as in claim 1 can be achieved. In this regard, the motor current minimum is only the copper loss minimum only by the idea of the maximum torque control method. Also, “Energy-saving and high-efficiency operation method of brushless DC motor”, Morimoto et al., Electrology D, vol. 112-3, pp. Although the concept of maximum efficiency control is also shown in H.285 (Hei 4-3), since the iron loss is a constant, the same model error problem as the problem (1) of the maximum torque control described above exists as well. ing. Therefore, the control including the iron loss can be realized by adopting the method of claim 5.

  According to the synchronous motor drive method of claim 5, the phase fluctuation amount is controlled based on the output of the torque control unit, and the amplitude fluctuation amount is controlled based on the detection amount related to the efficiency. In addition to realizing control including iron loss that has not been taken into account in the control method, the same action as in claim 1 can be achieved.

  In the synchronous motor driving method according to claim 6, since the variation corresponding to the fundamental wave and the low-order harmonic is adopted, the same effect as any one of claims 1 to 5 is achieved. can do. Further details will be described.

  Of the harmonic component of the load torque waveform, a component having a large contribution to vibration {for example, for a one-cylinder compressor that becomes a pulsating load, a first harmonic + second harmonic: a flywheel effect for a higher-order torque fluctuation component ( Since the effect of the moment of inertia) increases with frequency, it can be easily limited to the fluctuation band of the current amplitude and the current phase only by the rotational fluctuation and the influence on the vibration due to this. Thereby, unnecessary power consumption can be eliminated, and more efficient operation is possible. Of course, a voltage may be used instead of the current.

  In the synchronous motor driving method according to the seventh aspect, since the variation corresponding to the fundamental wave is adopted as the variation, the same effect as any one of the first to fifth aspects can be achieved. In addition, unnecessary power consumption can be eliminated, and more efficient operation is possible.

  According to the synchronous motor driving method of claim 8, since the third harmonic is superimposed on the amplitude fluctuation amount, the peak current can be easily suppressed, and the limitation on the inverter current limit is relaxed. In addition to being able to operate at an optimum operating point in a wider load torque range, it is possible to achieve the same effect as any one of claims 1 to 5. Note that there is almost no influence on vibration due to the superposition of the third harmonic component due to the action of the moment of inertia.

  According to the synchronous motor driving method of claim 9, the first neutral point voltage obtained by a resistor having one end connected to the output terminal of each phase of the inverter and the other end connected to each other. The magnetic pole position of the rotor of the synchronous motor is detected by integrating the difference with the second neutral point voltage obtained by connecting one end of the stator winding of each phase of the synchronous motor to each other. Therefore, it is not necessary to provide a Hall element, an encoder, or the like for detecting the magnetic pole position, and the same operation as any one of claims 1 to 8 can be achieved.

  In the compressor driving method according to claim 10, since the one-cylinder compressor is driven by the synchronous motor driven by the synchronous motor driving method according to any one of claims 1 to 9, energy saving and Cost reduction can be realized.

  In the synchronous motor drive device according to claim 11, when performing torque control for suppressing speed fluctuation during one rotation by a synchronous motor controlled by an inverter, for a load having periodic torque fluctuation, inverter control means Thus, the inverter can be controlled to superimpose the fluctuation amount on both the amplitude and phase of the current waveform or voltage waveform. Therefore, torque control for reducing low-speed vibration can be realized with a periodic pulsating load in a maximum efficiency state and a practical configuration.

  In the synchronous motor drive device according to the twelfth aspect, the inverter control means employs a control device that controls the phase fluctuation amount based on the amplitude fluctuation amount controlled by the output of the torque control unit. 11 can be achieved.

  In the synchronous motor drive device according to the thirteenth aspect, the inverter control means employs a device that controls the amount of variation in amplitude based on the amount of variation in phase controlled by the output of the torque control unit. 11 can be achieved.

  In the synchronous motor drive device according to claim 14, the inverter control means controls the amplitude fluctuation amount based on the output of the torque control unit, and controls the phase fluctuation amount based on the detected amount related to the efficiency. Since it employs, it is possible to realize control including iron loss that has not been taken into account in the maximum torque control method, and it is possible to achieve the same operation as in the eleventh aspect.

  In the synchronous motor drive device according to claim 15, the inverter control means controls the amount of phase variation based on the output of the torque control unit, and controls the amount of amplitude variation based on the detected amount related to efficiency. Since it employs, it is possible to realize control including iron loss that has not been taken into account in the maximum torque control method, and it is possible to achieve the same operation as in the eleventh aspect.

  In the synchronous motor drive device according to claim 16, the inverter control means adopts the variation corresponding to the fundamental wave and the low-order harmonics. The same action as any one can be achieved. In addition, unnecessary power consumption can be eliminated, and more efficient operation is possible.

  In the synchronous motor drive device according to claim 17, the inverter control means employs the one corresponding to the fundamental wave as the amount of variation, and thus the same effect as any one of claims 11 to 15. Can be achieved. In addition, unnecessary power consumption can be eliminated, and more efficient operation is possible.

  In the synchronous motor drive device according to claim 18, since the inverter control means employs a component that superimposes the third harmonic on the fluctuation amount of the amplitude, the peak current can be easily suppressed. In addition, the limitation on the inverter current limit can be relaxed, operation at an optimum operating point can be performed in a wider load torque range, and the same operation as any one of claims 11 to 15 can be achieved. Note that there is almost no influence on vibration due to the superposition of the third harmonic component due to the action of the moment of inertia.

  In the synchronous motor drive device according to claim 19, a first neutral point voltage is obtained by a resistor having one end connected to an output terminal of each phase of the inverter and the other end connected to each other, The second neutral point voltage is obtained by the stator windings of the respective phases of the synchronous motor, the ends of which are connected to each other, and the difference between the first neutral point voltage and the second neutral point voltage is integrated by the integrating means Then, the magnetic pole position detection means can detect the magnetic pole position of the rotor of the synchronous motor based on the integration signal. Therefore, it is not necessary to provide a Hall element, an encoder, or the like for magnetic pole position detection, and the same effect as any one of claims 11 to 18 can be achieved.

  In the compressor driving device according to claim 20, since the one-cylinder compressor is driven by the synchronous motor driven by the synchronous motor driving device according to any one of claims 11 to 19, energy saving and Cost reduction can be realized.

  Further details will be described.

  When the three-phase / dq coordinate transformation of the synchronous motor is set as shown in Equation 1, the voltage equation of the synchronous motor becomes as shown in Equation 2, and the generated torque is given as shown in Equation 3 by the dq axis current. Here, the d-axis is an axis indicating the direction of the magnetic flux generated by the permanent magnet, and the q-axis is an axis that is electrically shifted by 90 ° from the d-axis.

At this time, the actual current flowing through the synchronous motor becomes Equation 5 by the conversion of Equation 1, and the synchronous motor applied voltage can be calculated by Equations 4 and 5.

  Here, p is the number of pole pairs, R is a winding resistance, Lq and Ld are self-inductances converted into a dq coordinate system, and Ke is a speed electromotive force constant. Θ is an electrical angle.

  Here, for a synchronous motor having a surface magnet structure, since Lq = Ld, it can be seen from Equation 3 that the d-axis current is not involved in the torque. Therefore, in order to perform the torque control with the minimum motor current, that is, high efficiency, the d-axis current may be controlled to 0. When this condition is applied in Equation 6, the desired current phase is 0 (fixed). I understand. However, although the current phase at the time of maximum torque control is fixed, it can be seen that the voltage phase needs to be changed as shown in FIG.

Here, FIG. 2 shows a simulation result of fluctuations in voltage amplitude and phase under the condition that the current phase is set to 0 rad for efficient torque control in the case of a brushless DC motor having a surface magnet structure. In addition, it can be seen that since the inductance is small, the amount of phase fluctuation is smaller than that of a brushless DC motor having an embedded magnet structure.

  On the other hand, according to the above-mentioned document “Control Method Suitable for Embedded Magnet Structure PM Motor”, the maximum torque (minimizing the motor current) condition of the synchronous motor of the embedded magnet structure of the synchronous motor is the dq axis of Equation 7 Given in current. The generated torque at this time is represented by equation 8 from equations 3 and 7.

That is, it is understood that when performing torque control to minimize the current, it is necessary to appropriately adjust the distribution of the dq axis current according to the magnitude of the torque.

  Further, for a reluctance motor (a motor driven only by reluctance torque) which is a kind of synchronous motor, the current phase = 45 ° can be understood as the maximum torque control condition by setting the speed electromotive force constant Ke in Equation 7 to 0. In general, reluctance motors are designed to increase inductance L in order to obtain reluctance torque. Therefore, it is necessary to set a large voltage phase fluctuation, and the voltage amplitude and phase are both changed to compensate for the torque control. It has the effect of improving efficiency as a brushless DC motor with a built-in magnet structure.

  Here, when considering a load in which the load torque pulsates during one rotation, for example, the case where the compressor is driven by a synchronous motor, based on Equations 7 and 8, the dq-axis current is expressed as shown in FIG. It can be seen that it is necessary to vary the rotation position (magnetic pole position).

  Further, when the obtained dq-axis current of FIG. 1 is converted into the amplitude and phase of the actual current by Equation 6, as shown in FIG. 3, in order to generate the motor torque that matches the pulsating load when the motor current is minimal. It is understood that both the current amplitude and the current phase should be changed. Based on this knowledge, it is possible to achieve torque control with minimal motor current by simply adjusting the magnitude and phase of the fluctuations in the current amplitude and current phase, and use many model constants. Complicated calculations are no longer necessary, and many model constant measurement steps are not required under each condition in order to take into account the effects of temperature rise and magnetic saturation.

  Further, at this time, the required applied voltage is obtained from Equations 2 and 5, as shown in FIG. 4. Like the motor current, the amplitude and phase of the applied voltage are varied in synchronization with the pulsation of the load torque. You can see that

  In addition, the broken line has shown the average value in FIG. 3, FIG.

  Hereinafter, embodiments of the present invention will be described in detail with reference to the accompanying drawings.

  FIG. 5 is a block diagram showing an embodiment of the synchronous motor driving apparatus of the present invention.

  This synchronous motor driving device performs a predetermined calculation (for example, PI calculation) by using the calculated deviation as an input and a speed deviation calculation unit 1 that calculates a deviation between the speed command ω * and the motor speed ω. A speed control unit 2 that outputs an average value command, and a current that outputs a current amplitude command by adding an average value command of current amplitude and a variation in current amplitude for each rotational position output from a torque control unit 10 described later. A three-phase alternating current calculation unit that outputs a three-phase alternating current command based on, for example, Equation 9 using the amplitude command output unit 3 and the current amplitude command and a current phase command output from a current phase command output unit 11 described later as inputs. 4, the current source inverter 5 that receives this three-phase AC command, the synchronous motor 6 to which the output of the current source inverter 5 is supplied, and the magnetic pole position of the rotor of the synchronous motor 6 to detect the position angle θ. Output rotor position The output unit 7, the speed calculation unit 9 that calculates and outputs the motor speed ω with the position angle θ as an input, and the torque control calculation with the motor speed ω and the position angle θ as an input to change the current amplitude for each rotational position And a phase control calculation (for example, multiplying by a predetermined coefficient and performing a phase shift calculation) by inputting the fluctuation amount of the current amplitude at each rotational position as an input, and a fluctuation amount of the current phase. A phase control unit 8 that calculates and outputs a command, and a current phase command output unit 11 that calculates and outputs a current phase command by adding an average phase command β * obtained in a conventionally known manner and a variation command. And have.

  Examples of the rotor position detection unit 7 include a rotation position sensor such as an encoder and an output counter circuit, a position detection circuit that filters the motor terminal voltage, and a circuit that calculates a position based on the electrical quantity of the motor.

  In this embodiment and the following embodiments, examples of the synchronous motor 6 include a brushless DC motor having a surface magnet structure, a brushless DC motor having an embedded magnet structure, and a reluctance motor.

  FIG. 6 is a flowchart for explaining the operation of the synchronous motor driving device of FIG.

  In step SP1, the rotor position (position angle) θ is input. In step SP2, the rotational speed (motor speed) ω is calculated from the rotor position θ. In step SP3, the actual speed ω and the speed command ω * are calculated. An average current amplitude command is obtained by PI (proportional / integral) calculation of the difference, and in step SP4, torque control calculation is performed with the actual speed ω and the rotor position θ as inputs, and the current amplitude is calculated from the actual speed fluctuation amount. In step SP5, the average current amplitude command and the current amplitude variation are added to obtain and store the amplitude command, and in step SP6, the current amplitude variation is multiplied by a factor and shifted. A phase fluctuation is obtained (where the coefficient and shift amount are determined empirically, for example), and in step SP7, the average phase command β * from the outside and the current phase fluctuation are added. In step SP8, the stored current amplitude and phase commands are supplied to the three-phase alternating current calculation unit. In step SP9, each phase current is obtained and supplied to the current source inverter. Return to the process.

  Therefore, vibration is reduced by adding the average current amplitude command and the current amplitude fluctuation to obtain the amplitude command, and the phase is obtained by adding the average phase command β * from the outside and the current phase fluctuation. By obtaining a command, an improvement in efficiency can be achieved. As a result, the periodic pulsating load can be torque controlled in the maximum efficiency state, and vibration can be reduced.

  FIG. 7 is a block diagram showing another embodiment of the synchronous motor driving apparatus of the present invention.

  In this synchronous motor driving device, instead of the current source inverter 5, a current deviation calculation unit 5d that calculates a deviation between a three-phase AC command and a winding current detection value output from a winding current detection unit 5c described later; A current control unit 5a that performs current control using the calculated deviation as an input and converts the current command into a voltage command, a voltage source inverter 5b that receives the converted voltage command, and a winding current of the synchronous motor 6 described later 5 is different from the synchronous motor driving device of FIG. 5 only in that the winding current detecting unit 5c for detecting the above is used, and the other components are the same as those of the synchronous motor driving device of FIG.

  FIG. 8 is a flowchart for explaining the operation of the synchronous motor driving apparatus of FIG.

  In step SP1, the rotor position (position angle) θ is input. In step SP2, the rotational speed (motor speed) ω is calculated from the rotor position θ. In step SP3, the actual speed ω and the speed command ω * are calculated. An average current amplitude command is obtained by PI (proportional / integral) calculation of the difference, and in step SP4, torque control calculation is performed with the actual speed ω and the rotor position θ as inputs, and the current amplitude is calculated from the actual speed fluctuation amount. In step SP5, the average current amplitude command and the current amplitude variation are added to obtain and store the amplitude command, and in step SP6, the current amplitude variation is multiplied by a factor and shifted. A phase fluctuation is obtained (where the coefficient and shift amount are determined empirically, for example), and in step SP7, the average phase command β * from the outside and the current phase fluctuation are added. In step SP8, the stored current amplitude and phase commands are supplied to the three-phase alternating current calculation unit. In step SP9, the respective phase currents are obtained and supplied to the current control type inverter. Return to the original process.

  Therefore, vibration is reduced by adding the average current amplitude command and the current amplitude fluctuation to obtain the amplitude command, and the phase is obtained by adding the average phase command β * from the outside and the current phase fluctuation. By obtaining a command, an improvement in efficiency can be achieved. As a result, the periodic pulsating load can be torque controlled in the maximum efficiency state, and vibration can be reduced.

  Further, the synchronous motor driving device of FIG. 7 employs a simple voltage source inverter as compared with the current source inverter in the main circuit configuration, and therefore the configuration can be simplified as a whole.

  FIG. 9 is a block diagram showing still another embodiment of the synchronous motor driving apparatus of the present invention.

  This synchronous motor driving device performs a predetermined calculation (for example, PI calculation) with the calculated deviation as an input and a speed deviation calculation unit 21 that calculates a deviation between the speed command ω * and the motor speed ω. Speed control unit 22 that outputs an average value command, and a voltage that outputs a voltage amplitude command by adding an average value command of voltage amplitude and a variation in voltage amplitude for each rotational position output from torque control unit 30 described later A three-phase AC calculation unit that outputs a three-phase AC command based on, for example, Equation 10 using the amplitude command output unit 3 and the voltage amplitude command and a voltage phase command output from a voltage phase command output unit 31 described later as inputs. 24, the voltage source inverter 25 that receives this three-phase AC command, the synchronous motor 6 to which the output of the voltage source inverter 25 is supplied, and the magnetic pole position of the rotor of the synchronous motor 6 to detect the position angle θ. Output The rotor position detection unit 27, the speed calculation unit 29 that calculates and outputs the motor speed ω with the position angle θ as an input, the torque control calculation with the motor speed ω and the position angle θ as input, and the voltage for each rotation position A torque control unit 30 that outputs amplitude fluctuations and a phase control calculation (for example, multiplication of a predetermined coefficient and phase shift calculation) with the voltage amplitude fluctuation for each rotational position as an input for voltage phase The phase control unit 28 that calculates and outputs the fluctuation command of the voltage, and the voltage phase command that calculates and outputs the voltage phase command by adding the average phase command α * and the fluctuation command obtained in a conventional manner. And an output unit 31.

  Examples of the rotor position detection unit 27 include a rotation position sensor such as an encoder and an output counter circuit, a position detection circuit that filters the motor terminal voltage, and a circuit that calculates a position based on the electrical quantity of the motor.

  FIG. 10 is a flowchart for explaining the operation of the synchronous motor driving apparatus of FIG.

  In step SP1, the rotor position (position angle) θ is input. In step SP2, the rotational speed (motor speed) ω is calculated from the rotor position θ. In step SP3, the actual speed ω and the speed command ω * are calculated. An average voltage amplitude command is obtained by PI (proportional / integral) calculation of the difference, and in step SP4, torque control calculation is performed with the actual speed ω and the rotor position θ as inputs, and the voltage amplitude is calculated from the actual speed fluctuation amount. In step SP5, the average voltage amplitude command and the voltage amplitude variation are added to obtain and store the amplitude command, and in step SP6, the voltage amplitude variation is multiplied by a factor and shifted. A phase variation is obtained (where the coefficient and shift amount are determined empirically, for example), and in step SP7, the external average phase command α * and the voltage phase variation are added. In step SP8, the stored voltage amplitude and phase commands are supplied to the three-phase alternating current calculation unit. In step SP9, the phase voltages are obtained and supplied to the voltage source inverter. Return to the process.

  Therefore, vibration is reduced by adding the average voltage amplitude command and the voltage amplitude variation to obtain the amplitude command, and the phase is obtained by adding the average phase command β * from the outside and the voltage phase variation. By obtaining a command, an improvement in efficiency can be achieved. As a result, the periodic pulsating load can be torque controlled in the maximum efficiency state, and vibration can be reduced.

  FIG. 11 is a block diagram showing still another embodiment of the synchronous motor driving device of the present invention.

  This synchronous motor drive device performs a predetermined calculation (for example, PI calculation) with the calculated deviation as an input, and a speed deviation calculation unit 41 that calculates a deviation between the speed command ω * and the motor speed ω, and the voltage phase is calculated. Speed control unit 42 that outputs an average value command, and a voltage that outputs a voltage phase command by adding a voltage phase average value command and a variation in voltage phase for each rotational position output from torque control unit 50 described later A three-phase alternating current calculation unit that outputs a three-phase alternating current command based on, for example, Formula 10, using the phase command output unit 43 and a voltage amplitude command output from the voltage amplitude command output unit 51 described later as an input. 44, the voltage source inverter 45 that receives this three-phase AC command, the synchronous motor 6 to which the output of the voltage source inverter 45 is supplied, and the magnetic pole position of the rotor of the synchronous motor 6 to detect the position angle θ. output Rotor position detection unit 47, a speed calculation unit 49 that calculates and outputs a motor speed ω with the position angle θ as an input, and performs a torque control calculation with the motor speed ω and the position angle θ as inputs, for each rotational position. The torque control unit 50 that outputs the voltage phase fluctuation amount, and the voltage phase fluctuation calculation for each rotational position as an input to perform phase control calculation (for example, multiply by a predetermined coefficient and perform phase shift calculation) A voltage controller that calculates and outputs a voltage amplitude command by adding a phase control unit 48 that calculates and outputs an amplitude variation command, and an average amplitude command Vm * and a variation command that are obtained in a conventional manner. And a command output unit 51.

  Examples of the rotor position detection unit 47 include a rotation position sensor such as an encoder and an output counter circuit, a position detection circuit that filters the motor terminal voltage, and a circuit that calculates a position based on the electrical quantity of the motor.

  FIG. 12 is a flowchart for explaining the operation of the synchronous motor driving apparatus of FIG.

  In step SP1, the rotor position (position angle) θ is input. In step SP2, the rotational speed (motor speed) ω is calculated from the rotor position θ. In step SP3, the actual speed ω and the speed command ω * are calculated. An average voltage phase command is obtained by PI (proportional / integral) calculation of the difference, and in step SP4, torque control calculation is performed with the actual speed ω and the rotor position θ as inputs, and the voltage phase is calculated from the actual speed fluctuation amount. In step SP5, the average voltage phase command and the voltage phase variation are added to obtain and store the phase command, and in step SP6, the voltage phase variation is multiplied by a factor and shifted. A variation in amplitude is obtained (where the coefficient and shift amount are determined empirically, for example), and in step SP7, the average amplitude command Vm * from the outside and the variation in voltage amplitude are obtained. In step SP8, the stored voltage amplitude and phase commands are supplied to the three-phase AC calculation unit, and in step SP9, each phase voltage is obtained and supplied to the voltage source inverter. Return to the original process.

  Therefore, the vibration is reduced by adding the average voltage phase command and the voltage phase variation to obtain the phase command, and the amplitude is obtained by adding the average amplitude command Vm * and the voltage amplitude variation from the outside. By obtaining a command, an improvement in efficiency can be achieved. As a result, the periodic pulsating load can be torque controlled in the maximum efficiency state, and vibration can be reduced.

  FIG. 13 is a block diagram showing still another embodiment of the synchronous motor driving apparatus of the present invention.

  This synchronous motor drive device performs a predetermined calculation (for example, PI calculation) with the calculated deviation as an input, and a speed deviation calculation unit 61 that calculates a deviation between the speed command ω * and the motor speed ω. Speed control unit 62 that outputs an average value command, and a voltage that outputs a voltage amplitude command by adding an average value command of voltage amplitude and a variation in voltage amplitude for each rotational position output from torque control unit 70 described later A three-phase alternating current calculation unit that outputs a three-phase alternating current command based on, for example, Equation 10 using the amplitude command output unit 63 and the voltage amplitude command and a voltage phase command output from the voltage phase command output unit 71 described later as inputs. 64, the voltage source inverter 65 that receives this three-phase AC command, the synchronous motor 6 to which the output of the voltage source inverter 65 is supplied, and the magnetic pole position of the rotor of the synchronous motor 6 to detect the position angle θ. output Rotor position detection unit 67, a speed calculation unit 69 that calculates and outputs a motor speed ω with the position angle θ as an input, a torque control calculation with the motor speed ω and the position angle θ as inputs, A torque control unit 70 that outputs a voltage amplitude variation and an inverter input current detected by an inverter input current detection unit 72 (to be described later) are input to perform phase control calculation to calculate and output a voltage phase variation command. The phase control unit 68, the voltage phase command output unit 71 that calculates and outputs the voltage phase command by adding the average phase command α * and the variation command obtained as conventionally known, and the voltage from the commercial power source 73 And an inverter input current detector 72 that detects an inverter input current (a kind of detection amount related to efficiency) supplied to the inverter 65.

    Examples of the rotor position detection unit 67 include a rotation position sensor such as an encoder and an output counter circuit, a position detection circuit that filters the motor terminal voltage, and a circuit that calculates a position based on the electrical quantity of the motor.

  Further, the current control can be performed by adding a winding current detection unit and a current control unit, and a current source inverter can be employed instead of the voltage source inverter.

  FIG. 14 is a flowchart for explaining the operation of the synchronous motor driving apparatus of FIG.

  In step SP1, the rotor position (position angle) θ is input. In step SP2, the rotational speed (motor speed) ω is calculated from the rotor position θ. In step SP3, the actual speed ω and the speed command ω * are calculated. An average voltage amplitude command is obtained by PI (proportional / integral) calculation of the difference, and in step SP4, torque control calculation is performed with the actual speed ω and the rotor position θ as inputs, and the voltage amplitude is calculated from the actual speed fluctuation amount. In step SP5, the amplitude command is obtained by adding the average voltage amplitude command and the voltage amplitude variation, and stored. In step SP6, the inverter input current is determined according to the magnitude of the inverter input current. In order to control and calculate the voltage phase fluctuation, and to add the average phase command α * from the outside and the voltage phase fluctuation in step SP7, In step SP8, the stored voltage amplitude and phase commands are supplied to the three-phase alternating current calculation unit. In step SP9, each phase voltage is obtained, supplied to the voltage source inverter, and the process returns to the original process.

  Therefore, vibration is reduced by adding the average voltage amplitude command and the fluctuation of the voltage amplitude to obtain the amplitude command, and the voltage is set according to the magnitude of the inverter input current (to minimize the inverter input current). By calculating the amount of phase fluctuation and adding the average amplitude command Vm * from the outside to obtain the phase command, it is possible to achieve an improvement in efficiency by performing control in consideration of iron loss. As a result, the periodic pulsating load can be torque controlled in the maximum efficiency state, and vibration can be reduced.

  In addition, instead of calculating and calculating the voltage phase fluctuation according to the magnitude of the inverter input current, the inverter input power is calculated by adding voltage detection, and control is performed so that the inverter input power is minimized. Good.

  FIG. 15 is a block diagram showing still another embodiment of the synchronous motor driving apparatus of the present invention.

  This synchronous motor driving device performs a predetermined calculation (for example, PI calculation) using the calculated deviation as an input, and a speed deviation calculation unit 81 that calculates a deviation between the speed command ω * and the motor speed ω, and thereby calculates the voltage phase. Speed control unit 82 that outputs an average value command, and a voltage that outputs a voltage phase command by adding a voltage phase average value command and a variation in voltage phase for each rotational position output from torque control unit 90 described later. A three-phase AC calculation unit that outputs a three-phase AC command based on, for example, Formula 10, using the phase command output unit 83 and this voltage phase command and a voltage amplitude command output from a voltage amplitude command output unit 91 described later as inputs. 84, the voltage source inverter 85 that receives this three-phase AC command, the synchronous motor 6 to which the output of the voltage source inverter 85 is supplied, and the magnetic pole position of the rotor of the synchronous motor 6 to detect the position angle θ. output A rotor position detector 87, a speed calculator 89 that calculates and outputs a motor speed ω with the position angle θ as an input, a torque control calculation with the motor speed ω and the position angle θ as inputs, and A torque control unit 90 that outputs a voltage phase variation, and an inverter input current detected by an inverter input current detection unit 92 (to be described later) are input to perform phase control calculation to calculate and output a voltage amplitude variation command. The phase control unit 88, the voltage amplitude command output unit 91 that calculates and outputs the voltage amplitude command by adding the average amplitude command Vm * and the variation command obtained as conventionally known, and the voltage from the commercial power supply 93 And an inverter input current detection unit 92 that detects an inverter input current (a kind of detection amount related to efficiency) supplied to the inverter 85.

    Examples of the rotor position detection unit 87 include a rotation position sensor such as an encoder and a counter circuit for the output thereof, a position detection circuit for filtering the motor terminal voltage, and a circuit for calculating a position from the electric quantity of the motor.

  Further, the current control can be performed by adding a winding current detection unit and a current control unit, and a current source inverter can be employed instead of the voltage source inverter. In addition, instead of calculating and calculating the voltage phase variation according to the magnitude of the inverter input current, the inverter input power is calculated by adding voltage detection, and control is performed so that the inverter input power is minimized. Good.

  FIG. 16 is a diagram showing fluctuations in load torque corresponding to the rotation angle of the one-cylinder compressor, and FIG. 17 is a diagram showing frequency distribution of the load torque.

  As is clear from these figures, the actual pulsation load includes many frequency components, and when this torque control is performed to completely compensate for this and suppress the speed fluctuation that causes vibration, the motor current There is a problem that the effective value and the peak value of the increase.

  Here, since the speed fluctuation of the high-frequency torque pulsation component is reduced by the flywheel effect due to the inertia moment of the synchronous motor and the load, it hardly contributes to the vibration. Therefore, by setting the frequency of the pulsating torque to be compensated by torque control to only the fundamental wave and the lower harmonics, unnecessary current for dealing with torque pulsation that hardly contributes to vibration can be eliminated. A more efficient synchronous motor control can be realized by combining the motor driving device and the synchronous motor driving method. Specifically, for example, it is possible to easily cope with the problem by simply adding a filtering function to the torque control unit included in the synchronous motor driving device.

  In addition, when incorporating a compressor into an air conditioner, it is possible to absorb vibrations transmitted to the outdoor unit housing by devising a piping shape that connects the heat exchanger and the compressor, and supporting with a rubber foot. Even if the frequency of the pulsating torque to be compensated by torque control is limited to only the fundamental wave, there is no practical problem, and more efficient control can be performed. Of course, the same applies to devices other than air conditioners, such as refrigerators.

  FIG. 18 is a block diagram showing still another embodiment of the synchronous motor driving apparatus of the present invention.

  The synchronous motor driving device performs a predetermined calculation (for example, PI calculation) with the calculated deviation as an input, and a speed deviation calculation unit 101 that calculates a deviation between the speed command ω * and the motor speed ω, thereby calculating a voltage amplitude. The speed control unit 102 that outputs an average value command, and the voltage amplitude command by adding the voltage amplitude average value command and the voltage amplitude variation for each rotational position output from the voltage amplitude variation output unit 113 described later. The three-phase AC command output unit 103 outputs a three-phase AC command based on, for example, Equation 10 using the output voltage amplitude command output unit 103 and the voltage amplitude command and a voltage phase command output from the voltage phase command output unit 111 described later as inputs. AC calculation unit 104, current source inverter 105 that receives this three-phase AC command, synchronous motor 6 to which the output of current source inverter 105 is supplied, and the magnetic pole position of the rotor of synchronous motor 6 Rotor position detector 107 that detects the position angle θ and outputs the position angle θ, the speed calculation section 109 that calculates and outputs the motor speed ω with the position angle θ as an input, and the torque with the motor speed ω and the position angle θ as an input A torque control unit 110 that performs a control calculation and outputs a variation in voltage amplitude for each rotational position, and a phase control calculation using the voltage amplitude variation for each rotational position output from the voltage amplitude variation output unit 113 as input. For example, the phase control unit 108 that calculates and outputs a voltage phase variation command by multiplying a predetermined coefficient and performing a phase shift calculation), and an average phase command α * obtained in a conventionally known manner Current phase command output unit 111 that calculates and outputs a voltage phase command by adding the command and the variation command, a third harmonic generation unit 112 that generates a third harmonic with the position angle θ as an input, torque control Part 110 And a voltage amplitude variation output unit 113 for outputting a variation of the voltage amplitude for each rotation position by adding the voltage change and the third-order harmonics of the amplitude of the al output.

  Examples of the rotor position detection unit 107 include a rotation position sensor such as an encoder and a counter circuit for the output thereof, a position detection circuit for filtering the motor terminal voltage, and a circuit for calculating a position from the electric quantity of the motor.

  FIG. 19 is a flowchart for explaining the operation of the synchronous motor driving device of FIG.

  In step SP1, the rotor position (position angle) θ is input. In step SP2, the rotational speed (motor speed) ω is calculated from the rotor position θ. In step SP3, the actual speed ω and the speed command ω * are calculated. An average voltage amplitude command is obtained by PI (proportional / integral) calculation of the difference, and in step SP4, torque control calculation is performed with the actual speed ω and the rotor position θ as inputs, and the voltage amplitude is calculated from the actual speed fluctuation amount. In step SP5, the third harmonic component of the phase for reducing the peak is added to the voltage amplitude variation to calculate a new voltage amplitude variation. In step SP6, the average voltage amplitude command and The amplitude command is obtained by adding the new voltage amplitude variation and stored, and in step SP7, the new voltage amplitude variation is multiplied and shifted to obtain the voltage phase variation. Here, the coefficient and the shift amount are determined empirically, for example.) In step SP8, the phase command is obtained by adding the average phase command α * from the outside and the voltage phase variation, and is stored. In SP9, the stored voltage amplitude and phase commands are supplied to the three-phase AC calculation unit, and in step SP10, each phase current is obtained, supplied to the current source inverter, and the process returns to the original process.

  Therefore, vibration is reduced by adding the average voltage amplitude command and the voltage amplitude variation to obtain the amplitude command, and the phase is obtained by adding the average phase command β * from the outside and the voltage phase variation. By obtaining a command, an improvement in efficiency can be achieved. As a result, the periodic pulsating load can be torque controlled in the maximum efficiency state, and vibration can be reduced.

  Further, it is possible to perform current control by adding a winding current detection unit and a current control unit. 18 and 19, the third harmonic component is superimposed on the voltage command, but the third harmonic component may be superimposed on the current command. In any case, how much the third harmonic component should be superimposed is determined by the specification of the synchronous motor drive system.

  FIG. 20 shows a torque waveform when the third harmonic of the current waveform is superimposed so that the third harmonic component of about 10% of the fundamental wave of the torque is superimposed on the torque waveform {see (A) in FIG. 20], FIG. 21 is a diagram showing a current amplitude waveform {see FIG. 20B) and a current phase waveform {see FIG. 20C}. In any of the figures, “a” indicates a waveform after the third harmonic is superimposed, “b” indicates a waveform before the third harmonic is superimposed, and “c” indicates an average value.

  As is apparent from the figure, the peak of the current amplitude (the peak of the motor current) can be suppressed by superimposing the third harmonic, thereby shifting the operating point caused by the current capacity limitation of the inverter. There is no need to operate, and the control of the synchronous motor driving device and the synchronous motor driving method of FIGS. 5 to 15 can be realized in a wider range.

  In the case of performing the control for superimposing the third harmonic in this way, the third harmonic may not be included in the load torque, or the speed fluctuation may be increased when different, but compared with the fundamental wave. Considering the fact that the flywheel has a large effect due to its high frequency, and that anti-vibration measures are taken at the time of incorporation, there is no practical problem. It is preferable to apply to a synchronous motor driving method.

  FIG. 21 is a block diagram showing still another embodiment of the synchronous motor driving apparatus of the present invention.

  In this synchronous motor drive device, a voltage source inverter 121 is configured by connecting three series connection circuits of two switching transistors in parallel between both terminals of a DC power source 120. A protective diode is connected in parallel with each switching transistor. Then, three Y-connected resistors 122u, 122v, 122w are connected to the midpoint of each series connection circuit, and Y-connected stator windings 6u, 6v, 6w of the synchronous motor 6 are connected. ing. In addition, 6a has shown the rotor. Then, the first neutral point voltage VN obtained at the midpoint of the resistors 122u, 122v, 122w is supplied to the non-inverting input terminal of the operational amplifier 123a, and the second obtained at the midpoint of the stator windings 6u, 6v, 6w. The neutral point voltage VM is supplied to the inverting input terminal of the operational amplifier 123a through the resistor 123b. A resistor 123c is connected between the inverting input terminal and the output terminal of the operational amplifier 123a. Therefore, a difference voltage VNM corresponding to the difference between the first neutral point voltage VN and the second neutral point voltage VM is obtained at the output terminal of the operational amplifier 123a. This differential voltage VNM is supplied to an integration circuit formed by connecting a resistor 124a and a capacitor 124b in series, and an integration signal ∫VNOdt obtained at the midpoint between the resistor 124a and the capacitor 124b is supplied to the non-inverting input terminal of the operational amplifier 125. By connecting the inverting input terminal of the operational amplifier 125 to the ground, a zero cross comparator is configured, and an output signal from the zero cross comparator is supplied to the microprocessor 126 as a position signal (magnetic pole position detection signal). Further, the integration signal ∫VNOdt is supplied to the integration signal level detection circuit 127 and the detection signal from the integration signal level detection circuit 127 is supplied to the microprocessor 126. The microprocessor 126 is also supplied with a speed command and a speed fluctuation command, and outputs a signal for controlling the inverter 121 via the base drive circuit 128.

  FIG. 22 is a block diagram showing the configuration of the microprocessor 126.

  The microprocessor 126 receives the position signal from the interrupt processing 1 to stop, reset, and restart the period measurement timer 131, and inputs the timer value when the period measurement timer 131 is stopped. The position signal cycle calculation unit 132 to be calculated, the speed calculation unit 133 which calculates the speed by inputting the cycle of the position signal output from the position signal cycle calculation unit 132, calculates and outputs the current speed, and is given from the outside A deviation calculator 134 that calculates a difference between a speed command to be output and a current speed output from the speed calculator 133 and outputs it as a speed fluctuation; a speed fluctuation output from the deviation calculator 134; and a speed fluctuation given from the outside A switching signal calculation unit 135 that calculates and outputs a switching signal with a command as an input, and a deviation calculation unit 13 Is output from the deviation calculating unit 134 and the primary component compensation model calculating unit 136 that calculates and outputs the primary component compensation model with the speed fluctuation output from the switching signal and the switching signal output from the switching signal calculating unit 135 as inputs. The PI calculation unit 137 that performs the PI calculation with the speed variation as an input and outputs the calculation result, the primary component compensation model output from the primary component compensation model calculation unit 136, and the calculation result output from the PI calculation unit 137 An adder 138 that adds and outputs as a voltage command and a position signal cycle output from the position signal cycle calculation unit 132 and a phase amount command provided from an adder 147 described later are input to calculate and output a timer value. The timer value calculator 139 and the timer value output from the timer value calculator 139 are set and the position signal is received. The phase correction timer 140 that outputs the count over signal by counting the timer value that is started and started by the interrupt processing 1 and the timer value that is output from the timer value calculation unit 139 are set, and the phase correction timer The energization width control timer 141 that outputs the count over signal when the set timer value is counted by the interrupt processing 2 based on the count over signal output from the 140 and the count output from the phase correction timer 140 An inverter mode selection unit 142 that reads and outputs a voltage pattern from the memory 143 by an interrupt process 2 by an over signal or an interrupt process 3 by a count over signal output from the energization width control timer 141, and a voltage command output from an adder 138 And In addition, the voltage pattern output from the inverter mode selection unit 142 is input to perform pulse width modulation, and the PWM unit 144 that outputs a switch signal and the primary component compensation model output from the primary component compensation model calculation unit 136 are shifted. From the delay processing unit 145, the coefficient unit 146 for multiplying the shifted first-order component compensation model output from the delay processing unit 145 by a predetermined coefficient and outputting a compensation phase amount command, and the coefficient unit 146 An adder 147 that adds the output compensation phase amount command and the average phase amount command given from the outside to output the phase amount command is provided.

  The primary component compensation model calculation unit 136 performs compensation with a gain of 0 other than the primary component. Therefore, there is no problem even if the speed variation is used as the input of the primary component compensation model calculation unit 136. That is, the speed command is constant (direct current) in a steady state, and the primary component compensation model calculation unit 136 outputs the output even if direct current (or a signal having a frequency different from the output signal of the signal model) is input. It becomes zero. In other words, even if a speed fluctuation (= motor speed-speed command) is used as an input to the primary component compensation model calculation unit 136, the output of the primary component compensation model calculation unit 136 is determined only by the motor speed. Therefore, the control performance is not affected.

  FIG. 23 is a schematic diagram showing a control model of a system in which a compressor is driven by a synchronous motor driven by the synchronous motor driving device of FIG.

  In this control model, a subtraction unit 151 that calculates a difference between a speed command and the rotational speed of the synchronous motor 6 and a proportional control and an integral control (PI control) are performed by using the difference output from the subtraction unit 151 as an input. PI control unit 152 that outputs the result and the integration control result, and a speed fluctuation average value calculation unit that calculates the average magnitude Δω of speed fluctuations of N rotations (N is a natural number) with the difference output from the subtraction unit 151 as an input 153, an average speed fluctuation value Δω output from the speed fluctuation average value calculation unit 153 as an input, a switching unit 154 that outputs 0 or 1, and a rotation speed of the synchronous motor 6 and an output from the switching unit 154 And a multiplication unit 155 that outputs a multiplication result, and a variable structure primary component compensation that outputs a compensation value by performing primary component compensation with the multiplication result output from the multiplication unit 155 as an input. Unit 156, an addition unit 157 that adds a proportional control result, an integration control result, and a compensation value to output a voltage command, and an amplifier 157 ′ that receives the voltage command output from addition unit 157 and compensates for this, A subtractor 158 that calculates and outputs a difference between an output voltage output from the amplifier 157 ′ and a portion Eτ− of the motor speed electromotive force that is involved in the generation of current corresponding to torque, and a difference that is output from the subtractor 158. , The voltage / current transfer function (first-order lag element determined by the resistance and inductance of the motor winding) 159, the current output from the motor voltage / current transfer function 159, and the rotor A subtractor 160 that calculates and outputs a difference from a current iτ− that equivalently represents a torque error component associated with not directly controlling the current waveform (phase / amplitude) according to the position; The motor current / torque transfer function 161 that outputs the motor torque with the difference output from the unit 160 as an input, and the motor torque output from the motor current / torque transfer function 161 and the compressor load torque are subtracted and compressed. It has a subtractor 162 that outputs the axle torque, and a motor torque / speed transfer function 163 that receives the compressor shaft torque output from the subtractor 162 and outputs the speed. The subtraction unit 158, the voltage / current transfer function 159, the subtraction unit 160, the current / torque transfer function 161, the subtraction unit 162, and the torque / speed transfer function 163 constitute the synchronous motor 6.

  24 to 26 are flowcharts for explaining the processing of the microprocessor 126. FIG. 24 illustrates the interrupt process 1, FIG. 25 illustrates the interrupt process 2, and FIG. 26 illustrates the interrupt process 3.

  The process of the flowchart in FIG. 24 is performed every time a position signal is received.

  In step SP1, the value of the phase correction timer 140 is calculated from the phase amount command. In step SP2, the phase correction timer value is set in the phase correction timer 140. In step SP3, the phase correction timer 140 is started. In step SP4. The period measurement timer 131 is stopped, the value of the period measurement timer 131 is read in step SP5, the value of the period measurement timer 131 is reset in step SP6, and the period measurement timer 131 is started for the next period measurement. . In step SP7, the cycle of the position signal is calculated. In step SP8, the motor rotation speed is calculated from the calculation result of the position signal cycle. In step SP9, the speed fluctuation is calculated based on the motor rotation speed and the speed command. In step SP10, PI calculation is performed on the speed fluctuation, an average voltage amplitude command is calculated, and in step SP11, an average value of the magnitude of the speed fluctuation is calculated, and a switching signal is generated based on the obtained average value. In step SP12, the compensation voltage amplitude is calculated based on the speed fluctuation and the switching signal. In step SP13, the compensation voltage amplitude is added to the average voltage amplitude. In step SP14, delay processing is performed {eg, compensation Stores voltage amplitude and reads compensation voltage amplitude before M samples (M is a positive integer) } In step SP15, a compensation phase is calculated by multiplying a predetermined coefficient by the coefficient unit 146, and in step SP16, the compensation phase is added to the average phase command, and stored as the next phase command, and the original phase command is stored as it is. Return to processing.

  The process of the flowchart of FIG. 25 is performed every time a count over signal is output from the phase correction timer 140.

  In step SP1, the inverter mode is advanced by one step. In step SP2, a voltage pattern corresponding to the advanced inverter mode is output. In step SP3, the timer value of the conduction width control timer 141 is calculated from the conduction width command. In SP4, a timer value {= (timer value for 120 energization angle−120) deg} is set in the energization width control timer 141, and in step SP5, the energization width control timer 141 is started, and the process returns to the original process.

  The process of the flowchart of FIG. 26 is performed every time a count over signal is output from the energization width control timer 141.

  In step SP1, the inverter mode is advanced by one step. In step SP2, a voltage pattern corresponding to the advanced inverter mode is output, and the process returns to the original process.

  FIG. 27 is a diagram showing signal waveforms of respective parts of the synchronous motor driving device shown in FIGS. 21 and 22.

  When the compressor is driven by the synchronous motor 6, a differential voltage VNM is obtained as shown in FIG. 27A, and an integration signal ∫VNOdt is obtained as shown in FIG. A position signal is obtained as shown in (C) of FIG.

  By the interrupt process 1 based on this position signal, the phase correction timer 140 is started as shown in FIG. 27D (see the starting point of the arrow shown in FIG. 27D), and in FIG. 27 is output every time a count over signal is output from the phase correction timer 140 whose timer value is controlled based on the phase amount command shown in FIG. 27 (see the end point of the arrow shown in FIG. 27D). As shown in (E), the energization width control timer 141 starts {see the starting point of the arrow shown in (E) in FIG. 27}.

  A count over signal is output from the phase correction timer 140 {refer to the end point of the arrow shown in FIG. 27D) and a count over signal is output from the energization width control timer 141 {in FIG. 27E Referring to the end point of the arrow shown in FIG. 27], the inverter mode is advanced by one step as shown in (N) of FIG. 27, and the inverter circuit 121 is changed from (F) of FIG. 27 to (K) of FIG. The on / off states of the switching transistors 121u1, 121u2, 121v1, 121v2, 121w1, 121w2 are switched corresponding to the inverter mode. Further, each switching transistor is chopper-controlled by the PWM unit 144 based on the inverter output voltage as shown in FIG. The broken line shown in (L) in FIG. 27 is the output (average voltage) of the PI calculating unit 137, and the solid line shown in (L) in FIG. 27 is the primary component compensation model calculating unit 136. Output (compensation voltage).

  The phase control is a control system that is delayed by one sample due to the timer processing.

  FIG. 28 is based on the idea of maximum torque control, and the voltage phase waveform {see FIG. 28] when the compressor is driven by controlling the voltage phase and the voltage amplitude in association with each other (see FIG. 28A). Middle (see (B)}, phase current waveform {see (C) in FIG. 28} and inverter DC current waveform {see (D) in FIG. 28], FIG. 29 is based on the idea of maximum torque control, and voltage phase Voltage amplitude waveform {see FIG. 29 (A)}, voltage phase waveform {see FIG. 29 (B)}, phase current waveform {FIG. It is a figure which shows (C) reference (29)} and an inverter DC current waveform {refer (D) in FIG. 29}. The operating conditions are low pressure: 5 kg / cm 2, high pressure: 13 kg / cm 2, and rotation speed: 20 rps.

  Comparing FIG. 28 and FIG. 29, FIG. 28 shows that the peak value of the motor winding current is small and the DC current flowing into the voltage source inverter is also small. In other words, it can be seen that more efficient driving is possible.

  FIG. 30 is a block diagram showing a configuration of a microprocessor which is a main part of still another embodiment of the synchronous motor driving apparatus of the present invention. Since the configuration other than the microprocessor is the same as that shown in FIG.

  This microprocessor calculates the period of the position signal using the period measurement timer 151 that is stopped, reset, and restarted by the interrupt process 1 upon receipt of the position signal, and the timer value when the period measurement timer 151 is stopped as input. Position signal cycle calculation unit 152, speed calculation unit 153 that calculates a speed by inputting the cycle of the position signal output from position signal cycle calculation unit 152, and outputs the current speed, and is given from the outside Speed control is performed using the speed command and the current speed output from the speed calculation unit 153 as input, and the speed control unit 154 that outputs an average voltage command and the current speed output from the speed calculation unit 153 are input. The compensation phase coefficient generation unit 155 that generates the compensation phase coefficient and the current speed output from the speed calculation unit 153 are input. As a compensation voltage coefficient generation unit 161 for generating a compensation voltage coefficient, a compensation phase pattern mode selection unit 156, a compensation voltage pattern mode selection unit 165 operated by an interrupt process 1 upon receipt of a position signal, and an average given from the outside A multiplier 157 that multiplies the phase amount command, the compensation phase coefficient output from the compensation phase coefficient generation unit 155, and the compensation phase pattern selected by the compensation phase pattern mode selection unit 156, and outputs a compensation phase amount command; The adder 158 that adds the average phase amount command given from the above and the compensation phase amount command output from the multiplier 157 and outputs the phase amount command, and the cycle of the position signal output from the position signal cycle calculation unit 152 and Timer value calculation unit that calculates and outputs a timer value with the phase amount command output from the adder 158 as an input 59, the timer value output from the timer value calculation unit 159 is set, started by the interrupt processing 1 when the position signal is received, and the count over signal is output by counting the set timer value. Phase correction timer 160, inverter mode selection unit 162 that reads and outputs a voltage pattern from memory 163 by interrupt processing 2 using a count over signal output from phase correction timer 160, and average voltage command output from speed control unit 154 A multiplier 166 that multiplies the compensation voltage coefficient output from the compensation voltage coefficient generator 161 and the compensation voltage pattern selected by the compensation voltage pattern mode selector 165 and outputs a compensation voltage command, and the speed controller 154. Output average voltage command and output from multiplier 166 An adder 167 that outputs a voltage command by adding the compensation voltage command to be input, and a voltage command output from the adder 167 and a voltage pattern output from the inverter mode selection unit 162 are input to perform pulse width modulation. And a PWM unit 144 that outputs a switch signal.

  FIG. 31 is a flowchart for explaining the processing of the microprocessor of FIG. FIG. 31 shows the interrupt process 1.

  The process of the flowchart of FIG. 31 is performed every time a position signal is received.

  In step SP1, the period measurement timer 151 is stopped. In step SP2, the value of the period measurement timer 151 is read. In step SP3, the value of the period measurement timer 151 is reset, and the period measurement timer 151 is read for the next period measurement. Start. In step SP4, the period of the position signal is calculated. In step SP5, the current speed of the motor is calculated from the calculation result of the position signal period. In step SP6, the compensation phase pattern is read based on the compensation phase pattern mode. In step SP7, the compensation phase pattern mode is advanced by one step. In step SP8, the compensation phase coefficient is read based on the current speed, and in step SP9, the compensation phase pattern is multiplied by a coefficient multiple of the average phase quantity command. In step SP10, the compensation phase amount command is added to the average phase amount command to calculate the phase amount command. In step SP11, the timer value of the phase correction timer 160 is calculated from the phase amount command, and in step SP12. , Phase correction timer 160 to correction timer In step SP13, the phase correction timer 160 is started, in step SP14, speed control is performed based on the speed command and the current speed, an average voltage command is calculated, and in step SP15, compensation is performed based on the compensation voltage pattern mode. In step SP16, the compensation voltage pattern mode is advanced by one step. In step SP17, the compensation voltage coefficient is read based on the current speed. In step SP18, the compensation voltage pattern is multiplied by the coefficient multiple of the average voltage command. The compensation voltage command is calculated, and in step SP19, the compensation voltage command is added to the average voltage command, the voltage command is output, and the process returns to the original process.

  FIG. 32 is a diagram showing signal waveforms at various parts of the synchronous motor driving apparatus shown in FIGS.

  When the compressor is driven by the synchronous motor 6, a differential voltage VNM is obtained as shown in FIG. 32A, and an integration signal 積分 VNOdt is obtained as shown in FIG. A position signal is obtained as shown in (C) of FIG.

  As shown in FIG. 32D, the phase correction timer 160 is started by the interrupt processing 1 based on the position signal {see the starting point of the arrow shown in FIG. 32D}, and in FIG. 32M 32 is output every time a count over signal is output from the phase correction timer 160 whose timer value is controlled based on the phase amount command shown in FIG. 32 (see the end point of the arrow shown in FIG. 32D). The energization width control timer 169 starts as shown in (E) {see the starting point of the arrow shown in (E) in FIG. 32}.

  A count over signal is output from the phase correction timer 160 {refer to the end point of the arrow shown in FIG. 32D), and a count over signal is output from the energization width control timer 169 {in FIG. 32E Referring to the end point of the arrow shown in FIG. 32], the inverter mode is advanced by one step as shown in (N) of FIG. 32, and the inverter circuit 121 is changed from (F) of FIG. 32 to (K) of FIG. The on / off states of the switching transistors 121u1, 121u2, 121v1, 121v2, 121w1, 121w2 are switched corresponding to the inverter mode. Further, each switching transistor is chopper-controlled by the PWM unit 164 based on the inverter output voltage as shown in FIG. The broken line shown in (L) in FIG. 32 is the output (average voltage) of the speed control unit 154, and the solid line shown in (L) in FIG. 32 is the output (compensation voltage) of the multiplier 166. It is.

  The phase control is a control system that is delayed by one sample due to the timer processing.

  FIG. 33 is a block diagram showing a configuration of a microprocessor which is a main part of still another embodiment of the synchronous motor driving apparatus of the present invention.

  The synchronous motor driving device is different from the synchronous motor driving device of FIG. 30 in that an integrated signal level detection signal output from an integrated signal level detection circuit 127 (see FIG. 21) and compensation are used in place of the compensation phase coefficient generator 155. The compensation phase coefficient generation unit 168 that obtains and outputs the compensation phase coefficient with the compensation phase pattern output from the phase pattern mode selection unit 156 as an input, the compensation phase pattern output from the compensation phase pattern mode selection unit 156, and The compensation phase coefficient output from the compensation phase coefficient generator 168 is multiplied by the multiplier 158 to obtain and output a compensation phase amount command.

  FIG. 34 is a flowchart for explaining processing for obtaining a compensation phase amount command.

  In step SP1, the compensation phase pattern mode is read. In step SP2, a compensation phase pattern (for example, sin θn) is read based on the compensation phase pattern mode. In step SP3, the integrated signal level detection signal is compared with a predetermined value, and integration is performed. It is determined whether the signal level detection signal is large.

  If it is determined that the integrated signal level detection signal is large, the compensation phase coefficient K is increased by δsin θn in step SP4. Conversely, if it is determined that the integrated signal level detection signal is small, step In SP5, the compensation phase coefficient K is decreased by δsinθn. However, δ is a constant determined empirically.

  After performing the processing of step SP4 or step SP5, in step SP6, the compensation phase pattern is multiplied by a coefficient to calculate a compensation phase amount command (= K × sin θn), and the series of processing is terminated as it is.

  Therefore, the phase fluctuation pattern is sequentially corrected so that the integrated signal level detection signal becomes a predetermined value, and efficient control can be reliably performed.

  Further details will be described.

  FIG. 35 is a diagram showing the relationship between the integrated signal level and the amplitude corresponding to the phase fluctuation. The broken line shown in FIG. 35 is the integrated signal level at which the efficiency is maximum.

Therefore, in the case where the polarity of the compensation phase pattern sin θn is negative (the phase variation advances and the compensation period),
(1) When the integral signal level detection signal is larger than a predetermined value, the coefficient K is decreased and the compensation phase amount is decreased (equivalent to decreasing the phase advance amount). As a result, the integrated signal level becomes small.

  (2) When the integral signal level detection signal is smaller than the predetermined value, the coefficient K is increased and the compensation phase amount is increased (equivalent to increasing the phase advance amount). As a result, the integrated signal level increases.

Conversely, in the case where the polarity of the compensation phase pattern sin θn is positive (the phase variation is a delay compensation period),
(1) When the integral signal level detection signal is larger than a predetermined value, the coefficient K is increased and the compensation phase amount is increased (equivalent to increasing the phase advance amount). As a result, the integrated signal level becomes small.

  (2) When the integral signal level detection signal is smaller than the predetermined value, the coefficient K is decreased and the compensation phase amount is decreased (equivalent to decreasing the phase advance amount). As a result, the integrated signal level increases.

  As a result, efficient control can be reliably performed.

It is a figure which shows the relationship between a rotor position angle and a torque, the relationship between a torque and a dq axis current, and the relationship between a rotor position angle and a dq axis current. In a brushless DC motor with a surface magnet structure, it is a figure which shows the simulation result for the fluctuation | variation of the voltage amplitude and phase on the conditions which set the electric current phase to 0 rad, and the fundamental wave of load torque. It is a figure which shows the fundamental wave of the phase and amplitude of motor three-phase current for obtaining the dq-axis current of FIG. 1, and load torque. It is a figure which shows the phase and amplitude of a motor three-phase applied voltage for obtaining the dq-axis current of FIG. It is a block diagram which shows one embodiment of the synchronous motor drive device of this invention. It is a flowchart explaining the effect | action of the synchronous motor drive device of FIG. It is a block diagram which shows the other embodiment of the synchronous motor drive device of this invention. It is a flowchart explaining the effect | action of the synchronous motor drive device of FIG. It is a block diagram which shows the further another embodiment of the synchronous motor drive device of this invention. It is a flowchart explaining the effect | action of the synchronous motor drive device of FIG. It is a block diagram which shows the further another embodiment of the synchronous motor drive device of this invention. It is a flowchart explaining the effect | action of the synchronous motor drive device of FIG. It is a block diagram which shows the further another embodiment of the synchronous motor drive device of this invention. It is a flowchart explaining the effect | action of the synchronous motor drive device of FIG. It is a block diagram which shows the further another embodiment of the synchronous motor drive device of this invention. It is a figure which shows the relationship between the compression torque of a 1 cylinder compressor, and a rotation angle. It is a figure which shows frequency distribution of compression torque. It is a block diagram which shows the further another embodiment of the synchronous motor drive device of this invention. It is a flowchart explaining an effect | action of the synchronous motor drive device of FIG. It is a figure which shows the change of a torque, a current amplitude, and a current phase when adjusting the third harmonic of a current waveform so that the third harmonic component of about 10% of the fundamental wave of torque is superimposed on the torque waveform. . It is an electric circuit diagram which shows the further another embodiment of the synchronous motor drive device of this invention. It is a block diagram which shows the structure of the microprocessor of FIG. It is a figure which shows the control model corresponding to FIG. It is a flowchart explaining the interruption process 1 of FIG. It is a flowchart explaining the interruption process 2 of FIG. It is a flowchart explaining the interruption process 3 of FIG. It is a figure which shows the signal waveform of each part of the synchronous motor drive device shown to FIG. 21, FIG. It is a figure which shows the change of a line voltage amplitude, a voltage phase, a phase current, and an inverter DC current at the time of operating a real machine by relating and controlling a voltage phase and an amplitude based on the idea of maximum torque control. It is a figure which shows the change of a line voltage amplitude, a voltage phase, a phase current, and an inverter DC current at the time of drive | operating a real machine without relating and controlling a voltage phase and an amplitude based on the idea of maximum torque control. It is a block diagram which shows the structure of the microprocessor which is the principal part of the further another embodiment of the synchronous motor drive device of this invention. FIG. 31 is a flowchart for explaining interrupt processing 1 in FIG. 30. FIG. It is a figure which shows the signal waveform of each part of the synchronous motor drive device shown to FIG. 21, FIG. It is a block diagram which shows the structure of the microprocessor which is the principal part of the further another embodiment of the synchronous motor drive device of this invention. It is a flowchart explaining the process for obtaining a compensation phase amount command. It is a figure which shows the relationship between an integral signal level and a phase variation part amplitude.

Explanation of symbols

5, 5b, 25, 45, 65, 85, 105, 121 Inverter 6 Brushless DC motor 6a Rotor 6u, 6v, 6w Stator winding 8, 28, 48, 68, 88, 108 Phase control unit 10, 30, 50, 70, 90, 110 Torque controller 18, 126 Microprocessor 19a Timer value calculator 19m Compensation phase pattern mode selector 19n Adders 122u, 122v, 122w Resistor 124a Resistor 124b Capacitor 125 Zero cross comparator

Claims (20)

    For a load having a periodic torque fluctuation, the synchronous motor (6) controlled by the inverters (5) (5b) (25) (45) (65) (85) (105) (121) A synchronous motor driving method characterized by superimposing a fluctuation amount on both an amplitude and a phase of a current waveform or a voltage waveform when torque control for suppressing speed fluctuation is performed.     The synchronous motor driving method according to claim 1, wherein the phase fluctuation amount is controlled based on the amplitude fluctuation amount controlled by the output of the torque control units (10), (30), (70), and (110).     The synchronous motor drive method according to claim 1, wherein the amount of amplitude fluctuation is controlled based on the amount of phase fluctuation controlled by the output of the torque control section (50) (90).     2. The synchronization according to claim 1, wherein the amplitude fluctuation amount is controlled based on outputs of the torque control units (10), (30), (70), and (110), and the phase fluctuation amount is controlled based on a detection amount related to efficiency. Motor drive method.     2. The synchronous motor driving method according to claim 1, wherein the phase fluctuation amount is controlled based on the output of the torque control unit (50) (90), and the amplitude fluctuation amount is controlled based on the detection amount related to the efficiency.     6. The synchronous motor driving method according to claim 1, wherein the fluctuation amount corresponds to a fundamental wave and a lower order harmonic.     The synchronous motor driving method according to claim 1, wherein the fluctuation amount corresponds to a fundamental wave.     The synchronous motor driving method according to claim 1, wherein a third-order harmonic is superimposed on the amplitude fluctuation amount.     A first neutral point voltage obtained by a resistor (122u) (122v) (122w) having one end connected to the output terminal of each phase of the inverter (121) and the other end connected to each other; The synchronous motor (6) is integrated by integrating the difference from the second neutral point voltage obtained by connecting one end of the stator winding (6u) (6v) (6w) of each phase of the synchronous motor. 9. A synchronous motor driving method according to claim 1, wherein the magnetic pole position of the rotor (6a) is detected.     10. A compressor driving method, wherein a one-cylinder compressor is driven by a synchronous motor (6) driven by the synchronous motor driving method according to any one of claims 1 to 9.     For a load having a periodic torque fluctuation, the synchronous motor (6) controlled by the inverters (5) (5b) (25) (45) (65) (85) (105) (121) When performing torque control that suppresses speed fluctuations, inverters (5) (5b) (25) (45) (65) (85) are used to superimpose fluctuation amounts on both the amplitude and phase of the current waveform or voltage waveform. ) (105) (121) Inverter control means (10) (28) (30) (48) (50) (68) (70) (88) (90) (108) (110) ( 126). A synchronous motor driving device comprising:     The inverter control means (8) (10) (28) (30) (48) (50) (68) (70) (88) (90) (108) (110) (126) 12. The synchronous motor driving apparatus according to claim 11, wherein the phase fluctuation amount is controlled based on the amplitude fluctuation amount controlled by the outputs of (30), (70) and (110).     The inverter control means (8) (10) (28) (30) (48) (50) (68) (70) (88) (90) (108) (110) (126) 12. The synchronous motor drive device according to claim 11, wherein the amplitude fluctuation amount is controlled based on the phase fluctuation amount controlled by the output of (90).     The inverter control means (8) (10) (28) (30) (48) (50) (68) (70) (88) (90) (108) (110) (126) 12. The synchronous motor driving device according to claim 11, wherein the amount of fluctuation in amplitude is controlled based on the outputs of (30), (70), and (110), and the amount of fluctuation in phase is controlled based on a detected amount related to efficiency. .     The inverter control means (8) (10) (28) (30) (48) (50) (68) (70) (88) (90) (108) (110) (126) 12. The synchronous motor driving device according to claim 11, wherein the phase fluctuation amount is controlled based on the output of (90), and the amplitude fluctuation amount is controlled based on the detection amount related to the efficiency.     The inverter control means (8) (10) (28) (30) (48) (50) (68) (70) (88) (90) (108) (110) (126) The synchronous motor drive device according to any one of claims 11 to 15, wherein one corresponding to a wave and a lower harmonic is adopted.     The inverter control means (8) (10) (28) (30) (48) (50) (68) (70) (88) (90) (108) (110) (126) The synchronous motor drive device according to any one of claims 11 to 15, wherein a device corresponding to a wave is adopted.     The inverter control means (8) (10) (28) (30) (48) (50) (68) (70) (88) (90) (108) (110) (126) The synchronous motor drive device according to any one of claims 11 to 15, wherein a third-order harmonic is superimposed on the synchronous motor drive.     Resistors (122u) (122v) (122w) having one end connected to the output terminal of each phase of the inverter (121) and the other ends connected to each other to obtain the first neutral point voltage; The stator windings (6u) (6v) (6w) of the respective phases of the synchronous motor having one end connected to each other to obtain the second neutral point voltage, the first neutral point voltage, and the second neutral point voltage Integration means (124a) (124b) for integrating the difference from the point voltage, and magnetic pole position detection means (125) for detecting the magnetic pole position of the rotor (6a) of the synchronous motor (6) based on the integration signal. The synchronous motor drive device according to any one of claims 11 to 18.     A compressor drive device, wherein a one-cylinder compressor is driven by a synchronous motor (6) driven by the synchronous motor drive device according to any one of claims 11 to 19.

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