JP2010200498A - Motor control apparatus - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide a motor control apparatus which eliminates torque fluctuations, even if the sampling of motor electrical angle and motor current is carried out at unequal intervals. <P>SOLUTION: A carrier period variable section 6 diffuses the spectra of higher harmonics generated by an inverter 2, by changing a carrier period used by a PWM control section 18. An electrical angle compensating section 15 outputs an electrical angle compensating an electrical angle of a motor 3 detected by an angle sensor 5, to a 2-phase/3-phase conversion section 14, according to a carrier period changed by the carrier period variable section 6. The PWM control section 18 controls the inverter 2 at the carrier period, changed by the carrier period variable section 6 so as to obtain a three-phase target value which is outputted by the 2-phase/3-phase conversion section 14. <P>COPYRIGHT: (C)2010,JPO&INPIT

Description

  The present invention relates to a motor control device that controls a motor by digitally controlling the output power of an inverter.

  For example, a configuration described in Patent Document 1 is known as a three-phase AC motor digital control device that controls a three-phase motor by digitally controlling the output power of the inverter.

  This conventional digital motor control device acquires a motor electrical angle and a motor current value at every predetermined sampling period, calculates a motor rotational speed from the motor electrical angle, and d-axis according to the motor rotational speed and a torque command value. , Q-axis current command value is set. Next, based on this current command value, the d-axis and q-axis voltage command values are set by PI control, converted to voltage command values for U, V, and W phases by 2-phase / 3-phase conversion, and the motor is controlled by PWM control. The current was controlled.

JP-A-11-332298

  However, in the above conventional digital control method, if the frequency of the PWM carrier is made variable in order to reduce the influence of noise caused by the harmonics of the inverter, the sampling of the motor electrical angle and the motor current value becomes unequal intervals. For this reason, the control cycle is also unequal, and the motor electrical angle is calculated during one control cycle in which the voltage command values for the U, V, and W phases are calculated from the d-axis and q-axis voltage command values by 2-phase / 3-phase conversion. When the amount of advance also changes, the two-phase current values of the d-axis and q-axis also change, and there is a problem that torque fluctuation occurs.

  The present invention has been made in view of the above-described conventional problems. The motor electrical angle and the motor current are sampled at unequal intervals, and the torque command value and the actual torque average are also used in the motor digital control used for the control calculation. An object of the present invention is to provide a motor control device that can eliminate the deviation from the value and eliminate the torque fluctuation.

  In order to achieve the above object, the present invention provides a power converter that supplies power to a motor, a motor control unit that controls the motor by controlling the power that the power converter supplies to the motor, and detects the motor current. In a motor control device having a current detection means and an electrical angle detection means for detecting a motor electrical angle, the motor control unit generates a current command based on a torque command value for the motor and the number of rotations of the motor. Based on this, the frequency of the carrier signal of the PWM control means for outputting the control signal to the power converter is made variable, the motor electrical angle and the motor current are sampled in synchronization with the carrier cycle, and the carrier cycle changing means is changed. The power converter is controlled by the electrical angle compensated according to the carrier frequency. .

  According to the present invention, even if the carrier period changes, the power converter is configured to estimate the motor electrical angle that travels between the sampling of the motor current and electrical angle and the output of the power converter and corrects the current command value. Since the deviation between the torque command value and the actual torque is eliminated, there is an effect that torque fluctuations during the sampling period can be eliminated.

It is a block diagram explaining the structure of the motor control apparatus of Example 1. FIG. It is a block diagram explaining the detail of the electrical angle compensation part of Example 1. FIG. It is a figure explaining the example of the motor which Example 1 controls. (A) The time change of the carrier frequency which the carrier frequency variable part of Example 1 outputs, (b) It is a figure explaining the example of a waveform of the carrier signal which the carrier frequency variable part of Example 1 outputs. It is a figure explaining the motor which Example 1 controls. (A) Time variation of the carrier frequency output from the carrier frequency variable unit according to the first embodiment, (b) Output from the electrical angle correction value calculating unit according to the first example, and (c) The carrier cycle variation calculating unit according to the first example. It is a figure explaining an output. It is a block diagram explaining the principal part structure of the motor control apparatus of Example 2. FIG. It is a time chart explaining the process of the electrical angle compensation part of Example 2. (A) The time change of the carrier frequency changed with the carrier frequency variable part, (b) It is a figure explaining the output of the average electrical angle calculating part of Example 2. FIG. It is a block diagram explaining the principal part structure of the motor control apparatus of Example 3. FIG. (A) The flowchart of the speed control process in Example 3, (b) The flowchart of the electric current control process in Example 3. FIG. It is a block diagram explaining the principal part structure of the motor control apparatus of Example 4. FIG. It is a time chart explaining the process of the electrical angle compensation part of Example 4. It is a figure explaining the electrical angle after external input, carrier frequency, carrier period variation compensation, and external input delay compensation in Example 4.

  Next, embodiments of the present invention will be described in detail with reference to the drawings. In addition, although each Example described below is not specifically limited, it is an Example suitable for the motor control of a hybrid vehicle or an electric vehicle.

  FIG. 1 is a block diagram showing a first embodiment of a motor control device according to the present invention. In FIG. 1, a motor control device that controls a motor 3 to be controlled includes an inverter 2 that supplies three-phase power to the motor 3, current sensors 7u and 7w that detect a current supplied from the inverter 2 to the motor 3, and a motor. 3, an angle sensor 5 that detects an electrical angle 3, and a motor control unit 1 that controls the inverter 2 based on the electrical angle detected by the angle sensor 5 and a torque command value given from the outside.

  The angle sensor 5 detects the motor electrical angle θ of the motor 3, and the current sensors 7 u and 7 w detect U- and W-phase current values Iu and Iw of the three-phase current of the motor 3. The inverter 2 supplies motor driving power to the motor 3 based on the command value of the motor control unit 1.

  The motor control unit 1 performs feedback control of the motor current supplied to the motor 3 using the motor electrical angle θ detected by the angle sensor 5 and the current values Iu and Iw detected by the current sensors 7u and 7w. .

  The configuration of FIG. 1 is different from the conventional digital control device in that the carrier cycle variable unit 6 and the electrical angle compensation unit 15 are new, and the other elements are common. In reality, the motor control unit 1 is constituted by a single microcomputer, and the arithmetic processing of each part shown in FIG. 1 is executed by a control arithmetic program. Here, the functional configuration is shown separately.

Hereinafter, a digital control method of the motor 3 by the motor control unit 1 will be described. The motor control unit 1 takes in a motor electrical angle θ and a three-phase alternating current detection value from the angle sensor 5 and the current sensors 7 u and 7 w at every predetermined sampling period. The rotation speed detection unit 17 calculates the motor rotation speed N from the detection value θ of the angle sensor 5 and outputs it to the current command value calculation unit 12.
The current command value calculation unit 12 refers to the current map stored in advance in the ROM based on the torque command value input from the outside and the motor rotation number N from the rotation number detection unit 17. Current command values Id * and Iq * are calculated and output to the current PI controller 13. The three-phase / two-phase converter 16 performs a three-phase / two-phase conversion operation using the motor electrical angle θ on the current detection values Iu, Iw detected by the current sensors 7u, 7w, and a two-phase current detection value Id. (ave) and Iq (ave) are calculated and output to the current PI controller 13.

  The current sensors 7u and 7w detect only the U-phase and W-phase AC currents of the U, V, and W three-phase AC currents, but the three-phase current values have a relationship of Iu + Iv + Iw = 0, so the v-phase current Iv Can be calculated from the current values of the other two phases. The 2-phase / 3-phase conversion calculation by the 2-phase / 3-phase converter 14 will be described later.

  The current PI controller 13 performs a two-phase voltage command value Vd by proportional integral (PI) control calculation from the input two-phase current command values Id * and Iq * and the two-phase current detection values Id (ave) and Iq (ave). * And Vq * are calculated and output to the 2-phase / 3-phase converter 14. The two-phase / three-phase converter 14 converts the two-phase voltage command values Vd * and Vq * into U, V, and W three-phase voltage command values Vu *, Vv *, and Vw * by a two-phase / three-phase conversion process described later. The converted signal is output to the PWM controller 18. The PWM control unit 18 converts the voltage command values Vu *, Vv *, and Vw * into PWM signals, thereby controlling the inverter 2 and controlling the power supplied to the motor 3.

  FIG. 2 is a diagram showing a detailed configuration of the electrical angle compensator 15. The electrical angle compensation unit 15 includes an electrical angle correction value calculation unit 151, a carrier cycle variation calculation unit 152, and an adder 153. The electrical angle compensator 15 calculates the electrical angle θ detected from the angle sensor 5 and the output θ ′ of the electrical angle correction value calculator 151 that calculates the compensation value for a certain period, and the electrical angle according to the change of the carrier period. The electrical angle θ ″ compensated by adding the output Δθ (n) of the carrier period variation calculation unit 152 that calculates the value by the adder 153 is output to the two-phase / three-phase conversion unit 14.

The two-phase / three-phase conversion process by the two-phase / 3-phase converter 14 is performed as follows. Two-phase / 3-phase conversion means coordinate conversion of d, q axis to U, V, W axis. Here, d, q axis, U, V, W axis are 3 homologous in FIG. Axis that penetrates the three-phase coils 30u, 30v, and 30w arranged every 120 degrees in the initial motor model are the U axis, V axis, and W axis, respectively, and the axis that penetrates the N pole of the permanent magnet 31 that is the rotor in the positive direction. Is the d-axis, the axis orthogonal to the d-axis is the q-axis, the electrical angle θ represents the angle between the U-axis and the d-axis, and the two-phase / three-phase conversion of the voltage command value Calculated by the formula.

  Here, the carrier cycle variable unit 6 will be described. The carrier cycle variable unit 6 is a functional block that changes the frequency (cycle) of a carrier signal when performing PWM comparison in the PWM control unit 18. FIG. 4A shows a change in the carrier period, and FIG. 4B shows a PWM comparison carrier signal at that time. FIG. 4A shows an example in which the carrier frequency changes every time. Therefore, when the carrier frequency is low, the reciprocal carrier period becomes long, and thus the time is long. The reverse is also true when the carrier frequency is high. For this reason, the time with a lower carrier frequency is longer than that with a higher carrier frequency. In order to reduce this time difference, the number of times may be changed between a long carrier frequency and a short carrier frequency.

  In the example of FIG. 4 (a), the carrier frequency changes repeatedly from fcmin to fcmax to fcmin. That is, when the carrier frequency changes with time such as fcmin to fcmax, the period of the triangular wave of PWM comparison changes from Tcmax (= fcmin) to Tcmin (= fcmax) as shown in FIG. In FIG. 4B, the carrier signal in the PWM comparison is shown in a triangular wave shape, but the present invention is not limited to this, and a sawtooth wave shape is also possible.

  As described above, in the carrier cycle variable unit 6, by changing the carrier cycle, the cycle of the carrier signal of the PWM control unit 18 is changed, and the control processing is performed in the motor control unit 1 in synchronization with the cycle of the carrier signal. FIG. 5 shows the motor electrical angle that advances during the control cycle.

As shown in FIG. 5, when the three-phase current values Iu, Iv, Iw and the electrical angle θ are sampled, the start time Start is set, and θ after the carrier cycle Tc (ave) estimated from the rotation speed N of the motor 1 is obtained. The value is θ ′, and the change value of θ from the carrier period Tc (min) to Tc (max) is Δθ (n). When the two-phase / three-phase conversion calculation is performed using the θ ′ and Δθ (n) in the equation 1, and the result is output, the actual electrical angle (= rotor position) has a control cycle of Tc (min) Torque fluctuation occurs due to the causal relationship of d, q-axis voltage value fluctuation → d, q-axis current value fluctuation → torque fluctuation. In general, in a three-phase synchronous motor, the torque T is expressed by Equation 2.

  Where p is the number of pole pairs, φa is the number of armature coil linkage magnetic fluxes, and Iq is the torque shaft current.

  A method of reducing the torque fluctuation as described above when the calculation process is performed in synchronization with the control period of the carrier signal that changes over time, that is, the estimated value of the motor electrical angle according to the change of the control period The method of changing the is performed as follows.

When the rotational angular velocity ω, the average period Tc (ave) of the changing control period, the control period Tc (n), and the sampling electrical angle is θ,
The electrical angle advance amount θ ′ during the control average period Tc (ave) is θ ′ = ω × Tc (ave)
The difference in electrical angle advance Δθ from the control average period is Δθ = ω × {Tc (ave) −Tc (n)}.
The estimated electrical angle θ ′ after the control cycle Tc (n) is θ ″ = θ ′ + Δθ.

FIG. 6 shows changes in the carrier frequency and changes in the estimated value of the motor electrical angle. FIG. 6A shows the change over time of the carrier frequency. This will be described with the same changes as in FIG. Now, since the carrier frequency and the control period are synchronized, when the carrier frequency is low, the reciprocal carrier period becomes long. The reverse is also true when the carrier frequency is high. FIG. 6B shows an output waveform of the electrical angle correction value calculation unit 151. FIG. 6C shows an output waveform of the carrier cycle variation calculation unit 152. The sum of the output of the electrical angle correction value calculation means 151 and the output of the carrier period variation calculation unit 152 is θ ″. Two-phase / three-phase conversion of the voltage command value using this θ ″ is the following number: Calculated using equation (3).

  According to the first embodiment described above, even if the carrier period changes, the motor electrical angle that travels between the sampling of the motor current and electrical angle and the output of the power converter is estimated, and the current command value is corrected. In addition, since the power converter is controlled, the deviation between the torque command value and the actual torque is eliminated, so that there is an effect that torque fluctuations between sampling periods can be eliminated.

  FIG. 7 is a block diagram showing an electrical angle compensator 15 which is a main part of the motor control apparatus according to the second embodiment of the present invention. In the present embodiment, the configuration other than the electrical angle compensation unit 15 is the same as that of the first embodiment shown in FIG. The electrical angle compensator 15 according to the second embodiment has an average electrical angle calculator 154 and an adder 155 added to the first embodiment. The average electrical angle calculation unit 154 outputs an average value Δθ2 of the electrical angle of the motor that rotates during the current sampling period in accordance with the carrier cycle changed by the carrier cycle variable unit 6. The adder 155 outputs θ ″ ′ obtained by adding Δθ2 to θ ″ to the 2-phase / 3-phase converter 14.

  This operation will be described with reference to FIG. When discrete control is performed by digital control and the control cycle is unequal, as shown in FIG. 8, assuming that the control cycle is Tc (n), the current value sampled in the interrupt 1 and the electrical angle are used to calculate Tc. The voltage command value is output after (n) / 2, and the voltage command value is output after Tc (n) + Tc (n + 1) / 2. The same control cycle is repeated while counting n. At this time, the electrical angle used in the calculation estimates the electrical angle at the time of actual output, and normally the value after Tc (n) from the time of sampling the electrical angle is used in the calculation.

  However, in practice, the rotor of the motor also rotates during the control cycle Tc (n) where the control output value (voltage command value) is constant, and the estimated electrical angle during this period and the actual rotating electrical power There is a deviation between the angle θ, and this deviation also causes fluctuations in the d and q axis current values. This fluctuation of the current value appears almost as a torque fluctuation.

That is, it is necessary to estimate the motor electrical angle that will be in the average time of the output period. Referring to FIG. 8, interrupt 1 occurs at time 0, and each value is sampled and output after time (1/2) Tc (n). The output is held until the next interrupt 2 output is reflected. For this reason, since the motor electrical angle changes from the time when the output of interrupt 1 is reflected until the time when the output of interrupt 2 is reflected, the motor electrical angle in the average time of the calculation result output time is estimated. .
This average time is Tc (n) + (1/2) Tc (n + 1)-(1/2) Tc (n) = {Tc (n) + Tc (n + 1)} / 2.

  Assuming that the rotational angular velocity is ω, the estimated value Δθ2 of the average electrical angle can be obtained by the following equation.

Δθ2 = ω × {Tc (n) + Tc (n + 1)} / 2
θ ″ ′ = θ ″ + Δθ2
FIG. 9 shows changes in carrier frequency and changes in Δθ2 ′ when the rotation speed is constant. Using the estimated θ ′ ″ after this carrier cycle {Tc (n) + Tc (n + 1)} / 2, a 2-phase / 3-phase conversion operation is performed, and the result is output to reduce the torque fluctuation. it can.

  According to the second embodiment described above, the motor electrical angle that travels from the sampling of the motor current and electrical angle to the output of the power converter even if the carrier period changes is estimated and the current command value is corrected. In addition, since the power converter is controlled, the deviation between the torque command value and the actual torque is eliminated, so that there is an effect that torque fluctuations between sampling periods can be eliminated.

  According to the second embodiment, the power converter is controlled so that the average torque fluctuation value matches the torque command value by feeding back the average value of the rotating motor electrical angle during the current sampling period. There is an effect that torque fluctuation between cycles can be eliminated.

  FIG. 10 is a block diagram illustrating a configuration of a main part of a third embodiment of the motor control device according to the present invention. In the present embodiment, a torque command calculator 156 and a voltage command calculator 157 are added to the electrical angle compensator 15, and a voltage command between the current PI controller 13 and the two-phase / three-phase converter 14. The compensation unit 158 is added. The other configuration is the same as that of the first embodiment, and a duplicate description is omitted.

  The voltage command compensation unit 158 is provided in the motor control unit 1 to correct the voltage command values Vd * and Vq * output from the PI controller 13 in consideration of the torque fluctuation caused by the change in the carrier cycle. Therefore, the operation is shown in the following (1) to (4).

  (1) The provisional three-phase voltage command value is Equation 3.

(2) From the estimated electrical angle θ {Tc (n) / 2} to θ {Tc (n) + Tc (n + 1) / 2} expressed by Formula 5 when the temporary three-phase voltage command value is output A control cycle {Tc (n) + Tc (n + 1) obtained by integrating the average value of the two-phase voltages from θ {Tc (n) / 2} to θ {Tc (n) + Tc (n + 1) / 2}. )} / 2.

(3) Deviations ΔVd and ΔVq between the command value and the average value of each of the two axes are calculated by Equation 6.

Therefore, the corrected two-phase voltage command values Vd ** and Vq ** are expressed by Equation 7.

(4) The corrected three-phase voltage command values Vu **, Vv **, and Vw ** are as shown in Equation 8.

  Next, torque control calculation processing according to the third embodiment will be described with reference to the control flowchart of FIG. The motor control unit 1 executes the speed control processing routine of FIG. 11A and the current control processing routine of FIG. In this embodiment, the speed control processing routine is executed at a constant cycle, the current control processing routine is executed at unequal intervals, and the speed control processing routine is executed at a longer cycle than the current control processing routine. To do.

  In the speed control processing routine, first, a torque command value is read in step S10, the rotational speed is calculated from the output of the angle sensor 2 in step S12, and the process proceeds to step S14.

  In step S14, d-axis and q-axis current command values Id * and Iq * are calculated from the torque command value and the rotational speed read in the previous step with reference to map data stored in advance in the ROM. The process ends.

  Next, in the current control processing routine, first, in step S20, the three-phase current detection values Iu and Iw of the current sensors 7u and 7w and the electrical angle θ that is the detection value of the angle sensor 5 are read.

  Next, in step S22, using the three-phase current values Iu, Iv, Iw and the electrical angle θ, a three-phase / two-phase conversion operation is executed to calculate the two-phase current values Id, Iq.

  Next, in step S24, the d-axis and q-axis current command values Id * and Iq * calculated in the speed control processing routine are read, and the process proceeds to step S26.

  In step 26, the two-phase current values Id and Iq calculated in step S22 and the two-phase current command values Id * and Iq * acquired in step S24 are subjected to proportional-integral (PI) control calculation to execute the biaxial voltage Command values Vd * and Vq * are obtained.

  Next, in step S28, using the read electrical angle θ and the current carrier cycle Tc (n) (= 1 / fc (n)) w, the motor electrical angle advance amount Δθ between the control cycles and the estimated electrical angle θ ′. Is calculated.

  In step S30, using the d-axis and q-axis voltage command values Vd * and Vq * calculated in step S26 and the estimated electrical angle θ ′ calculated in step S28, the two-phase / three-phase conversion operation of Formula 8 is performed. And three-phase voltage command values Vu **, Vv **, and Vw ** are calculated.

  In step S32, these three-phase voltage command values Vu **, Vv **, Vw ** are converted into PWM signals and output to the inverter 2.

  In step S34, the carrier frequency counter n is counted up, a carrier cycle Tc (n) to be used next is set, and the process proceeds to step S36.

  In step S36, it is determined whether the counter n has reached the upper limit m. If the counter n has reached the upper limit m, the process returns to the main routine. Thereafter, when the current control process routine is started again, the counter n is reset to 1. If the upper limit m has not been reached, the process returns to step S22. And control of these steps S20-S36 is repeatedly performed at unequal intervals for every interruption of current control processing.

  According to the third embodiment described above, the motor electrical angle that travels from the sampling of the motor current and electrical angle to the output of the power converter even if the carrier period changes is estimated and the current command value is corrected. In addition, since the power converter is controlled, the deviation between the torque command value and the actual torque is eliminated, so that there is an effect that torque fluctuations between sampling periods can be eliminated.

  Further, according to the third embodiment, by feeding back the average value of the motor current, the power converter is controlled so that the average current value matches the current command value, so that torque fluctuations between sampling periods can be eliminated. There is an effect.

  FIG. 12 is a block diagram illustrating a configuration of a main part of the motor control device according to the fourth embodiment of the present invention. In this embodiment, the motor control unit 1 has an external input detector 19 and the electrical angle compensator 15 is different from the first embodiment shown in FIGS. It has the composition which has. The other configuration is the same as that of the first embodiment, and a duplicate description is omitted.

  When the output of the external input detector 19 is input to the external input calculation unit 159, the external input calculation unit 159 calculates and outputs an electrical angle Δθloss corresponding to a calculation delay time for processing of the external input. . The adder 160 outputs θ ″ ″ obtained by adding θ ″ and Δθloss to the 2-phase / 3-phase converter 14. This is a method for correcting a motor electrical angle in consideration of torque fluctuation caused by an external input being input and a change in calculation time, and its operation will be described below with reference to FIG.

  FIG. 13 shows an example in which processing 1 and processing 2 are different from FIG. 8, and processing 2 has a higher processing priority than processing 1.

  Consider a case where the output of another process 2 is input while the interrupt 1 in the process 1 for motor control is performed. As this external input, for example, human input is conceivable, and AD conversion or the like is given as external input processing. It is assumed that this AD conversion starts during the calculation of interrupt 1 in process 1. In the process 1, the output of the interrupt 1 is finished and the interrupt 2 is started. When the external input process of the process 2 is output during the interrupt 2 calculation, the interrupt 2 calculation is interrupted. If this interruption time is Tloss, the output time of interrupt 2 changes from the case where there is no interruption.

  When the above interruption is likely to occur, that is, when AD conversion of high priority processing is likely to be output, the output interruption time and the estimated electrical angle are used in the calculations of the first, second, and third embodiments. This reduces torque fluctuations.

  Further, by calculating Δθloss as follows, as shown in FIGS. 13 and 14, Δθloss is calculated as an interruption time in the electrical angle estimation in the control period after the external input (high priority) AD conversion is started. It may be added to the estimated motor electrical angle θ ″.

Δθloss = ω × Tloss
θ ”” = θ ”+ Δθloss
In the above case, the estimated value of the motor electrical angle in the average time of the calculation result output time is as follows. The average output time, including the time at which it is interrupted, is
Tc '(ave) = Tc (n) + (1/2) Tc (n + 1) + Tloss- (1/2) Tc (n)
= [{Tc (n) + Tc (n + 1)} / 2 + Tloss].

If the rotational angular velocity is ω, the estimated average electrical angle θ ”is
θ ″ ”= ω × Tc ′ (ave)
The calculation may be performed as follows.

  Thus, according to the motor control device of the fourth embodiment, instead of directly using the sampling value as the motor electrical angle θ used for the two-phase / three-phase conversion, as described above, the motor electrical angle that advances during the control cycle is determined. Since the inverter is controlled by compensation, torque fluctuations between control cycles can be eliminated.

  According to the fourth embodiment described above, the motor electrical angle that travels from the sampling of the motor current and electrical angle to the output of the power converter even if the carrier period changes is estimated and the current command value is corrected. In addition, since the power converter is controlled, the deviation between the torque command value and the actual torque is eliminated, so that there is an effect that torque fluctuations between sampling periods can be eliminated.

  Further, according to the fourth embodiment, even when the calculation time for calculating the voltage command value from the sampling of the current value and the electrical angle is changed by the output process due to the external input, the torque fluctuation due to the change of the sampling period There is an effect that can be eliminated.

1 Motor control unit 2 Inverter (power converter)
3 Motor 5 Angle sensor (electrical angle detection means)
6 Carrier cycle variable section 7u, 7w Current sensor (current detection means)
12 current command value calculation unit 13 current PI controller 14 2-phase / 3-phase conversion unit (second coordinate conversion means)
15 electrical angle compensation unit 16 3-phase / 2-phase conversion unit (first coordinate conversion means)
17 Rotation speed detector 18 PWM controller

Claims (4)

A power converter for supplying power to the motor;
A motor control unit for controlling the motor by controlling the power supplied to the motor by the power converter;
Current detection means for detecting a current value supplied to the motor;
An electrical angle detection means for detecting an electrical angle of the motor;
In a motor control device having
The motor control unit is
A rotational speed detection means for detecting the rotational speed of the motor from the output of the electrical angle detection means;
Current command value calculating means for calculating a current command value from the torque command value for the motor and the rotation speed of the motor;
First coordinate conversion means for converting the coordinates of the current from the current value detected by the current detection means and the electrical angle detected by the electrical angle detection means;
A current PI controller that PI-controls the output of the first coordinate conversion means and the current command value to generate a voltage command value;
Second coordinate conversion means for performing coordinate conversion of the voltage command value;
PWM control means for PWM-controlling the output of the second coordinate conversion means and outputting a control signal to the power conversion device;
Carrier cycle varying means for varying the frequency of the carrier signal of the PWM control means;
The electrical angle of the motor and the current value of the motor are sampled in synchronization with the carrier cycle, and the electrical angle compensated for the electrical angle detected by the electrical angle detection means based on the carrier frequency is the second coordinate. Electrical angle compensation means for outputting to the converter,
The motor control apparatus characterized by controlling the said power converter device by the electrical angle compensated according to the carrier frequency which the said carrier cycle variable means changed. The electrical angle compensation means includes
An average electrical angle calculation unit that calculates an average value of a motor electrical angle that rotates during a sampling period of the motor current and outputs the average value to the second coordinate conversion unit,
2. The motor control device according to claim 1, wherein the power conversion device is controlled by changing an output of the average electrical angle calculation means in accordance with a change in the sampling period. The electrical angle compensation means includes
Voltage command compensation means for estimating a motor electrical angle that proceeds during the sampling period, estimating a torque fluctuation value output during this period, and generating a voltage command value whose average value of the fluctuation values matches the torque command value With
The motor control device according to claim 1, wherein the power conversion device is controlled by changing an output of the voltage command compensation means in accordance with a change in the sampling period. The motor control unit is
Comprising external input detection means for detecting an external input and outputting a signal to the electrical angle compensation means;
4. The motor control device according to claim 1, wherein the electrical angle compensation unit controls the power conversion device in accordance with an output of the external input detection unit. 5.

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