JP2004166248A - High frequency component, high frequency module and communication device using them - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide a high frequency compound component which attains a wide band matching in transmission bands and minimizes an insertion loss in a fundamental frequency band. <P>SOLUTION: In the high frequency compound component, a high frequency amplifier circuit and a high frequency switch circuit (antenna switch module) on the post-stage are connected via a phase adjusting circuit, and a phase θ2 of impedance Z2 when watching the high frequency switch circuit side from a connection reference surface of the phase adjusting circuit is adjusted into areas from -125° to 90°, such that wide band matching in transmission bands is realized. Further, when a phase is defined as θ1 wherein a relation of impedance Z1 when watching the high frequency amplifier side from the connection reference surface and the impedance Z2 when watching the high frequency switch circuit side is a relation of conjugate matching of a phase θ of Z1, in the high frequency compound component, the phase θ2 of the impedance Z2 is adjusted by overlapping θ2 on a range within the conjugate matching region of θ1±90° in the required fundamental frequency band, such that the insertion loss may be also reduced. <P>COPYRIGHT: (C)2004,JPO

Description

  The present invention relates to a radio communication system that transmits and receives signals of two or more different frequencies using a single antenna, and relates to a multiband high-frequency component in which a switch circuit and a high-frequency amplifier circuit are combined and a single laminate. The present invention relates to a multiband high-frequency module and a communication device using the same.

For example, EGSM (Extended Global System for Mobile Communications) method and DCS (Digital Cellular System) method, which are popular in Europe, PCS (Personal Communication Service) method, which is popular in the United States, are adopted in Japan. There are various systems using time division multiple access (TDMA) such as PDC (Personal Digital Cellular). With the rapid spread of mobile phones in recent years, especially in major metropolitan areas in developed countries, it is difficult to connect system users in the frequency bands allocated to each system, making it difficult to connect, or connecting in the middle of a call There are problems such as disconnection. Therefore, it has been proposed that the user can use a plurality of systems to increase the number of frequencies that can be substantially used, and further expand the service area and effectively use the communication infrastructure of each system.
Conventionally, as a small and light high-frequency circuit component compatible with a plurality of systems, for example, a dual-band high-frequency switch module used in a portable communication device compatible with two systems of EGSM and DCS is disclosed in Patent Document 1. Patent Document 2 proposes a triple-band compatible high-frequency switch module used in a portable communication device compatible with three systems of EGSM, DCS, and PCS.

  FIG. 20 shows an example of a block configuration of a triple band high frequency switch module. Signals in the EGSM frequency band and DCS / PCS frequency band by a demultiplexing circuit Dip (hereinafter sometimes referred to as a demultiplexer or diplexer) connected to the shared antenna ANT terminal The first high-frequency switch SW1 switches between the EGSM transmission terminal Tx and the EGSM reception terminal Rx, and the second high-frequency switch SW2 Switches between the DCS / PCS transmission terminal Tx, the DCS reception terminal Rx, and the PCS reception terminal Rx. The low-pass filters LPF1 and LPF2 are inserted in the transmission path to reduce the amount of harmonic distortion generated by the high power amplifier. The band pass filters SAW1, SAW2, and SAW3 remove unnecessary frequency components from the received signal from the antenna ANT, and send only the necessary components to the low noise amplifier. Therefore, high power amplifiers HPA1 and HPA2 are provided before the EGSM transmission terminal Tx and the DCS / PCS transmission terminal Tx, and low noise amplifiers LNA1 and LNA2 are provided after the EGSM reception terminal Rx, the DCS reception terminal Rx, and the PCS reception terminal Rx. , LNA3 is provided.

  There is still a strong demand for reducing the size and weight of portable communication devices, and parts sharing and modularization of functions are being promoted. For example, the circuit component surrounded by the dotted line in FIG. 20 includes a transmission line and a capacitor formed by an electrode pattern in a laminated body in which dielectric sheets such as LTCC (Low Temperature Co-fired Ceramics) are stacked in multiple layers, and a diode or the like. It is realized as a multi-band antenna switch module ASM mounted on a laminate (here, antenna switches including antennas may be referred to as high-frequency switch modules; the same shall apply hereinafter). In addition, modularization within the range surrounded by the alternate long and short dash line is also realized by mounting a discrete SAW filter on the laminate.

  On the other hand, in order to output a relatively high power signal on the transmission side of the portable communication device, a high power amplifier of about several watts (in the present invention, it is called a high frequency amplifier without being distinguished from a power amplifier etc. Is used). Since mobile phones and the like need to be small and have low power consumption, a high-power amplifier that consumes most of DC power has high DC-RF power conversion efficiency (also referred to as power added efficiency) and is small. Is required. Especially in mobile phones, the high power amplifier is downsized and highly efficient because the device is small and the length of talk time per battery charge is an important selling point of the product. Is essential. Although modularization of circuit components including high-power amplifiers is an issue to be studied, it has not been realized to make a multilayer module with LTCC or the like.

As a conventional technique related to these amplifiers and antennas or antenna switch modules, Patent Document 3 discloses an antenna apparatus in which a reception-only antenna and an amplifier are mounted on a laminate, and a phase adjustment circuit is provided between them. However, this is for adjusting the phase deviation of the closed loop when the reception antenna (patch antenna) itself receives the electromagnetic wave leaking from the amplifier. That is, it is not a composite of the high frequency switch function.
Further, Patent Document 4 discloses that a multilayer substrate formed by laminating a plurality of dielectric layers incorporates a transmission line and a capacitor constituting a high-frequency switch and an amplifier, and a transistor is mounted on the multilayer substrate to form a module. Has been. However, in this case, only a concept was shown and practical problems and means when the two were integrated were not disclosed and it was difficult to realize.
Further, Patent Document 5 discloses a high-frequency transmission module in which a high power amplifier and a coupler for monitoring the output power are integrated, and matching between the two is set to non-conjugated matching at a spurious frequency. However, this high-frequency transmission module is intended for modularization of high-power amplifiers and couplers, and specific solutions for reducing insertion loss and high-frequency attenuation characteristics between high-frequency components are presented. Not.

Japanese Patent Laid-Open No. 11-225088 JP 2000-165288 A JP 2000-183612 A JP-A-10-126307 JP 2002-171137 A

As described above, there has been no example in which a multi-band antenna switch module and a high-frequency amplifier circuit are realized as a composite module in one laminated body. Therefore, portable communication device assembly manufacturers have various combinations of existing high-frequency amplifier circuits and existing high-frequency switch modules, and as a result of monitoring transmission characteristics such as conversion efficiency and harmonic generation, or antenna output characteristics, a combination with good compatibility is obtained. The reality is that the design method to be selected remains.
Therefore, it is conceivable to form a composite module in one laminate while matching the high-frequency amplifier HPA and the antenna switch module ASM (range ASM + HPA in FIG. 20). However, even if the size can be reduced by using a composite module, the same problem as that in the case of a discrete combination is not solved. For example, the output terminal of the high-frequency amplifier and the transmission terminal of the antenna switch module are each designed to match approximately 50Ω, but each is not strictly 50Ω, and is set around 50Ω as much as possible. It is common to have For example, even if the high frequency amplifier HPA side is 45Ω and in the position of 120 degrees and the antenna switch module ASM side is in the position of 52Ω and the phase of 80 degrees, it is regarded as 50Ω matching with each other. In the end, we are aiming for the vicinity of 50Ω at the center of the Smith chart in the transmission band, but we do not consider the impedance relationship including the mutual reactance. As a result, there arises a problem that the loss in the necessary frequency band becomes large and the amount of attenuation in the unnecessary frequency band is insufficient.

  On the other hand, even if the antenna switch module itself is intended to widen the transmission band, the transmission band may not be wide when measuring the overall characteristics after being connected to the high frequency amplifier. This is because a load connected to the subsequent stage of the high-frequency amplifier, in this case, the antenna switch module causes a change in the high-frequency amplifier. Specifically, the output terminal side of the high-frequency amplifier is composed of a semiconductor element in the final amplification stage and an output matching circuit that connects the output terminal in the subsequent stage, but a change occurs in the input impedance Z4 of this output matching circuit. Because.

  In view of such problems, the present invention provides a high-frequency component that realizes broadband matching in a transmission band, minimizes insertion loss in a necessary fundamental frequency band, and realizes these simultaneously. . In addition, the present invention provides a high-frequency module that is configured in a single laminated body and is reduced in size and weight, and further provides a communication device using the same.

  The present invention relates to a high-frequency component in which a high-frequency amplifier circuit and a subsequent high-frequency circuit that processes a high-frequency signal amplified by the high-frequency amplifier circuit are connected via a phase adjustment circuit. This is a high-frequency component in which the phase θ2 of the impedance Z2 when adjusting the phase of the subsequent high-frequency circuit from the connection reference plane to the circuit is adjusted in the range of −125 degrees to 90 degrees. Here, the range of −125 degrees to 90 degrees refers to the range of −125 degrees to −180 degrees clockwise from 0 point and the range of 90 degrees to 180 degrees counterclockwise on the Smith chart. The degree is set to 90 degrees.

  The present invention relates to a high-frequency component in which a high-frequency amplifier circuit and a subsequent high-frequency circuit that processes a high-frequency signal amplified by the high-frequency amplifier circuit are connected via a phase adjustment circuit. The relationship between the impedance Z1 when the high frequency amplifier side is viewed from the connection reference plane to the circuit and the impedance Z2 when the subsequent high frequency circuit side is viewed from the connection reference plane is the conjugate matching of the phase θ of Z1 Is a high-frequency component in which the phase θ2 of the impedance Z2 is phase-adjusted in a conjugate matching region of θ1 ± 90 degrees in the required fundamental frequency band, where θ1 is the phase having the above relationship.

  The present invention relates to a high-frequency component in which a high-frequency amplifier circuit and a subsequent high-frequency circuit that processes a high-frequency signal amplified by the high-frequency amplifier circuit are connected via a phase adjustment circuit. The relationship between the impedance Z1 when the high frequency amplifier side is viewed from the connection reference plane to the circuit and the impedance Z2 when the subsequent high frequency circuit side is viewed from the connection reference plane is the conjugate matching of the phase θ of Z1 When the phase in the relationship is θ1, the phase θ2 of the impedance Z2 is within the conjugate matching region of θ1 ± 90 degrees in the required fundamental frequency band and overlaps the range of −125 degrees to 90 degrees This is a high-frequency component with phase adjustment.

As the above-described high-frequency circuit, a demultiplexing circuit for demultiplexing a signal into a plurality of transmission / reception systems having different pass bands, and a demultiplexing circuit connected to the demultiplexing circuit, and switching between the transmission system and the reception system for each of the transmission / reception systems. A high-frequency switch circuit having a low-pass filter in each transmission system of the switch circuit.
Further, the subsequent high-frequency circuit can be composed of at least one of a coupler circuit, an isolator circuit, and a filter circuit, and a coupler circuit, an isolator circuit, and a filter circuit between the high-frequency amplifier circuit and the high-frequency switch circuit. It can also be configured by inserting at least one of them.

  The present invention is a high-frequency component that handles a plurality of transmission / reception systems having different passbands, a demultiplexing circuit that demultiplexes a high-frequency signal and a low-frequency signal, and a transmission system that is connected to the demultiplexing circuit. Or a switching circuit that switches connection with a receiving system, the branching circuit is configured by an LC circuit, and the switching circuit mainly includes a switching element and a transmission line, and the LC circuit of the branching circuit, the switch A switch in which at least a part of a transmission line of a circuit is constituted by an electrode pattern in a laminated body of an electrode pattern and a dielectric layer, and a chip element constituting a part of the switching element or the LC circuit is arranged on the laminated body The module unit has at least a semiconductor element, a power supply circuit, and a matching circuit, and at least part of the transmission line and the LC circuit constituting the power supply circuit and the matching circuit are A high-frequency amplifier module unit configured by an electrode pattern in the stacked body of the polar pattern and the dielectric layer, and a chip element constituting a part of the semiconductor element or the LC circuit disposed on the stacked body, and the high-frequency amplifier module The phase adjustment circuit between the switch module unit and the switch module unit is configured by a transmission line or an LC circuit, and at least a part of the transmission line or the LC circuit has an electrode pattern in the laminate of the electrode pattern and the dielectric layer. The phase adjustment circuit allows the phase θ2 of the impedance Z2 when viewed from the connection reference plane between the high-frequency amplifier module unit and the switch module unit to be in the region of −125 degrees to 90 degrees. This is a high-frequency module with phase adjustment.

  The present invention is a high-frequency component that handles a plurality of transmission / reception systems having different passbands, a demultiplexing circuit that demultiplexes a high-frequency signal and a low-frequency signal, and a transmission system that is connected to the demultiplexing circuit. Or a switching circuit that switches connection with a receiving system, the branching circuit is configured by an LC circuit, and the switching circuit mainly includes a switching element and a transmission line, and the LC circuit of the branching circuit, the switch A switch in which at least a part of a transmission line of a circuit is constituted by an electrode pattern in a laminated body of an electrode pattern and a dielectric layer, and a chip element constituting a part of the switching element or the LC circuit is arranged on the laminated body The module unit has at least a semiconductor element, a power supply circuit, and a matching circuit, and at least a part of the transmission line and the LC circuit constituting the power supply circuit and the matching circuit are: A high-frequency amplifier module unit configured by an electrode pattern in the stacked body of the polar pattern and the dielectric layer, and a chip element constituting a part of the semiconductor element or the LC circuit disposed on the stacked body, and the high-frequency amplifier module The phase adjustment circuit between the switch module unit and the switch module unit is configured by a transmission line or an LC circuit, and at least a part of the transmission line or the LC circuit has an electrode pattern in the laminate of the electrode pattern and the dielectric layer. The impedance adjustment Z1 when the high-frequency amplifier module unit side is viewed from the connection reference plane between the high-frequency amplifier module unit and the switch module unit by the phase adjustment circuit, and the switch module unit from the connection reference plane When viewed from the side, the relationship with the impedance Z2 is the conjugate matching relationship of the phase θ of Z1. When the phase is θ1, the high-frequency module is obtained by adjusting the phase θ2 of the impedance Z2 in the conjugate matching region of θ1 ± 90 degrees in the necessary fundamental frequency band.

  The present invention is a high-frequency component that handles a plurality of transmission / reception systems having different passbands, a demultiplexing circuit that demultiplexes a high-frequency signal and a low-frequency signal, and a transmission system that is connected to the demultiplexing circuit. Or a switching circuit that switches connection with a receiving system, the branching circuit is configured by an LC circuit, and the switching circuit mainly includes a switching element and a transmission line, and the LC circuit of the branching circuit, the switch A switch in which at least a part of a transmission line of a circuit is constituted by an electrode pattern in a laminated body of an electrode pattern and a dielectric layer, and a chip element constituting a part of the switching element or the LC circuit is arranged on the laminated body The module unit has at least a semiconductor element, a power supply circuit, and a matching circuit, and at least a part of the transmission line and the LC circuit constituting the power supply circuit and the matching circuit are: A high-frequency amplifier module unit configured by an electrode pattern in the stacked body of the polar pattern and the dielectric layer, and a chip element constituting a part of the semiconductor element or the LC circuit disposed on the stacked body, and the high-frequency amplifier module The phase adjustment circuit between the switch module unit and the switch module unit is configured by a transmission line or an LC circuit, and at least a part of the transmission line or the LC circuit has an electrode pattern in the laminate of the electrode pattern and the dielectric layer. The impedance adjustment Z1 when the high-frequency amplifier module unit side is viewed from the connection reference plane between the high-frequency amplifier module unit and the switch module unit by the phase adjustment circuit, and the switch module unit from the connection reference plane When viewed from the side, the relationship with the impedance Z2 is the conjugate matching relationship of the phase θ of Z1. When the phase is θ1, the phase θ2 of the impedance Z2 is adjusted in a region within a conjugate matching region of θ1 ± 90 degrees in the necessary fundamental frequency band and overlapping with a range of −125 degrees to 90 degrees. It is a high frequency module.

  The high-frequency module of the present invention has a low-pass filter in each transmission system of the switch circuit, and the low-pass filter includes an LC circuit, and the LC circuit is configured by an electrode pattern in a laminate of an electrode pattern and a dielectric layer. It is desirable that

  The present invention can be a communication device in which one common antenna is connected to the high-frequency component or high-frequency module described above.

  According to the present invention, it is possible to obtain a high-frequency component having excellent characteristics with high conversion efficiency and no loss in which transmission band is matched in the transmission band and insertion loss in the necessary fundamental frequency band is minimized. Furthermore, an inexpensive, small, and light weight multiband high-frequency module in which functions are integrated in one laminated body can be obtained. A high-performance communication device using these can be provided.

  First, the inventor of the present application examined the influence of the phase relationship between the high-frequency amplifier circuit and the subsequent high-frequency circuit on the matching bandwidth and insertion loss during transmission. As shown in FIG. 5, a model in which a high-frequency amplifier PA, an output matching circuit MN, and an antenna switch module ASM are connected and a phase shifter PS is inserted in the preceding stage is used. In FIG. 5, regarding the input impedance Z4 of the output matching circuit MN, the relationship between the phase change on the ASM side and the deviation in the transmission band of the input impedance Z4 of the output matching circuit MN was examined. FIG. 1 shows the relationship between the ASM-side phase and the Z4 impedance deviation, and FIG. 2 shows the position on the Smith chart. In FIG. 1, the horizontal axis represents the input phase on the antenna switch module ASM side, and the left vertical axis represents the MAG (impedance absolute value, radius on Smith chart) deviation of the impedance Z4 viewed from the input end of the output matching circuit. The right vertical axis shows the DEG (phase angle, angle from the center right short position on the Smith chart) deviation of the impedance Z4 as viewed from the input end of the output matching circuit. Since it is a deviation, the smaller each, the smaller the impedance change in the band, that is, the wider band matching can be achieved in a region where both are small. As an allowable range, the MAG deviation was set to 0.09 or less indicated by the horizontal axis X1. The DEG deviation was set to ± 0.6 degrees in the range indicated by the horizontal axis X2 centered on 0.

Next, place the phase shifter PS in front of the antenna switch module ASM, fix the impedance Z1 on the high frequency amplifier side seen from the connection reference plane, change the impedance Z2 on the antenna switch module side, and change the phase on the ASM side The insertion loss, fundamental frequency, and attenuation at the second and third harmonic frequencies were investigated. The results at the fundamental frequency are shown in FIG. 3 as a characteristic diagram of the ASM side input phase and insertion loss, and FIG. 4 shows the phase relationship on the Smith chart.
In the present invention, for the sake of convenience, the impedance Z1 on the high-frequency amplifier side is fixed and the impedance Z2 on the antenna switch module (high-frequency circuit) side in the subsequent stage is changed and adjusted, but conversely, the antenna switch module (high-frequency circuit) The present invention can also be implemented by fixing the impedance Z2 on the) side and changing and adjusting the impedance Z1 on the high frequency amplifier side. Furthermore, the present invention can also be implemented by appropriately adjusting both sides (high frequency amplifier HPA, antenna switch module ASM).
In the present invention, when an antenna is connected to the high frequency switch circuit and the high frequency switch module, they are called an antenna switch circuit and an antenna switch module, and there is no substantial difference between them. This is also true for the claims.

As a result of the above study, we found that there is a close relationship between phase change and insertion loss and harmonic attenuation. In summary,
(1) Broadband matching of the transmission band can be achieved by suppressing the change amount (deviation) of the impedance Z4. That is, a broadband can be realized by adjusting the phase range in which both the MAG deviation and the DEG deviation are small. This corresponds to the range in which the DEG deviation is within the range of X2 and the MAG deviation is less than or equal to X1 from FIG. The end point of the range is indicated by Δ. Thus, it can be seen that the range is −125 degrees to −180 degrees and 90 degrees to 180 degrees. Therefore, if the phase θ2 of the impedance Z2 on the antenna switch module (the subsequent high-frequency circuit) side as seen from the connection reference plane is adjusted to the range of −125 degrees to 90 degrees shown on the Smith chart of FIG. Consistency can be realized.
(2) When the phase θ2 on the antenna switch module side viewed from the reference plane is a phase θ1 (−θ) obtained by inverting the sign of the phase θ on the high frequency amplifier side, the insertion loss and the attenuation amount are minimized. That is, the insertion loss is minimized at the phase θ1 corresponding to the conjugate impedance of the impedance Z1. In FIG. 3 showing the fundamental frequency band, a peak near −130 degrees is the minimum insertion loss, which is a conjugate matching position. In addition, there is a valley at a position shifted 180 degrees from that, indicating a high insertion loss. Since Fig. 3 shows a sinusoidal waveform, if the center of the waveform is allowed to be higher, a low insertion loss can be achieved by adjusting the phase θ2 of the impedance Z2 within the range of θ1 ± 90 degrees at the fundamental frequency. it can.
(3) The insertion loss and the attenuation amount are maximized when the phase θ2 on the antenna switch module side viewed from the reference plane is the reverse phase θ0 (θ1 ± 180 degrees) of θ1. Therefore, with respect to the unnecessary frequency band, the second and third harmonics can be effectively attenuated by adjusting the phase θ2 of the impedance Z2 to the non-conjugated matching region including the antiphase θ0 of θ1.

  The conjugate impedance is an impedance having a complex conjugate relationship with a certain impedance. The relationship between impedance components having equal resistance components and reactance components having equal magnitudes and opposite signs is shown. The impedance of θ (R + jX) and the impedance of θ1 (R-jX) are complex conjugates. Such a relationship is called conjugate matching, and impedance matching is realized. Here, the conjugate matching by (R + jX) and (R−jX) is ideal and optimal, but it has been found by this examination that adjustment is possible only by considering only the reactance part (R′−jX). . Further, it is difficult to match all the impedances (R + jX), but it has been found that there is a sufficient effect if at least the phase is taken into consideration, and it is a practical adjustment means.

From the above, the first invention is intended to achieve wideband matching in the transmission band by adjusting the phase θ2 of the impedance Z2 to a region of −125 degrees to 90 degrees.
In the second invention, the phase θ2 of the impedance Z2 is adjusted in the region of the conjugate impedance θ1 ± 90 degrees to achieve low insertion loss in the fundamental frequency band.
In the third aspect of the invention, the phase θ2 of the impedance Z2 is adjusted within the conjugate matching region of θ1 ± 90 degrees and the region overlapping the range of −125 degrees to 90 degrees, thereby achieving broadband matching in the transmission band. At the same time, low insertion loss is realized in the fundamental frequency band.

  Also, in order to maximize the harmonic attenuation in the unnecessary 2nd harmonic and 3rd harmonic n frequency bands, the phase θ2 of the impedance Z2 is set to a range corresponding to non-conjugated matching in this frequency band. The phase is adjusted. The adjustment range of θ2 is a region of θ0 ± 120 degrees when 5 dB deterioration is allowed with respect to the maximum attenuation, preferably θ0 ± 90 degrees allowing 3dB deterioration, and more preferably θ0 ± 45 degrees. The phase θ2 here is for the nth harmonic frequency band, and is different from θ2 in the fundamental frequency band of the first and second inventions described above. As described above, in the harmonic circuit, the impedance Z changes depending on the frequency f. Therefore, even if the impedance Z2 is the same, if the frequency to be viewed is different, there is a different θ2. For example, in the first invention, θ2 is aimed at wideband matching, and in the second invention, θ2 is aimed at low insertion loss in the fundamental wave. And there is θ2 aiming at the maximum attenuation of the n-th harmonic.

An embodiment of the high-frequency component and high-frequency module according to the present invention will be described below by taking a cellular phone system as an example using an antenna switch module and a high-frequency amplifier.
First, in general, in a mobile phone system, in order to avoid interference with surrounding mobile phones, a control signal (power control signal) is set so that the outgoing output becomes the minimum power necessary for communication from the base station to the mobile phone. Has been sent. With an APC (Automatic Power Control) circuit that operates based on this control signal, the gate voltage is controlled in the high-frequency amplifier at the transmission side output stage so that the transmission output becomes an output necessary for a call. This is controlled by comparing the detection signal obtained by actually monitoring the power output from the high-frequency amplifier with the power control signal from the base station. In this way, mobile phone communication systems are built to make it difficult for interference to occur with other mobile phones and to maintain stable call quality by changing the output to adapt to the surrounding environment. Has been. In European digital cellular phone systems, there are two main types of output detection circuits for high-frequency amplifiers. One is to attach a coupler circuit to the output terminal of the high-frequency amplifier and detect the output power, and the other is to attach a resistor of about 1 to 10Ω to the high-frequency amplifier section and obtain the power consumption from the voltage drop. There are two ways to convert to In general, the former is realized by forming a circuit in a laminate, and the latter has various types of derivation means. For example, it is realized by circuit integration on a mounted component or a semiconductor chip. The following embodiment is a case where a semiconductor chip provided with a function of monitoring output power by the latter method in advance is used.

  FIG. 6 of the embodiment shows an equivalent circuit diagram of an antenna switch module for EGSM, DCS, and PCS triple bands. FIG. 7 shows an equivalent circuit diagram of the high-frequency amplifier, and FIG. 8 shows a circuit for explaining the phase adjustment circuit. 9 is a diagram illustrating phase adjustment, FIG. 10 is an example of a low-pass filter, and FIG. 11 is an example of a high-pass filter. FIG. 12 shows the relationship between the frequency after phase adjustment and power added efficiency, and FIG. 13 is a Smith chart showing the state of phase adjustment. FIG. 14 is a partial development view of the dielectric sheet of the composite laminated module.

  In FIG. 6, the diplexer Dip includes transmission lines L1 to L4 and capacitors C1 to C4. The transmission line L2 and the capacitor C1 form a series resonance circuit, and in the DCS band (transmission frequency: 1710 to 1785 MHz, reception frequency: 1805 to 1880 MHz) and PCS band (transmission frequency: 1850 to 1910 MHz, reception frequency: 1930 to 1990 MHz) Design to have a resonant frequency. In this example, the attenuation pole is set to 1.8 GHz. The transmission line L4 and the capacitor C3 form a series resonance circuit and are designed to have a resonance frequency in the EGSM band (transmission frequency: 880 to 915 MHz, reception frequency: 925 to 960 MHz). In this example, the attenuation pole is set to 0.9 GHz. This circuit makes it possible to demultiplex and synthesize an EGSM signal and a DCS / PCS signal. The transmission lines L1 and L3 are preferably set to a certain length so as to have a high impedance with respect to the frequency of the DCS / PCS signal. This makes it difficult to transmit DCS / PCS signals to EGSM routes. On the other hand, the capacitors C2 and C4 are preferably set to relatively small capacitance values so as to have a high impedance for the frequency of the EGSM signal. This makes it difficult for EGSM signals to be transmitted to the DCS / PCS route.

  The first switch circuit SW1 includes capacitors C5 and C6, transmission lines L5 and L6, PIN diodes D1 and D2, and a resistor R1. The lengths of the transmission lines are set so that the transmission lines L5 and L6 are λ / 4 resonators in the EGSM transmission frequency band. However, the transmission line L5 can be replaced with a choke coil whose ground level appears to be open (high impedance state) at the EGSM transmission frequency. In this case, the inductance value is preferably about 10 to 100 nH. The resistor R1 determines the current flowing through the first and second diodes D1 and D2 when the control power supply VC1 is in the high state. In this example, 100Ω to 200Ω was used. Capacitances C5 and C6 are necessary for DC cut of the control power supply. When the control power source VC1 is high, the PIN diode D2 has a parasitic inductance such as a connection wire, and therefore, series resonance with the capacitor C6 is performed to cancel this. The capacitance value of the capacitor C6 is set as appropriate.

  As described above, when the control power supply VC1 is high, both the first and second diodes D1 and D2 are turned on, the connection point between the second diode D2 and the transmission line L6 becomes the ground level, and the transmission is a λ / 4 resonator. The impedance on the opposite side of the line L6 becomes infinite. Therefore, when the control power supply VC1 is High, a signal cannot pass through the path between the diplexers Dip and EGSM Rx, and the signal easily passes through the path between the diplexers Dip and EGSM Tx. On the other hand, when the control power source VC1 is low, the first diode D1 is also turned off, and no signal can pass through the path between the diplexer Dip and EGSM Tx, and the second diode D2 is also turned off, so that between the diplexer Dip and EGSM Rx. It is easier for signals to pass through this route. With the above configuration, transmission / reception of EGSM signals can be switched.

  The second switch circuit SW2 includes capacitors C7 to C10, transmission lines L7 to L10, PIN diodes D3 to D6, and resistors R2 and R3. The lengths of the transmission lines are set so that the transmission lines L7 to L10 are λ / 4 resonators at the frequency of the DCS / PCS signal. However, the transmission lines L7 and L9 can also be replaced with a choke coil whose DCS transmission frequency is such that the ground level appears to be open (high impedance state) at the PCS transmission frequency. In this case, the inductance value is desirably about 5 to 60 nH. The resistor R2 determines the current flowing through the third and fourth diodes D3 and D4 when the control power supply VC2 is in the high state. In this example, 100Ω to 200Ω was used. The resistor R3 determines the current flowing through the fifth and sixth diodes D5 and D6 when the control power supply VC3 is in the high state. In this example, 100Ω to 2 kΩ was used. Capacitances C7, C8, and C10 are necessary for DC cut of the control power supply. When the control power supply VC2 is high, the PIN diode D4 has a parasitic inductance such as a connection wire, so the capacitance value of the capacitor C7 is set so as to resonate in series with the capacitor C7.

  As described above, when the control power supply VC2 is High, the third and fourth diodes D3 and D4 are both turned ON, and the connection point between the fourth diode D4 and the transmission line L8 becomes the ground level, and the transmission is a λ / 4 resonator. The impedance on the opposite side of the line L8 becomes infinite. Therefore, when the control power source VC2 is High, signals cannot pass through the path between the diplexers Dip to PCS Rx and the diplexers Dip to DCS Rx, and signals easily pass through the path between the diplexers Dip to DCS / PCS Tx. On the other hand, when the control terminal VC2 is low, the third diode D3 is also turned OFF, and the signal cannot pass through the path between the diplexer Dip and DCS / PCS Tx, and the fourth diode D4 is also OFF, so the diplexer Dip to PCS Rx. The signal easily passes through the path between the diplexer Dip and the DCS Rx.

  Further, when the control terminal VC3 is High, since there is a parasitic inductance such as a connection wire in the PIN diode D6, the capacitance value of the capacitor C10 is set so as to resonate in series with the capacitor C10. As a result, when the control terminal VC3 is High, both the fifth and sixth diodes D5 and D6 are turned on, the connection point of the sixth diode D6 and the transmission line L10 becomes the ground level, and the transmission is a λ / 4 resonator. The impedance on the opposite side of the line L10 becomes infinite. Therefore, when the control terminal VC3 is High, no signal can pass through the path between the DCS Rx, and the sixth diode D6 is also OFF, so that the signal easily passes through the path between the PCS Rx. On the other hand, when the control terminal VC3 is Low, the fifth diode D5 is also turned OFF, and the signal cannot pass through the path between the PCS Rx, and the signal easily passes through the path between the DCS Rx. With the above configuration, switching to DCS / PCS Tx when the control terminal VC2 is High, switching to PCS Rx when the control terminals VC2 and VC3 are Low and High respectively, and switching to DCS Rx when the control terminal VC2 and the control terminal VC3 are Low Is possible.

  The first low-pass filter LPF1 is a π-type low-pass filter including a transmission line L11 and capacitors C11 to C13. Here, L11 and C11 constitute a parallel resonance circuit, and the resonance frequency is set to be twice or three times the transmission frequency of EGSM. In this embodiment, the frequency is set to 3 times 2.7 GHz. With the above configuration, harmonic distortion included in the transmission signal on the EGSM side input from the power amplifier can be removed. Therefore, it is preferable in terms of characteristics to provide the low-pass filter LPF1 in the transmission path, but it is not always essential. The first low-pass filter LPF1 is arranged between the first diode D1 of the first high-frequency switch SW1 and the transmission line L5. This is arranged between the diplexer Dip and the first high-frequency switch SW1. Alternatively, it may be arranged between the transmission line L5 and the EGSM Tx. If a capacitor connected to the ground of the first low-pass filter LPF1 is arranged in parallel with the transmission line L5, a parallel resonant circuit is formed, and the line length of the transmission line L5 can be made shorter than λ / 4. The inductance value of the choke coil can be reduced.

The second low-pass filter LPF2 is a π-type low-pass filter including the transmission line L12 and the capacitors C14 to C16. Here, the transmission line L12 and the capacitor C14 constitute a parallel resonance circuit, and the resonance frequency is set to be twice or three times the DCS / PCS transmission frequency. In this embodiment, the frequency is set to 3.6 GHz which is doubled. With the above configuration, harmonic distortion included in the DCS / PCS side transmission signal input from the power amplifier can be removed. Therefore, it is preferable in terms of characteristics to provide the low-pass filter LPF1 in the transmission path, but it is not always essential. Similarly to the first low-pass filter LPF1, the second low-pass filter LPF2 may be disposed between the diplexer Dip and the second high-frequency switch SW2, or the transmission line L7 and the DCS transmission terminal DCS Tx. You may arrange | position between. The first and second low-pass filters LPF1 and LPF2 are configured between the diode D1 and the transmission line L5 and between the diode D3 and the transmission line L7, and are provided in the switch circuit. This is preferable in terms of circuit design, but is not essential. The low-pass filter may be provided at any position on the transmission path between the diplexer through which the transmission signal passes and the transmission terminal.
Table 1 summarizes the control logic of the above high-frequency components (antenna switch module).

  Further, depending on the embodiment, the EGSM system can be further divided into GSM850 (transmission frequency: 824 to 849 MHz, reception frequency: 869 to 894 MHz) and EGSM to be compatible with quad band. In this case, the transmission system can use a common terminal, and the reception system can be configured by connecting a switch for switching between GSM850 and EGSM to the EGSM reception terminal section of the triple-band antenna switch. Further, it can be realized by using a transmission line which is a λ / 4 resonator in the GSM850 or EGSM band instead of the switch and dividing the frequency between the two.

  Next, FIG. 7 shows a circuit diagram on the high frequency amplifier side constituting the multiband high frequency composite component. The output terminal P0 at the matching circuit end on the high frequency amplifier side is connected to, for example, the transmission terminal P1 of the EGSM Tx of the antenna switch module of FIG. 6, and serves to send the amplified transmission signal to the antenna switch side. One end of a transmission line ASL1 is connected to the output terminal P0 via a DC cut capacitor Ca2. Capacitors Ca3 and Ca4 having one end grounded are connected to the transmission line ASL1 to constitute an output matching circuit. The other end of the transmission line ASL1 is connected to the drain of a field effect switching transistor (FET) Q1 which is a kind of semiconductor element. The source of the FET Q1 is grounded, and the gate is connected to the collector of the bipolar switching element (B-Tr) Q2.

  On the other hand, the connection point between the other end of the transmission line ASL1 and the drain D of the field effect switching transistor FET Q1 is grounded via a series circuit of an inductor SL1 composed of a λ / 4 stripline and the capacitor Ca5, and the inductor SL1 A connection point with the capacitor Ca5 is connected to the drain voltage terminal Vdd1. The connection point between the gate of the field effect switching transistor FET Q1 and the collector of the bipolar switching element Q2 is grounded via the capacitor Ca6 and also connected to the gate voltage terminal Vg.

  Furthermore, the emitter of the bipolar switching element Q2 is grounded, and the base is connected to one end of the transmission line SL3. The collector of the bipolar switching element Q2 is grounded via a series circuit of an inductor SL2 and a capacitor Ca7 made of a strip line or the like, and a connection point between the inductor SL2 and the capacitor Ca7 is connected to a collector voltage terminal Vc. The connection point between the inductor SL2 and the capacitor Ca7 is also connected to the connection point between the base of the bipolar switching element Q2 and one end of the transmission line SL3. The other end of the transmission line SL3 is grounded via a capacitor Ca8 and connected to the input terminal Pin.

  In the equivalent circuits of FIGS. 6 and 7, the transmission line and the inductor are often constituted by strip lines, but may be constituted by microstrip lines, coplanar guidelines, or the like. On the amplifier circuit side, the semiconductor element Q3 and the power supply circuit can be added in the same manner, so that the amplifier circuit has three stages or more and can be configured as a high power amplifier. The transistors Q1 is FET and Q2 is B-Tr, but other types of transistors may be used. For example, Si-MOSFET, GaAsFET, Si bipolar transistor, GaAsHBT (heterojunction bipolar transistor), HEMT (high electron mobility transistor) and the like can be mentioned. Of course, an MMIC (monolithic microwave integrated circuit) in which several transistors are integrated may be used. In the present embodiment, the transmission line SL3 and the transistor Q2 are directly connected, but they may be connected via a resistor.

  Phase adjustment between the antenna switch module and the high-frequency amplifier can be performed by inserting a transmission line or an LC circuit such as a high-pass filter or a low-pass filter as a phase adjustment circuit. For example, FIG. 8 shows an embodiment of a transmission line. On the amplifier output side, there is a semiconductor element FET (field effect transistor) Q1, and power is supplied to the drain terminal D of this transistor Q1 from a power supply terminal Vdd1 via a transmission line or an inductor SL1. The drain terminal D is connected to the output terminal P0 via the transmission line ASL1 and the DC cut capacitor Ca2. On the antenna switch transmission side, a DC cut capacitor Ca1 and a transmission line ASL2 are connected to the transmission terminal P1, and a transmission line L5, which is a part of the transmission system circuit, and a low-pass filter LPF composed of an LC circuit are connected behind the transmission terminal P1. When connecting the output terminal P0 and the transmission terminal P1, one of the capacitors Ca1 and Ca2 can be omitted. Then, an LC circuit is inserted here and / or the transmission lines ASL1 and ASL2 are used as phase adjustment circuits. Since the transmission line ASL2 is required to connect the transmission terminal P1 and the low-pass filter LPF in the laminated body, it is used at the same time. In actual setting, first, adjust the length or width of the transmission line ASL1 on the amplifier output side, insert capacitors Ca3 and Ca4 with appropriate capacities at appropriate positions, and match the input impedance of the antenna switch to approximately 50Ω. Plan. Next, the length or width of the transmission line ASL2 is adjusted as necessary. Furthermore, the length and width of L5, the capacitance value of parallel C on the LPF transmission terminal side, and the like may be adjusted as necessary.

Moreover, the following became clear about the adjustment means. As shown in FIG.
(A) When the phase θ3 of the input impedance Z3 of the antenna switch when viewed from the antenna switch side of the phase adjustment circuit at the fundamental frequency is in the counterclockwise direction with respect to θ1 on the Smith chart, the target phase Adjust the length to be longer than the current transmission line length so that the region θ2 is closer to the best phase θ1 than θ3. If necessary, the length and width of L5 and the capacitance value of parallel C on the LPF transmission terminal side Are adjusted so that θ3 is moved in the clockwise direction so as to approach θ1. Note that the width of the transmission line may be narrowed here, but qualitatively, when the position at this time is in the upper half circle of the Smith chart, it moves clockwise, but when it is in the lower half circle Moves counterclockwise. Therefore, the means for narrowing the width needs to consider the current position and the rotation direction.
(B) Similarly, when the phase θ3 is in the clockwise direction with respect to θ1, the target phase region θ2 is adjusted so as to be shorter than the length of the current transmission line so as to be closer to the best phase θ1 than θ3. If necessary, adjust the length and width of L5, the capacitance value of parallel C on the LPF transmission terminal side, etc., and move θ3 counterclockwise. It should be noted that the transmission line ASL1 side is preferably adjusted to be thicker and shorter. However, when the width of the transmission line is increased, it moves counterclockwise when it is in the upper half circle of the Smith chart as opposed to making it thinner, but it moves closer to clockwise when it is in the lower half circle. In this case as well, it is necessary to consider the current position and the rotation direction.

If these adjustments are not in time or you want to make fine adjustments after modularization,
(C) When the phase θ3 is in the counterclockwise direction with respect to θ1, the low pass comprising the LC circuit of FIGS. 10 (1) to (4) so that the target phase region θ2 is closer to the best phase θ1 than θ3. Insert a filter. Here, since one of the DC cut capacitors Ca1 and Ca2 can be omitted, Ca1 is omitted here and is shown by a dotted line. Further, if the capacitor Ca1 on the antenna switch side is omitted in (2) and (4), parallel resonance can be made with the transmission line L5 falling to the ground. In this case, the transmission line L5 can be short, which is convenient for layout adjustment during lamination.
(D) When the phase θ3 is in the clockwise direction with respect to θ1, the high-pass filter comprising the LC circuits of FIGS. 11 (1) to (4) so that the target phase region θ2 is closer to the best phase θ1 than θ3. Insert. In this case, either one of the DC cut capacitors Ca1 and Ca2 can be omitted, and the other C can operate without any substantial effect. As a result, only the capacitor Ca2 or C on the high frequency amplifier side is left, and the other end is grounded. The one end of the inductor L or transmission line SL may be connected to the antenna switch side. Therefore, the phase adjustment when the high-pass filter is inserted is desirable because the circuit is simplified. The above adjusting means is also a feature of the present invention.

In this embodiment, phase adjustment was performed by inserting a high-pass filter between the EGSM Tx terminal P1 on the antenna switch module side and the EGSM output terminal P0 on the high frequency amplifier side. As a result, the phase θ2 on the antenna switch side in the transmission band (f) can be adjusted to the above-described low insertion loss and wideband matching region, and the impedance positions of the second harmonic (2f) and the third harmonic (3f) can be adjusted. It was possible to adjust to the non-conjugated alignment position.
As a result, with regard to the power added efficiency, which is important in the transmission characteristics, the effect of widening the bandwidth is prominent. FIG. 12 shows the power added efficiency in the EGSM transmission band. This plots the change in power added efficiency versus frequency, and for comparison, before phase adjustment (comparative example) and after adjustment (example) are shown in FIG. In the comparative example, the efficiency peak value was low, the loss was very large, and the waveform near the peak was steep and narrow band. On the other hand, it can be seen that after the adjustment using the present embodiment, the peak value of the efficiency is greatly improved, the insertion loss is reduced, and the waveform near the peak is gentle and wideband.

In this example, a high-pass filter composed of an 18 nH inductor and a 6 pF capacitor was inserted. FIG. 13 shows the fundamental, second harmonic, and third harmonic impedance positions on the Smith chart before and after the high-pass filter is inserted. The phase θ1 corresponding to conjugate matching in the fundamental wave is about −130 degrees. After the high-pass filter is inserted, it can be adjusted to a very effective position of about θ1-10 degrees in the phase region targeting θ2. As for the n-th harmonic, 2f can be adjusted to a region of θ0-90 degrees closer to the target θ0, and 3f can be adjusted to a region of θ0-120 degrees. In this example, the same HPA part and ASM part are used before and after adjustment, and the characteristics of the single unit are already optimized. However, if they are connected as they are, only the efficiency of the level before adjustment shown in FIG. 12 can be obtained. However, by inserting the phase adjustment circuit from this state, the mutual phase relationship can be adjusted to the optimum phase, and after the adjustment. Low level insertion loss and broadband characteristics were obtained. From the above, it was confirmed that the present invention is an effective means for improving characteristics.
In addition, since it is explained in detail in another application that it is very effective in suppressing the nth-order harmonic, it is omitted in this application.
It has also been confirmed that the same effect can be obtained for the DCS / PCS system.

FIG. 14 is a schematic diagram of a high-frequency module in which the high-frequency (antenna) switch module shown in the equivalent circuit in FIG. 6 and the high-frequency amplifier shown in the equivalent circuit in FIG. It is the expanded view which extracted the dielectric material green sheet of 7-8 layers and 13-15 layers of lower layers. In the dielectric green sheet, (1) is the uppermost layer and is composed of 15 layers in the following order, and the last sheet (15) shows the back surface of the laminate.
The dielectric green sheet (hereinafter referred to as green sheet or sheet) used in this example is made of an LTCC material that can be fired at a low temperature of 950 ° C. or lower. For example, Al, Si, Sr, 10-60 mass% in terms of Al ₂ O ₃ , 25-60 mass% in terms of SiO ₂ , 7.5-50 mass% in terms of SrO, and 20 mass% or less in terms of TiO ₂ . Ti, 0.1 to 10% by mass in terms of Bi ₂ O ₃ , 0.1 to 5% by mass in terms of Na ₂ O, 0.1 to 5% by mass in terms of K ₂ O, 0.01 to in terms of CuO A dielectric composition containing 5% by mass and 0.01 to 5% by mass of Bi, Na, K, Cu, and Mn in terms of MnO ₂ is used.

  A green sheet having a sheet thickness of 40 to 200 μm was used so that a transmission line and a capacitor can be easily formed. The electrode material used was silver. A transmission line and a capacitor capacity were formed by electrode patterns on each layer of the green sheet, and through holes were appropriately provided to constitute a circuit. The green sheets are sequentially laminated and pressure-bonded and fired at 950 ° C. to obtain a laminate module in which high-frequency components are combined. The size of the laminated body is about 13.75 mm wide x 8 mm long x 0.75 mm high, and chip elements such as diodes, transistors, chip inductors, chip capacitors, resistors, etc. are mounted on the top surface of the laminated body. A metal case (not shown) is put on the finished product. The total height after completion is about 1.8mm. However, a resin sealed package may be used instead of the metal case, and the total height in this case is about 1.5 mm. In addition, other laminates having a size of 10 mm wide × 8 mm long × 0.75 mm high were also implemented. In this case as well, the total height of the finished product is about 1.8 mm in the case of the metal case and about 1.5 mm in the case of the resin sealed package.

  The schematic configuration in the stack is that the antenna switch stack module side has transmission lines L1, L2, L3, L4, etc. constituting the duplexer and low-pass filter in the upper layer, and the duplexer, switch circuit, and low-pass filter in the intermediate layer Are mainly formed, and transmission lines L5, L6, L7, L8, L9, L10, etc. constituting a switch circuit are formed in the lower layer. On the other hand, on the high-frequency amplifier side, the upper layer mainly includes the transmission line of the first-stage matching circuit, the intermediate layer mainly includes the first-stage and subsequent-stage matching circuit, the capacitor capacity, and the lower layer mainly includes the thermal via and the second-stage matching circuit mainly including the transmission line Supply lines are mainly formed. The ground electrodes are provided as G1, G2, G3, G4, G5 and G6 on the second, third, eighth, thirteenth, fourteenth and fifteenth layers, respectively. In this example, since the intermediate layer is omitted, not all of the ground electrode, the transmission line, and the capacitor capacity are shown, but they are each formed by an electrode pattern by the arrangement as described above. As described above, there are diodes D1 to D6, transistors Q1 to Q3, chip capacitors C5 and C8, Ca5 to Ca7, resistors R1 to R3, and the like, as described above.

  The connection between the high frequency amplifier and the antenna switch module is in the upper layer. To avoid mutual interference, the transmission line ASL1 (line on the high frequency amplifier side) of the green sheet (1) and the transmission line ASL2 (antenna switch side of the green sheet (2)) Is formed at a position where the layers are changed and do not overlap vertically. In this example, a high-pass filter for phase adjustment is interposed between the two, but this comprises an LC circuit composed of a chip inductor and a chip capacitor and mounted on the upper surface of the laminate. As a result, the adjustment can be performed even after the laminate module is created, so that it is desirable that the trial adjustment does not take time. Further, since the transmission lines ASL1 and ASL2 are provided in layers different from each other on the upper and lower sides and do not interfere with each other on the projection, interference between both high-frequency components can be avoided here. Also, on the surface layer, there is a circuit pattern closer to the antenna than the low-pass filter on the antenna switch side with ASL1 and the shield electrode in between.If ASL1 and this pattern are electromagnetically coupled, the low-pass filter is not passed. Unnecessary high frequency power is output from the antenna as it is. By separating the gap with a shield electrode as in the present embodiment, the above-described coupling can be avoided, which helps to improve high-frequency characteristics.

  As shown in FIG. 14, in this composite laminated module, the electrode pattern constituting the high-frequency amplifier is formed in the left region, while the electrode pattern constituting the antenna switch module is formed in the right region, and the green sheet is 2 in all the lamination directions. It is divided into two areas. Further, a strip-shaped shield electrode SG is provided between the left and right regions of the first layer, and through-hole electrodes HG are provided in cascade from the shield electrode SG to all layers in the stacking direction. The through-hole electrode HG is connected from the shield electrode SG to the third layer ground electrode G2, the eighth layer ground electrode G3, the thirteenth layer ground electrode G4, and the bottom layer ground electrode G6. Are effective in suppressing the mutual interference between the electrode patterns in the vertical direction. When there is an allowance in dimensional arrangement, it is desirable to provide a strip-shaped shield electrode SG on the green sheet of all layers. However, in many cases, this cannot be done, so that the action of the shield electrode SG can be brought out by also using the ground electrode. Whether the shield electrode is provided or the ground electrode is also used, it is desirable that these are provided on a green sheet on which an electrode pattern of the transmission line is formed or on any one of the sheets above and below the sheet.

The interval between the through-hole electrodes HG arranged in a column needs to be taken into consideration. It is better not to widen the gap as much as possible, but the gap g between the through-hole electrodes HG is set to ¼ or less of the wavelength (λ) of the highest frequency for which interference is to be prevented from the viewpoint of manufacturing and the tendency to suppress mutual interference. There is a need. Actually, it has been found that the shielding effect increases at about λ / 10 to λ / 50. In this embodiment, the interval g is unequal, but is about λ / 20 (approximately 1 mm) to λ / 25 of the third harmonic of the DCS band (near 5.4 GHz).
Since the cascaded through-hole electrodes HG are provided intermittently, the adhesion strength between the layers is increased and the strength is increased. Here, the through holes do not necessarily have to be provided on a straight line, and may be provided by being appropriately shifted in consideration of, for example, the arrangement of electrode patterns as seen in the seventh layer and lower layers in FIG. Furthermore, the through-hole electrodes arranged in a row can be provided in parallel and shifted so as to fill the gap. In this case, a higher shielding effect can be expected. Of course, if there is a margin, it may be provided in parallel in three or more rows.
The above composite laminated module eliminates mutual interference such as noise between the high frequency components due to the ground shielding effect by the shield electrode SG and / or the ground electrode and the through hole electrode HG, and can prevent unstable operation such as oscillation of the high frequency amplifier. . In addition, it is possible to suppress the occurrence of spurious signals between the necessary signal (transmission signal) and the unnecessary signal, and it is possible to prevent the passage characteristics from deteriorating. In addition, since high-frequency components are consolidated in a single laminate, the mounting area on the printed wiring board in the mobile phone is 25% to 50% compared to the case where the conventional power amplifier and antenna switch are separately mounted on the board. Miniaturized. Therefore, it is possible to meet the needs for miniaturization and weight reduction by installing in communication devices such as small information terminals such as mobile phones and PDAs.

Further, by integrating the high-frequency amplifier HPA and the antenna switch module ASM in the laminate as in the embodiment, the length of the line connecting each circuit is shortened, and the line loss can be reduced even in the laminate, Mismatching does not occur. Further, the wiring required for connecting the two on the printed wiring board is no longer necessary, and the high-frequency module is directly mounted on the printed wiring board, so that line loss can be reduced and no matching deviation occurs. In addition, the high-frequency amplifier HPA and the antenna switch module ASM can be designed as an integrated unit, so that the impedance matching can be optimized and the matching circuits provided in each of the HPA and ASM can be simplified as before. Can be With such an optimal design, a high-frequency module that is small and has low insertion loss, high output power, and high output efficiency has been completed.
Regarding the characteristics, in the past, efficiency was about 35%, 2nd harmonic -25dBm, 3rd harmonic -25dBm at the time of GSM transmission, but according to this example, efficiency is 43% or more, 2nd harmonic. The characteristics were improved to -38 dBm or less and the triple harmonic -36 dBm or less. At this time, the characteristics at the time of DCS / PCS transmission were achieved with an efficiency of 34% or more, a second harmonic of -38 dBm or less, and a third harmonic of -36 dBm or less. As a result, the effectiveness of the present invention was confirmed not only in the GSM band but also in the DCS / PCS band.
Due to the above improvement in characteristics, when this high-frequency module is used in a mobile phone, the efficiency can be improved by about 5 to 10% compared to the case where components are separately mounted as in the conventional case. As a result, power consumption during transmission is reduced, battery life is improved, and a long-term call of about 10 to 20% is possible as a call time per charge.
In addition, as described above, mobile terminal manufacturers have previously evaluated a combination of multiple types of high-frequency amplifiers and antenna switch modules to obtain the desired characteristics, and matched them on the printed circuit board to make full use of them. Various ideas have been made such as forming a circuit, a harmonic filter, etc., but all of them can be omitted.

  The equivalent circuit of the antenna switch module shown in FIG. 6 and the equivalent circuit of the high-frequency amplifier shown in FIG. 7 are examples. For example, although the switch circuit has shown an example using a pin diode, the switch circuit can also be configured using SPnT type GaAs switches such as SPDT (Single Pole Dual Throw) and SP3T. In this case, if the pin diode switch is simply replaced with an SPDT GaAs switch, the λ / 4 line necessary for the pin diode switch is not necessary, so that there is enough room in the stack. For this reason, it is advantageous for further miniaturization and higher integration by reducing this space or forming a new functional element. Also, the whole including the duplexer can be replaced. In this case, taking a triple-band antenna switch as an example, if it is replaced with a SP5T GaAs switch, switching of all paths becomes possible. In addition, as described above, several λ / 4 lines are unnecessary, which is advantageous for further miniaturization and higher integration. However, the low-pass filter on the transmission side and various filters to be inserted to satisfy the characteristics are composed of a laminated body and / or a mounted component. In this case, since the GaAs switch is directly connected to the antenna, it is desirable that the GaAs switch has a countermeasure against electrostatic surge. When a switch that does not handle surge is used for the switch itself, it is necessary to insert a surge countermeasure circuit such as a filter composed of an LC circuit between the antenna and the GaAs switch. On the amplifier circuit side, the semiconductor element Q3 and the power supply circuit can be added in the same manner, so that the amplifier circuit has three stages or more and can be configured as a high power amplifier.

  An embodiment in which a triple-band antenna switch circuit is configured using GaAs switches will be described below with reference to FIGS. The integration with the high-frequency amplifier HPA can be implemented by replacing the left block of the block diagram of FIG. 20 with any one of the block diagrams of FIGS. In FIG. 15, SPDT (SP2T) switch SW1 and SP3T switch SW2 are used. First, for example, the diplexer Dip demultiplexes the EGSM band on the low frequency side and the DCS / PCS band on the high frequency side, switches EGSM band transmission (connected to PA1) and reception (RX1) with switch SW1, and DCS with switch SW2. It switches between three routes: DCS / PCS transmission (connected to PA2), DCS reception (RX2), or PCS reception (RX3) in the / PCS band. FIG. 16 shows an example in which only three SP2T switches are used. In this case as well, the diplexer Dip demultiplexes the EGSM band, which is the low frequency side, and the DCS / PCS band, which is the high frequency side, using the diplexer Dip, and transmits (connects to PA1) and receives the EGSM band using the switch SW1. Switch (RX1), switch DCS / PCS transmission (connected to PA2) and DCS / PCS reception with switch SW2, and switch DCS reception (RX2) and PCS reception (RX3) with switch SW3. Yes. FIG. 17 shows an example when the SP5T switch is used. In this case, the diplexer Dip is not used, and all routes are switched only by the switch. In FIGS. 15 and 16, the circuit block of the GaAs SPDT switch can be appropriately replaced with an SPDT switch using a PIN diode. Even if the antenna switch module is configured using GaAs switches as described above, the matching (phase) relationship between the high-frequency amplifier HPA and the antenna switch module ASM is described above, as in the embodiment using the PIN diode described above. The characteristics of the entire high-frequency module can be improved by performing phase adjustment with a broadband adjustment in the transmission band and with a minimum insertion loss in the fundamental frequency band by the phase adjustment circuit. Furthermore, it can also be configured as an integrated module including an APC circuit that controls the gate voltage of the output transistor based on the detection value of the output detector.

  The high-frequency module of the present invention may include a coupler circuit or an isolator circuit between the antenna switch module and the high-frequency amplifier, and a SAW filter may be inserted in the reception system path. In the above-described embodiment, the case where the function of monitoring the output power is combined with the semiconductor chip of the high-frequency amplifier has been described. On the other hand, there may be a configuration using a method of detecting power by a coupler circuit. An embodiment in this case will be described with reference to FIGS. As shown in FIG. 18, the coupler CPL is inserted between the low-pass filter LPF of the antenna switch module and the high-frequency amplifier PA. The transmission path CSL1 for high-frequency output from the high-frequency amplifier HPA shown in FIG. 19 is generally called a main line, and the transmission line CSL2 for taking out part of output power is generally called a coupled line. One end of the coupled line is sent to the detector as an output monitor and used for output power control of the high frequency amplifier HPA. On the other hand, the other end is terminated with a resistor R. Generally, it is terminated at 50Ω, but can be changed as appropriate in order to adjust the degree of coupling and isolation. In such a configuration, since a coupler is inserted between the high frequency amplifier HPA and the antenna switch module ASM, in this case, the relative phase between the high frequency amplifier and the antenna switch module is not adjusted, but the high frequency amplifier HPA is used. The phase adjustment circuit is provided for the relative phase between the coupler CPL and the coupler CPL as in the above-described embodiment to perform wideband matching in the transmission band, and to perform adjustment while minimizing the insertion loss in the fundamental frequency band. In this case, it is desirable to design the coupler CPL and the antenna switch module ASM so as to obtain desired characteristics. Furthermore, as a desirable improvement means, the relative phase between the coupler CPL and the antenna switch module ASM is also adjusted to a matching relationship in which, as described above, broadband matching is achieved in the transmission band and insertion loss in the fundamental frequency band is minimized. By implementing the present invention, the overall characteristics are remarkably improved.

  Normally, when a coupler CPL is used, a loss of about 0.2 to 0.3 dB due to a coupler circuit is generally generated. However, by optimizing the high-frequency amplifier HPA, the coupler circuit CPL, and the antenna switch module ASM in a single laminate, 25-50% compared to the conventional case where each circuit is separately mounted on a printed wiring board. Can be achieved, and high efficiency of about 2 to 7% can be achieved. This reduces power consumption during transmission and improves battery life. For example, a long-time call of about 5 to 15% is possible as a call time per charge. Furthermore, it can be configured as an integrated module including an APC circuit that controls the gate voltage of the output transistor based on the detection value of the coupler.

By connecting a common antenna to a composite laminated module in which the above-described high-frequency antenna switch module and high-frequency amplifier are combined, and using it for a wireless communication device such as a mobile phone, it is possible to meet the demand for reduction in size and weight.
Moreover, as a transmission / reception system used in the present invention, in addition to the above, PDC800 band (810 to 960 MHz), GPS band (1575.42 MHz), PHS band (1895 to 1920 MHz), Bluetooth band (2400 to 2484 MHz), The same effect can be expected for a multiband antenna switch circuit combining CDMA2000, which is expected to be popular in the US, TD-SCDMA, which is expected to be popular in China, and W-CDMA, which is expected to be popular in Europe. A multi-mode multi-band high-frequency antenna switch circuit such as a dual-band, 3-band, 4-band, or 5-band can be obtained by using the circuit in these cases.

  The high-frequency component or high-frequency module of the present invention can be used in a communication device such as an information terminal such as a mobile phone or a PDA.

In the present invention, it is a characteristic diagram showing the deviation of the impedance seen from the input phase of the antenna switch module side and the input side of the high frequency amplifier. FIG. 2 is an explanatory diagram on a Smith chart showing a phase adjustment range of impedance of FIG. 1 in the present invention. It is a characteristic diagram which shows the input phase and insertion loss by the side of the antenna switch module in a fundamental frequency in this invention. It is explanatory drawing on a Smith chart which shows the phase adjustment range of the input impedance of an antenna switch module in this invention. It is a block diagram of the model which examined phase adjustment in the present invention. 1 is an equivalent circuit diagram of a triple-band antenna switch module according to an embodiment of the present invention. 1 is an equivalent circuit diagram of a high-frequency amplifier showing an embodiment of the present invention. It is a circuit diagram near the output matching circuit terminal of the high frequency amplifier and the transmission terminal on the antenna switch module side showing an example of phase adjustment. It is explanatory drawing of the simple Smith chart explaining the procedure of phase adjustment. It is a circuit diagram which shows the low pass filter which is one of the concrete means of phase adjustment. It is a circuit diagram which shows the high pass filter which is one of the concrete means of phase adjustment. It is a characteristic view which shows the Example of the phase adjustment by this invention, and shows the characteristic of a frequency and power addition efficiency. It is a Smith chart figure which shows the condition of the phase adjustment of an Example. It is a partial development view of a dielectric green sheet showing one embodiment of a composite laminated module of the present invention. It is a block diagram of one Example which comprised the antenna switch module for triple bands using the GaAs switch. It is a block diagram of the Example which similarly used the GaAs switch. It is a block diagram of the Example using other GaAs switch. It is a block diagram which shows the Example at the time of comprising by inserting a coupler circuit between an antenna switch module and a high frequency amplifier. It is a figure which shows one Example of a coupler circuit. It is a block diagram explaining the form of the multiband antenna switch module of this invention.

Explanation of symbols

ASM: Antenna switch module (high frequency switch module)
HPA: High power amplifier (high frequency amplifier)
Dip: Diplexer
SW: Switch circuit
LPF: Low-pass filter circuit
SAW: SAW filter
L, SL, ASL: Inductor, transmission line
C, Ca: Capacitors Q1, Q2: Semiconductor switching elements
SG: Shield electrode
HG: Shield electrode with through hole

Claims (10)

A high-frequency component in which a high-frequency amplifier circuit and a subsequent high-frequency circuit for processing a high-frequency signal amplified by the high-frequency amplifier circuit are connected via a phase adjustment circuit,
The phase θ2 of the impedance Z2 when the rear high-frequency circuit side is viewed from the connection reference plane between the rear-stage high-frequency circuit and the high-frequency amplifier circuit is adjusted in a range of −125 degrees to 90 degrees. High frequency components. A high-frequency component in which a high-frequency amplifier circuit and a subsequent high-frequency circuit for processing a high-frequency signal amplified by the high-frequency amplifier circuit are connected via a phase adjustment circuit,
The relationship between the impedance Z1 when the high frequency amplifier side is viewed from the connection reference plane between the subsequent high frequency circuit and the high frequency amplifier circuit, and the impedance Z2 when the subsequent high frequency circuit side is viewed from the connection reference plane is The phase θ2 of the impedance Z2 is adjusted in the conjugate matching region of θ1 ± 90 degrees in the necessary fundamental frequency band, where θ1 is the phase in the conjugate matching relationship of the phase θ of Z1. High frequency components. A high-frequency component in which a high-frequency amplifier circuit and a subsequent high-frequency circuit for processing a high-frequency signal amplified by the high-frequency amplifier circuit are connected via a phase adjustment circuit,
The relationship between the impedance Z1 when the high frequency amplifier side is viewed from the connection reference plane between the subsequent high frequency circuit and the high frequency amplifier circuit, and the impedance Z2 when the subsequent high frequency circuit side is viewed from the connection reference plane is , When the phase in the conjugate matching relation of the phase θ of Z1 is θ1, the phase θ2 of the impedance Z2 is in the conjugate matching region of θ1 ± 90 degrees in the required fundamental frequency band and −125 degrees A high-frequency component characterized in that the phase is adjusted to an area that overlaps the range of ~ 90 degrees. The subsequent high-frequency circuit is a demultiplexing circuit that demultiplexes a signal into a plurality of transmission / reception systems having different pass bands, and a switch circuit that is connected to the demultiplexing circuit and switches the transmission system and the reception system to each of the transmission / reception systems. The high-frequency component according to claim 1, wherein each transmission system of the switch circuit is a high-frequency switch circuit having a low-pass filter. 5. The high-frequency component according to claim 1, wherein the subsequent high-frequency circuit includes at least one of a coupler circuit, an isolator circuit, and a filter circuit. A high-frequency component that handles multiple transmission / reception systems with different passbands,
A demultiplexing circuit for demultiplexing the signal on the high frequency side and the signal on the low frequency side;
A switch circuit that is connected to the branching circuit and switches a connection with a transmission system or a reception system;
The demultiplexing circuit is composed of an LC circuit, the switch circuit is mainly composed of a switching element and a transmission line, and at least a part of the LC circuit of the demultiplexing circuit and the transmission line of the switch circuit is an electrode pattern and a dielectric. A switch module part arranged on the laminate, the chip element constituting a part of the switching element or the LC circuit is constituted by an electrode pattern in the laminate with the layer;
At least a part of a transmission line and an LC circuit that includes at least a semiconductor element, a power supply circuit, and a matching circuit, and that constitutes the power supply circuit and the matching circuit are electrode patterns in the stacked body of an electrode pattern and a dielectric layer. A chip element that constitutes a part of the semiconductor element or the LC circuit, and a high-frequency amplifier module unit disposed on the laminate;
A phase adjustment circuit between the high-frequency amplifier module unit and the switch module unit is configured by a transmission line or an LC circuit, and at least a part of the transmission line or the LC circuit includes the electrode pattern and a dielectric layer. Composed of electrode patterns in the body,
The phase adjustment circuit adjusts the phase θ2 of the impedance Z2 when viewed from the connection reference plane between the high-frequency amplifier module unit and the switch module unit to the region of −125 degrees to 90 degrees. High frequency module characterized by A high-frequency component that handles multiple transmission / reception systems with different passbands,
A demultiplexing circuit for demultiplexing the signal on the high frequency side and the signal on the low frequency side;
A switch circuit that is connected to the branching circuit and switches a connection with a transmission system or a reception system;
The demultiplexing circuit is composed of an LC circuit, the switch circuit is mainly composed of a switching element and a transmission line, and at least a part of the LC circuit of the demultiplexing circuit and the transmission line of the switch circuit is an electrode pattern and a dielectric. A switch module portion arranged on the laminate, the chip element constituting a part of the switching element or the LC circuit is constituted by an electrode pattern in the laminate with the layer,
At least a part of a transmission line and an LC circuit that includes at least a semiconductor element, a power supply circuit, and a matching circuit, and that constitutes the power supply circuit and the matching circuit are electrode patterns in the stacked body of an electrode pattern and a dielectric layer. A chip element that constitutes a part of the semiconductor element or the LC circuit, and a high-frequency amplifier module unit disposed on the laminate;
A phase adjustment circuit between the high-frequency amplifier module unit and the switch module unit is configured by a transmission line or an LC circuit, and at least a part of the transmission line or the LC circuit includes the electrode pattern and a dielectric layer. Composed of electrode patterns in the body,
The phase adjustment circuit allows the impedance Z1 when the high-frequency amplifier module unit side is viewed from the connection reference plane between the high-frequency amplifier module unit and the switch module unit, and the switch module unit side is viewed from the connection reference plane. When the phase that is in the conjugate matching relationship with the phase θ of Z1 is θ1, the phase θ2 of the impedance Z2 is in the conjugate matching region of θ1 ± 90 degrees in the required fundamental frequency band. A high frequency module characterized by phase adjustment. A high-frequency component that handles multiple transmission / reception systems with different passbands,
A demultiplexing circuit for demultiplexing the signal on the high frequency side and the signal on the low frequency side;
A switch circuit that is connected to the branching circuit and switches a connection with a transmission system or a reception system;
The demultiplexing circuit is composed of an LC circuit, the switch circuit is mainly composed of a switching element and a transmission line, and at least a part of the LC circuit of the demultiplexing circuit and the transmission line of the switch circuit is an electrode pattern and a dielectric. A switch module portion arranged on the laminate, the chip element constituting a part of the switching element or the LC circuit is constituted by an electrode pattern in the laminate with the layer,
At least a part of a transmission line and an LC circuit that includes at least a semiconductor element, a power supply circuit, and a matching circuit, and that constitutes the power supply circuit and the matching circuit are electrode patterns in the stacked body of an electrode pattern and a dielectric layer. A chip element that constitutes a part of the semiconductor element or the LC circuit, and a high-frequency amplifier module unit disposed on the laminate;
A phase adjustment circuit between the high-frequency amplifier module unit and the switch module unit is configured by a transmission line or an LC circuit, and at least a part of the transmission line or the LC circuit includes the electrode pattern and a dielectric layer. Composed of electrode patterns in the body,
The phase adjustment circuit allows the impedance Z1 when the high-frequency amplifier module unit side is viewed from the connection reference plane between the high-frequency amplifier module unit and the switch module unit, and the switch module unit side is viewed from the connection reference plane. When the phase that is in the conjugate matching relationship with the phase θ of Z1 is θ1, the phase θ2 of the impedance Z2 is within the conjugate matching region of θ1 ± 90 degrees in the required fundamental frequency band. In addition, the high frequency module is characterized in that the phase is adjusted in a region overlapping with a range of -125 degrees to 90 degrees. Each transmission system of the switch circuit has a low-pass filter, and the low-pass filter includes an LC circuit, and the LC circuit is configured by an electrode pattern in a laminate of an electrode pattern and a dielectric layer. The high frequency module in any one of Claims 6-8. A communication device that transmits and receives two or more signals having different frequencies in common with one antenna, and the high-frequency component according to any one of claims 1 to 5 or the high-frequency component according to any one of claims 6 to 9. A communication device comprising the module and the one shared antenna connected thereto.

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