JP2003304697A - Motor-driven power steering device - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To enable a power source voltage to be utilized to the maximum limit, in driving a brushless motor. <P>SOLUTION: A first limit processing unti 230 limits d-axis, q-axis voltage command values v*<SB>d</SB>, v*<SB>q</SB>calculated by a proportional integration control calculation based on d-, q-axis current deviations e<SB>d</SB>, e<SB>q</SB>so that the amplitude v*<SB>m</SB>of a fundamental wave component of each phase voltage command value becomes v*<SB>m</SB>≤E<SB>d</SB>/√3 (E<SB>d</SB>is a power source voltage). A d-q/three-phase AC coordinate converter 132 converts the voltage command values v*<SB>d</SB>, v*<SB>q</SB>after being limited into phase voltage command values (fundamental wave components). A tertiary harmonic component calculation 202 calculates a tertiary harmonic component v*<SB>3</SB>of 1/6 of the fundamental component amplitude v*<SB>m</SB>in amplitude from the voltage command values v*<SB>d</SB>, v*<SB>q</SB>after the limit and a sine value sine 3θ<SB>re</SB>(3θ<SB>re</SB>is the electrical angle). Adders 221, 222 and the like add the component v*<SB>3</SB>to each phase voltage command value (fundamental component), to generate the command values v*<SB>u</SB>, v*<SB>v</SB>, v*<SB>w</SB>which correspond to the phase voltages to be applied to a motor 6. <P>COPYRIGHT: (C)2004,JPO

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to an electric power steering device that applies a steering assist force to a steering mechanism of a vehicle by a brushless motor.

[0002]

2. Description of the Related Art Conventionally, there has been used an electric power steering apparatus which applies a steering assist force to a steering mechanism by driving an electric motor according to a steering torque applied to a steering wheel (steering wheel) by a driver.

As an electric motor used in such an electric power steering apparatus, a brush motor has been widely used in the past, but in recent years, a brushless motor has also been used from the viewpoint of improving reliability and durability and reducing inertia. It is used. For this brushless motor, feedback control is performed with a sinusoidal current as follows.

A brushless motor is usually a rotor (hereinafter also referred to as "rotational field") as a field consisting of a permanent magnet.
And a stator composed of U-phase, V-phase and W-phase three-phase coils, and a target current value set according to the steering torque is I ^* , a command value of a current to be passed through each phase coil. Is expressed by the following equation. i ^* _u = I ^* sin (theta) _re ... (1a) i ^* _v = I ^* sin ((theta) _re-2 (pi) / 3) ... (1b) i ^* _w = I ^* sin ((theta) _re-4 (pi) / 3) =-i ^* _u− i ^* _v (1c) where i ^* _u is the u-phase current command value, i ^* _v is the v-phase current command value, i ^* _w is the w-phase current command value, and θ _re is the value shown in FIG. The angle of the rotating field taken clockwise with the u-phase coil as a reference (referred to as "electrical angle").

Command of voltage to be applied to brushless motor
The value is based on the deviation between the target current value and the detected motor current value.
It is calculated by the control calculation, but the phase delay is reduced.
Therefore, in this control calculation, the current command value is expressed in dq coordinates.
Voltage command value based on the d-axis and q-axis current command values
Is calculated. Here, the dq coordinates consist of permanent magnets.
In the rotating coordinate system that rotates in synchronization with the rotating field (rotor)
Therefore, the magnetic flux direction of the rotating field is the d-axis and is orthogonal to the d-axis.
The direction is defined as the q-axis. Current command value i of each phase ^* _u,
i^* _v, I^* _wIs an alternating current, but according to the dq coordinates,
The flow command value becomes DC.

The current command value of each phase expressed by the above equations (1a) to (1c) is represented by dq coordinates as follows. i ^* _d = 0 ... (2a) i ^* _q = -√ (3/2) I ^* ... (2b) Here, i ^* _d is a d-axis current command value, and i ^* _q is a q-axis current command value. .

On the other hand, regarding the motor current, when the u-phase current and the v-phase current are detected by the current detector, the d-axis current detection value i _d and the q-axis current detection value i _q are calculated from the detection results by the following equations. It is calculated. _{i d = √2 {i v sinθ} re -i u sin (θ re -2π / 3)} ... (3a) i q = √2 {i v cosθ re -i u cos (θ re -2π / 3)} (3b) where i _u is the u-phase current detection value, _iv is the v-phase current detection value, and θ _re is the electrical angle.

In the above control calculation, the brushless motor is operated on the basis of the command value i ^* _d and the detected value i _{d of the d-} axis current and the command value i ^* _q and the detected value i _{q of the} q-axis current thus calculated. The d-axis voltage command value v ^* _d and the q-axis voltage command value v ^* _q for feedback control with respect to are calculated by the following calculation (hereinafter referred to as “PI control calculation”). _{^{v * d = K p {e}} d + (1 / T i) ∫e d dt} ... (4a) v * q = K p {e q + (1 / T i) ∫e q dt} ... (4b) _{^{e d = i * d -i d}} ... (4c) e q = i * q -i q ... (4d) where, e _d is the deviation between the detected value and command value in the d-axis current (hereinafter, "d-axis current Deviation)), _eq is a deviation between a command value and a detected value in the q-axis current (hereinafter referred to as “q-axis current deviation”), Kp is a proportional gain, and Ti is
Is the integration time.

The d-axis voltage command value v ^* _d and the q-axis voltage command value v ^* _q are the voltage command values v ^* _u for the u-phase, v-phase, and w-phase.
When converted into v ^* _v and v ^* _w , that is, the voltage command value is dq
When the coordinates are converted to the three-phase AC coordinates, the following equation is obtained. v ^* _u = √ (2/3) {v ^* _d cosθ _re −v ^* _q sinθ _re } ... (5a) v ^* _v = √ (2/3) {v ^* _d cos (θ _re −2π / 3) −v ^* _q sin (θ _re −2π / 3)} (5b) v ^* _w = −v ^* _u −v ^* _v (5c)

In this way, the u-phase voltage command value v^* _u, V phase
Voltage command value v^* _v, W-phase voltage command value v ^* _wIs obtained,
These phase voltage command values v^* _u, V^* _v, V^* _wAccording to
Ratio PWM signals are generated, and the PWM signals
Power transistor that turns on / off according to duty ratio
The motor drive circuit configured using
Brushes u-phase, v-phase, and w-phase voltages according to duty ratio
Applied to the motor.

In the above control system, when the brushless motor is in a high rotation and high load state, the voltage command value, which is the operation amount of the control system, becomes large, but in reality, the power supply voltage is limited. , It is necessary to limit the voltage command value. Hereinafter, this point will be described.

When the above equations (5a) to (5c) are modified,
Each phase voltage command value v ^* _u , v ^* _v , v ^* _w can be expressed as follows. v ^* _u = −v ^* _m sin (θ _re + φ) (6a) v ^* _v = −v ^* _m sin (θ _re −2π / 3 + φ) (6b) v ^* _w = −v ^* _m sin (θ _re −4π / 3 + φ) (6c) v ^* _m = √ (2/3) √ {v ^* _d ² + (− v ^* _q ) ² } (6d) Here, when v ^* _q ≠ 0, φ = Tan ⁻¹ {v ^* _d / (− v ^* _q )}, and φ = −π / 2 when v ^* _q = 0.

Now, the waveform of each phase voltage command value is always a sine wave.
Amplitude of each phase voltage command value v^* _mTo v^* _m≤ E_d/ 2 (7) Restricted to. Where E_dIs the power supply voltage. This and
Voltage command value v for each phase at maximum amplitude^* _u, V^* _v, V^* _wIs v^* _u=-(E_d/ 2) sin (θ_re+ Φ) (8a) v^* _v=-(E_d/ 2) sin (θ_re−2π / 3 + φ) (8b) v^* _w=-(E_d/ 2) sin (θ_re-4π / 3 + φ) (8c) Thus, for example, the waveform is as shown in FIG.
Therefore, the interphase voltage v of each phase u, v, w^* _uv, V^* _vw,
v^* _wuIs     v^* _uv= V^* _u-V^* _v=-(√3 / 2) E_dsin (θ_re+ Π / 6 + φ) (9a)     v^* _vw= V^* _v-V^* _w=-(√3 / 2) E_dsin (θ_re-Π / 2 + φ) (9b)     v^* _wu= V^* _w-V^* _u=-(√3 / 2) E_dsin (θ_re+ 5π / 6 + φ) (9c) And has a waveform as shown in FIG. 9B,
All of these amplitudes are (√3 / 2) E_dIs. For this reason,
Maximum power supply voltage, ie E_d/ 2 − (− E_d/ 2) = E _dUntil
Cannot be used.

[0014]

In a conventional electric power steering system using a brushless motor, when the motor is rotating at a constant speed, a sine _{wave is} generated with respect to the angle (electrical angle) θ _re of the rotating field. Since the wave-shaped phase current is controlled so as to flow, each phase voltage (phase voltage command value) to be applied to the motor also has a sine wave shape. However, since the voltage applied to the motor is restricted by the power supply voltage E _d , if it is attempted that the voltage command values for each phase are always sinusoidal,
As described above, the maximum value of the amplitude v ^* _m of each phase voltage command value is E
_d / 2, and the maximum value of the amplitude of the interphase voltage is (√3 / 2) E _d . That is, in the conventional electric power steering apparatus using the brushless motor, there is a problem that the power supply voltage cannot be utilized to the maximum E _d when driving the motor.

Therefore, it is an object of the present invention to provide an electric power steering apparatus in which a brushless motor is used and which can utilize the power supply voltage to the maximum in driving the brushless motor. And

[0016]

According to a first aspect of the present invention, a brushless motor is driven by applying a phase voltage to the brushless motor according to a steering torque applied to an operating means for steering a vehicle. An electric power steering device for applying a steering assist force to a steering mechanism of the vehicle, the control calculation means calculating the value of each phase voltage to be applied to the brushless motor as a sinusoidal phase voltage command value; the amplitude of the sine wave representing the phase voltage command value calculated by the control operation unit, an amplitude limiting means for limiting the brushless motor to or lower than the power supply voltage E _d to drive, based on the power supply voltage E _d, the Each phase voltage command value whose amplitude is limited by the amplitude limiting means is limited to a range of -E _d / 2 or more and E _d / 2 or less when the neutral point is 0. A limiter unit that applies a limiter process to each phase voltage command value, and a motor drive unit that applies each phase voltage corresponding to the phase voltage command value after the limiter process to the brushless motor. To do.

According to the first aspect of the invention as described above, after the sinusoidal phase voltage command value having the amplitude equal to or less than the power supply voltage E _d is calculated, each phase voltage command value is equal to or more than −E _d / 2 _Since the limiter processing is performed to limit the range to _d / 2 or less, each phase voltage command value has a trapezoidal waveform when the brushless motor is in a high rotation and high load state. However, there is no period in which the voltage command values of different phases have the same value. As a result, the motor output is increased and a large steering assist force is obtained compared to the conventional case. Further, in this case, since the maximum value of the amplitude of the interphase voltage (line voltage) becomes equal to the power supply voltage, according to the present invention, the power supply voltage can be utilized to the maximum in driving the brushless motor.

A second aspect of the present invention applies a phase voltage to the brushless motor according to a steering torque applied to an operating means for steering the vehicle to drive the brushless motor, thereby assisting the steering mechanism of the vehicle. An electric power steering device for applying force, comprising: a control calculation means for calculating a value of each phase voltage to be applied to the brushless motor as a phase voltage command value of a waveform obtained by superimposing a third harmonic on a fundamental sine wave. A motor drive unit that applies each phase voltage according to each phase voltage command value calculated by the control calculation unit to the brushless motor.

According to the second aspect of the invention as described above, the absolute value of each phase voltage command value is calculated by calculating the phase voltage command value of the waveform in which the third harmonic is superimposed on the basic sine wave. Since the amplitude of the fundamental wave component (fundamental sine wave) in each phase voltage command value can be made larger than that of the conventional one while being suppressed to 1/2 or less, it is possible to improve the utilization rate of the power supply voltage in driving the brushless motor. it can. Further, in the case of the three-phase brushless motor, the third harmonic does not appear in the line voltage and the motor current does not include the harmonic component, so that the occurrence of torque ripple and vibration is reduced as compared with the first invention. Can be suppressed.

In a third aspect based on the second aspect, the control calculation means calculates a fundamental wave component value in each phase voltage to be applied to the brushless motor as a fundamental wave phase voltage command value. Calculating means, and a harmonic component calculating means for calculating a third-order harmonic component that is a harmonic component of a sine wave representing the fundamental wave phase voltage command value and has an amplitude of 1/6 of the amplitude of the sine wave. , Adding means for calculating each phase voltage command value by adding the third harmonic component to the fundamental wave phase voltage command value.

According to the third aspect of the invention, the phase voltage command value of the waveform in which the third harmonic having the amplitude of ⅙ of the amplitude of the basic sine wave is superimposed on the basic sine wave is calculated. , The phase voltage corresponding to the phase voltage command value is applied to the brushless motor. As a result, the amplitude of the fundamental wave component (fundamental sine wave) and the amplitude of the line voltage included in each phase voltage command value can be increased without distorting each phase voltage command value.

In a fourth invention based on the third invention, the fundamental wave component calculating means is a d-axis which is a magnetic flux direction of a rotating field in the brushless motor, and q which is orthogonal to the d-axis.
Introducing dq coordinates, which is a rotating coordinate system composed of an axis and the value of the voltage to be applied to the brushless motor,
The d-axis and q-axis currents in which the d-axis voltage command value and the q-axis voltage command value expressed in coordinates are calculated based on the deviation between the target current value determined according to the steering torque and the detected value of the current flowing through the brushless motor. Control operation means, and coordinate conversion means for generating the fundamental wave phase voltage command value by converting the d-axis voltage command value and the q-axis voltage command value into a voltage value of each phase of the brushless motor, The harmonic component calculating means calculates the third harmonic component v ^* _{3 according} to the following formula: v ^* ₃ = − (1/6) √ (2/3) [{− _{^{v * q + 4v * d 2}} v * q / (v * d 2
_{^{+ V * q 2)} sin3θ}} re + {- v * d + 4v * d v * q 2 / (v * d 2 +
_{^{v * q 2)} cos3θ re}} ] Here, v ^* _d is the d-axis voltage command value, v ^* _q is the q-axis voltage command value, theta _re indicates the rotational position of the rotating field magnet It is an electrical angle.

According to the fourth aspect of the invention, in the current control based on the dq axis coordinates, the third harmonic component to be superimposed on the fundamental wave phase voltage command value is calculated, for example, by four arithmetic operations and a sine wave ROM table. Since it can be obtained by referring to, the calculation load for calculating the third-order harmonic component v ^* ₃ can be reduced.

According to a fifth aspect of the present invention, in the third or fourth aspect of the invention, the control calculation means uses the amplitude of a sine wave representing the fundamental wave phase voltage command value as a power supply voltage for driving the brushless motor. It is characterized by further including an amplitude limiting means for limiting to 1 / √3 or less.

According to the fifth aspect, the amplitude of the fundamental wave phase voltage command value is limited to 1 / √3 of the power supply voltage, so that the brushless motor is in a high rotation and high load state. The maximum power supply voltage can be used without distorting the voltage command value for each phase. Further, since the line voltage does not include a harmonic component, the motor current does not include a harmonic component, so that the occurrence of torque ripple and vibration can be suppressed.

[0026]

DETAILED DESCRIPTION OF THE INVENTION Embodiments of the present invention will be described below with reference to the accompanying drawings. <1. First Embodiment><1.1 Overall Configuration> FIG. 1 is a schematic diagram showing a configuration of an electric power steering apparatus according to a first embodiment of the present invention together with a vehicle configuration related thereto. This electric power steering apparatus has a steering shaft 102 having one end fixed to a steering wheel (steering wheel) 100 as an operating means for steering, a rack and pinion mechanism 104 connected to the other end of the steering shaft 102, and a steering wheel. A torque sensor 3 that detects a steering torque applied to a steering shaft 102 by operating 100, a brushless motor 6 that generates a steering assist force for reducing a load on a driver in steering operation (steering operation), and a steering assist thereof. A ball screw drive unit 11 that transmits a force to a rack shaft, a position detection sensor 12 such as a resolver that detects a rotational position of a rotor of a brushless motor 6, and a power supply from an on-vehicle battery 8 via an ignition switch 9, Torque sensor 3 and vehicle speed sensor 4, Based on the sensor signals from 置検 detecting sensor 12 and an electronic control unit (ECU) 5 for controlling the driving of the motor 6. When the driver operates the steering wheel 100 in a vehicle equipped with such an electric power steering device, the torque sensor 3 detects the steering torque due to the operation and outputs a steering torque signal Ts indicating the steering torque. On the other hand, the vehicle speed sensor 4 detects the vehicle speed (vehicle speed) and outputs a vehicle speed signal Vs indicating the vehicle speed. The ECU 5 as a control device drives the motor 6 based on the steering torque signal Ts, the vehicle speed signal Vs, and the rotational position of the rotor detected by the position detection sensor 12.
As a result, the motor 6 generates a steering assist force, and the steering assist force is applied to the rack shaft via the ball screw drive unit 11, thereby reducing the driver's load in the steering operation. That is, the rack shaft reciprocates by the sum of the steering force due to the steering torque applied by the steering wheel operation and the steering assist force generated by the motor 6. Both ends of the rack shaft are connected to a wheel 108 via a connecting member 106 including a tie rod and a knuckle arm, and the direction of the wheel 108 changes according to the reciprocating motion of the rack shaft.

<1.2 Structure of Control Device> FIG. 2 shows a control device E in the electric power steering device.
It is a block diagram which shows the structure of CU5. This ECU 5
Is composed of a phase compensator 112, a microcomputer (hereinafter abbreviated as “microcomputer”) 10, and a motor drive unit. The microcomputer 10 executes a predetermined program stored in its internal memory, thereby executing a target current calculation unit 114, a command current direction designating unit 116, a convergence correction unit 118, an adder 120, and subtractors 122 and 124. The d-axis current PI control unit 126, the q-axis current PI control unit 128, the first limit processing unit 130, and the dq / 3-phase AC coordinate conversion unit 13
2, sign inversion adder 134, and second limit processing units 136 and 3
It functions as a motor control unit including a phase AC / dq coordinate conversion unit 138, a sine wave ROM table 140, and a rotor angular velocity calculation unit 142. The motor drive unit is hardware (circuit) that drives the three-phase brushless motor 6 including the u phase, the v phase, and the w phase based on the voltage command value output from the microcomputer 10 as the motor control unit. It is composed of a three-phase PWM modulator 150, a motor drive circuit 152, a u-phase current detector 156, a v-phase current detector 154, and a rotor angle position detector 162.

In the present embodiment, the steering torque applied to the steering shaft 102 by the operation of the steering wheel 100 is detected by the torque sensor 3, and the steering torque signal Ts output from the torque sensor 3 is the EC of the above configuration.
While being input to U5, the vehicle speed is detected by the vehicle speed sensor 4, and the vehicle speed signal Vs output from the vehicle speed sensor 4 is also input to the ECU 5. In the ECU 5, the input steering torque signal Ts is phase-compensated by the phase compensator 112, and the signal after the phase compensation is input to the target current calculation unit 114. On the other hand, the vehicle speed signal Vs output from the vehicle speed sensor 4 is the target current calculation unit 1 in the ECU 5.
14 and the convergence correction unit 118. Further, the sensor signal Sr output from the position detection sensor 12 attached to the motor 6 is input to the rotor angle position detector 162 in the ECU 5, and the rotor angle position detector 162 is detected.
Outputs a signal indicating the rotational position of the rotating field (permanent magnet) that is the rotor of the motor 6, that is, the electrical angle θ _re (see FIG. 8).
reference). The signal indicating the electrical angle θ _re is input to the sine wave ROM table 140 and the rotor angular velocity calculation unit 142.

The target current calculator 114 calculates a target current value It, which is the value of the current to be supplied to the motor 6, based on the steering torque signal Ts and the vehicle speed signal Vs. Specifically, a table (referred to as an “assist table”) showing the relationship between the value of the target current to be supplied to the motor 6 and the steering torque as a parameter to generate an appropriate steering assist force is called a target current calculation. The target current calculation unit 114, which is held in advance in the unit 114, sets the target current value It by referring to this assist table. However,
This target current value It is represented by the above-mentioned equation (2b) q
It is a value with a sign indicating a current command value corresponding to the shaft current, and its positive / negative value should generate torque in the direction of assisting steering, that is, in the direction of assisting rightward steering in the motor 6 or by steering leftward. It shows whether or not the torque in the assisting direction should be generated in the motor 6.

The command current direction designating section 116 generates a signal indicating whether the target current value It is positive or negative, that is, a signal indicating a steering assist direction (hereinafter referred to as "direction signal") Sdir, and the direction signal Sdir is converged. It is input to the section 118.
Further, the rotor angular velocity calculation unit 142 calculates the rotor angular velocity ωre on the basis of the signal indicating the electrical angle θ _re corresponding to the rotor angle, and the signal indicating the rotor angular velocity ωre is also input to the convergence correction unit 118. The convergence correction unit 118 calculates a compensation current value ic for ensuring vehicle convergence based on these signals and the vehicle speed signal Vs. The adder 120 is
This compensation current value ic is added to the above target current value It,
The added value obtained as a result is output as the q-axis current command value i ^* _q . The q-axis current command value i ^* _q is a current value corresponding to the torque that the motor 6 should generate, and is input to the subtractor 124. On the other hand, the d-axis current command value i ^* _d is input to the subtractor 122 as i ^* _d = 0 since it does not contribute to the torque.

U-phase current detector 156 and v-phase current detector 1
54 detects the u-phase current and the v-phase current of the current supplied from the motor drive circuit 152 to the motor 6, respectively.
The phase current detection value i _u and the v phase current detection value _iv are output, respectively. The sine wave ROM table 140 stores various values of the angle θ and various values of sin θ in association with each other, and stores the sine wave corresponding to the electrical angle θ _re indicated by the signal from the rotor angle position detector 162. Outputs the value sin θ _re . The three-phase AC / dq coordinate conversion unit 138 uses the sine wave value sin θ _re.
Is used to convert the above u-phase current detection value i _u and v-phase current detection value _iv into values on the dq coordinates, that is, d-axis current detection value i _d and q-axis current detection value _iq . The d-axis current detection value i _d and the q-axis current detection value i _q thus obtained
Are input to the subtractor 122 and the subtractor 124, respectively.

The subtractor 122 uses the d-axis current command value i^* _d= 0
And the above d-axis current detection value i_dD-axis current deviation which is the deviation from
Difference e_d= I^* _d-I_dAnd the d-axis current PI control unit 126 is calculated.
Is the d-axis current deviation e_dFor proportional integral control calculation for
Therefore, the d-axis voltage command value v^* _dAsk for. On the other hand, the subtracter 12
4 is the q-axis current command value i output from the adder 120 ^* _q
And the above q-axis current detection value i_qQ-axis current deviation which is the deviation from
Difference e_q= I^* _q-I_qIs calculated, and the q-axis current PI control unit 128 is calculated.
Is the q-axis current deviation e_qFor proportional integral control calculation for
Therefore, the q-axis voltage command value v^* _qAsk for.

The first limit processing unit 130 applies the voltage to the motor 6.
It is an amplitude limiting means for the phase voltage to be
The voltage command value for each phase is changed so that the voltage command value is always a sine wave.
Width v ^* _mV^* _m≤ E_d/ 2 (formula (7))
Unlike before, the above-mentioned d-axis voltage command value v^* _dAnd q-axis
Pressure command value v^* _qEach phase voltage command value (equation (6a) ~
(6c)) amplitude v^* _mIs v^* _m≤ E_dSo that
The above d-axis voltage command value v^* _dAnd q-axis voltage command value v^* _qTo
Restrict.

The dq / 3-phase AC coordinate conversion unit 132 is a first
1 of the above-mentioned restriction (limiter processing) by the restriction processing unit 130
D-axis voltage command value v after being applied^* _dAnd q-axis voltage command value
v^* _qIs the u-phase voltage command value v, which is the value on the three-phase AC coordinates^* _uOh
And v-phase voltage command value v^* _vConvert to. And sign inversion
The adder 134 receives those phase voltage command values v^* _uAnd v ^* _v
To w-phase voltage command value v^* _wTo calculate.

The second limit processing unit 136 is a limiter means for each phase voltage command value v ^* _u , v ^* _v , v ^* _w thus obtained, and each phase voltage command value v ^* _u , The phase voltage command values v ^* _u , v ^* _v , and v ^* _w are limited so that the amplitudes of v ^* _v and v ^* _w do not exceed _Ed / 2 when the neutral point is 0 ( _Ed is Power-supply voltage).

The three-phase PWM modulation section 150 responds to each phase voltage command value v ^* _u , v ^* _v , v ^* _w after the above-mentioned restriction (limiter processing) by the second restriction processing section 136. The duty ratio PWM signals Su, Sv, and Sw are generated.

The motor drive circuit 152 is, for example, a power M
A PWM voltage type inverter configured by using switching elements such as OS transistors, wherein each switching element is turned on / off by the PWM signals Su, Sv, Sw.
By turning off, the phase voltages v _u , v _v , v _w to be applied to the brushless motor 6 are generated. These respective phase voltages v _u , v _v , v _w are output from the ECU 5 and output to the motor 6
Applied to. In response to this voltage application, a current flows through the coils (not shown) of each phase u, v, w of the motor 6, and the motor 6
Generates a torque Tm for steering assistance according to the current.

Of the current flowing through the motor 6, the u-phase current i _u
And the v-phase current _iv are the u-phase current detector 156 as described above.
And v phase current detector 154 respectively detect 3
The phase alternating current / dq coordinate conversion unit 138 converts the current values i _d and i _q on the dq coordinates. Of the current values i _d and i _q on the dq coordinates, the d-axis current detection value i _d is the subtractor 12
2 and the q-axis current detection value i _q are input to the subtractor 124. As a result, the desired steering assist force is applied to the motor 6
Feedback control is performed so that the d-axis current detection value i _d becomes equal to the d-axis current command value i ^* _d and the q-axis current detection value i _q becomes equal to the q-axis current command value i ^* _q. (This control is called "current control").

<1.3 Current control operation> As described above
In the embodiment, appropriate steering compensation is performed according to the steering torque and the vehicle speed.
The target value of the motor current (d-axis current command
Value i^* _dAnd q-axis current command value i ^* _q) Is set, this eye
Feedback control is applied so that the standard current flows through the motor 6.
Control is done. Fig. 3 shows the matrix for such current control.
Calculation by Icon 10 (hereinafter referred to as "current control calculation")
3 is a flowchart showing the procedure of Microcomputer 10
By executing the current control calculation of
The mode of the current controller 20 corresponding to the part surrounded by the line is
Other than the drive unit (hereinafter referred to as "current control calculation unit")
Is realized. The book will be described below with reference to FIGS. 2 and 3.
Operation of the current control calculation unit in the embodiment (current control operation
Work) will be described.

In this embodiment, the ignition switch 9
When is turned on, the ECU 5 is activated, and the microcomputer 10 in the ECU 5 repeatedly executes the current control calculation shown in FIG. In this current control calculation, the microcomputer 10 operates as follows.

First, the microcomputer 10 receives the electrical angle θ _re indicating the magnetic pole position from the rotor angle position detector 162 (step S12), and the u-phase current detector 156 outputs the u-phase current detection value i _u to the v-phase current detection. The v-phase current detection value i _v is received from the device 154 (step S14). Next, to obtain a sine wave value sin [theta _re referring to sine wave ROM table 140 stored in the memory in the microcomputer 10, the sine wave value sin [theta _re and u-phase current detection value i _u and the v-phase current detection value Using i _v , the d-axis current detection value i _d and the q-axis current detection value i _q are calculated by the equations (3a) and (3b), that is, the following equation (step S16). The _{i d = √2 {i v sinθ} re -i u sin (θ re -2π / 3)} i q = √2 {i v cosθ re -i u cos (θ re -2π / 3)} This step S16 The three-phase AC / dq coordinate conversion unit 138 is realized.

[0042] Then the microcomputer 10 calculates the a deviation between the d-axis current detection value i _d and the preset d-axis current command value i ^* _d d-axis current deviation e _d = i ^* _d -i _d to, with obtaining the d axis voltage command value v ^* _d by the proportional integral control operations on the d-axis current deviation e _d, the q-axis current detection value i _q to be obtained by the adder 120 q-axis current command value i ^* is calculated which is the deviation between the _q q-axis current deviation ^{_{e q = i * q -i q}} , obtains the q-axis voltage command value v ^* _q by the proportional integral control operations on the q-axis current deviation e _q. That is, d
The axis voltage command value v ^* _d and the q-axis voltage command value v ^* _q are calculated (steps S18 and S20). ^{_{v * d = K p {e}} d + (1 / T i) ∫e d dt} v * q = K p {e q + (1 / T i) ∫e q dt} where, Kp is a proportional gain Yes, Ti is the integration time. By the above step S18, the subtracters 122 and 12
4 is realized, and the d-axis current PI control unit 126 and the q-axis current PI control unit 128 are realized by step S20.

Next, the microcomputer 10 controls the amplitude v of each phase voltage command value (see the equations (6a) to (6c)) corresponding to the d-axis voltage command value v ^* _d and the q-axis voltage command value v ^* _q. ^The d-axis voltage command value v ^* _d and the q-axis voltage command value v ^* _q are limited so that ^* _m becomes v ^* _m ≦ _Ed (10). Specifically, the formula (6d), as shown in _{^{v * m = √ (2/3)}} √ {v * d 2 + (- v * q) 2} because it is, the above equation based on the equation (10 The d-axis voltage command value v ^* _d and the q-axis voltage command value v ^* _q are limited so as to satisfy () (step S22). The first limit processing unit 130 is realized by the limiter processing that is the processing of step S22.

Next, the microcomputer 10 uses the sine wave ROM table.
Sine wave value sin θ obtained by referencing rule 140_reAnd the above
D-axis voltage command value v after mitter processing^* _dAnd q-axis voltage command
Value v^* _qUsing and, the voltage command value v for each phase^* _u, V^* _v, V^* _w
Is calculated by the following equation (step S24). v^* _u= √ (2/3) {v^* _dcos θ_re-V^* _qsin θ_re} v^* _v= √ (2/3) {v^* _dcos (θ_re−2π / 3) −v^* _qsin (θ_re
−2π / 3)} v^* _w= -V^* _u-V^* _v By this step S24, dq / 3-phase AC coordinate conversion
The unit 132 and the sign inversion adder 134 are realized.

Next, the microcomputer 10 obtains in this way
Each phase voltage command value v^* _u, V^* _v, V ^* _wAgainst limiter
Processing is performed (step S26). That is, each phase voltage finger
Command v^* _u, V^* _v, V^* _wWhere, when the neutral point is 0,
To limit. -E_d/ 2≤v^* _u≤ E_d/ 2… (11a) -E_d/ 2≤v^* _v≤ E_d/ 2… (11b) -E_d/ 2≤v^* _w≤ E_d/ 2… (11c) Where E_dIs the power supply voltage. To this step S26
As a result, the second restriction processing unit 136 is realized. The above formula (1
Each phase voltage finger after the limiter processing by 1a) to (11c)
Command v^* _u, V^* _v, V^* _wIs output from the microcomputer 10 (
(Step S28) One current control calculation ends.

As described above, the above-mentioned current control calculation is repeatedly executed, and the motor 6 is driven by the motor drive unit including the three-phase PWM modulation unit 150 and the motor drive circuit 152.
Each phase voltage command value v ^* _u output from the microcomputer 10,
_v ^* v, v ^* phase voltage according to the _{w v u, v v, v} w is applied.

<1.4 Operation and Effect> The above-mentioned first example
According to the embodiment, the motor 6 is in a high rotation and high load state.
In this case, each phase voltage command value becomes trapezoidal. Sanawa
Then, when the rotation speed and the load of the motor 6 become a certain amount or more,
Each phase voltage finger before the limiter processing in the second limit processing unit 136
Command v^* _u, V^* _v, V^* _wIs set by the first restriction processing unit 130.
Due to the width limitation (equation (10)), as shown in FIG.
Amplitude v^* _m= E_dOf the sine wave of the second limiting processing unit 13
After the limiter process (equations (11a) to (11c)) in 6
Each phase voltage command value v ^* _u, V^* _v, V^* _wIs shown in FIG.
It becomes a trapezoidal wavy shape. As a result, conventional (Fig. 9)
Compared with this, the motor output increases, so a large steering assist force
can get. In this case, the interphase voltage (line voltage) v^*
_uv= V^* _u-V^* _v, V^* _vw= V^* _v-V^* _w, V^* _wu= V^* _w-V
^* _uAlso has a trapezoidal wave shape as shown in FIG. Sanawa
The maximum value of the line voltage amplitude is E_dAnd compared to conventional
The voltage utilization rate is improved, and the power supply for driving the motor 6 is increased.
Maximum pressure can be used. By the way, each phase
Voltage command value v^* _u, V^* _v, V^* _wIs trapezoidal wavy as above
If the voltage command values of different phases are the same,
If there is a period of time, the current flowing through the motor 6
Therefore, there will be a period in which the magnetic field does not rotate. However
In the present embodiment, the first restriction processing unit 130 causes v^* _m≤
E_dAmplitude v so that (Equation (10))^* _mIs restricted
Therefore, as shown in Fig. 4 (b),
There is no period when the pressure command values are the same.

Although the first embodiment is premised on the use of the three-phase brushless motor 6, even when a brushless motor having a number of phases other than three is used, the same effect can be obtained by the same configuration as above. Is obtained.

<2. Second Embodiment> In the first embodiment, the power supply voltage can be utilized to the maximum when driving the brushless motor 6, and a large steering assist force can be obtained. However, since the interphase voltage (line voltage) applied to the motor 6 at the time of high rotation and high load of the motor 6 has a trapezoidal waveform (FIG. 4C), the motor current includes harmonic components, This causes the generation and increase of torque ripple and vibration. Therefore, hereinafter, an electric power steering apparatus in which the voltage utilization rate in driving the brushless motor 6 is improved while suppressing the occurrence of torque ripple and vibration will be described as a second embodiment of the present invention.

The overall structure of the electric power steering apparatus according to this embodiment is the same as that of the first embodiment and is as shown in FIG. Below, it demonstrates centering around the structure and operation | movement of ECU5 which is a control apparatus in this embodiment.

FIG. 5 is a block diagram showing the configuration of the ECU 5 in this embodiment. The same parts of the configuration of the ECU 5 as those of the first embodiment are designated by the same reference numerals, and detailed description thereof will be omitted. Also in the present embodiment, the ECU 5
Is composed of a phase compensator 112, a microcomputer 10, and a motor drive unit, but the configuration for current control differs from that of the first embodiment. That is, in the present embodiment, the motor control unit realized by the microcomputer 10 executing a predetermined program has the same components as those in the first embodiment.
Third harmonic component calculator 202 and adders 221, 222
In addition to the above, the sign inversion adder 134 is replaced with an adder / subtractor 224. In addition, the first restriction processing unit 2
The content of the limiter processing by 30 is also different from that of the first restriction processing unit 130 in the first embodiment.

<2.1 Configuration of Current Control Calculation Unit>
In the present embodiment, the current control arithmetic unit, which is a part that constitutes the current control unit 30 in the motor control unit realized by the microcomputer 10, will be described focusing on the differences from the first embodiment.

In this embodiment, in order to secure the amplitude of the interphase voltage (line voltage) to be applied to the motor 6 up to the power supply voltage E _d , the phase voltage command value v ^* shown in the equations (6a) to (6c) ^. _u ,
The third harmonic is superimposed on the fundamental sine wave of v ^* _v and v ^* _w . That is, each phase voltage command value v ^* _u , v shown by the following equation
^{Use *} _v , v ^* _w . ^{_{v * u = -v * m {}} sin (θ re + φ) + (1/6) sin3 (θ re + φ)} ... (12a) v * v = -v * m {sin (θ re -2π / 3 + φ) + (1/6) sin3 (θ _re −2π / 3 + φ)} (12b) v ^* _w = −v ^* _m {sin (θ _re −4π / 3 + φ) + (1/6) sin3 (θ _re −4π) / 3 + φ)} ... ( 12c) v * m = √ (2/3) √ {v * d 2 + (- v * q) 2} ... (12d) where, v ^* when _q ≠ 0 _φ = tan ⁻¹ {v ^* _d / (− v ^* _q )}, and φ = −π / 2 when v ^* _q = 0. Also, sin3
(θ _re + φ) = sin3 (θ _re −2π / 3 + φ) = sin3 (θ _re −4π /
3 + φ), the third harmonic does not appear in the line voltage (phase voltage).

In order to prevent the respective phase voltage command values v ^* _u , v ^* _v , v ^* _w represented by the above equations (12a) to (12c) from being distorted, v ^* _m {sinθ _re + (1 / 6) sin3θ _re } ≦ E _d / 2. Incidentally, the maximum value of sinθ _re + (1/6) sin3θ _re when changing the rotor angular position, that is the electrical angle theta _re is √3 / 2. Therefore, (√3 / 2) v ^* _m ≤ E _d /
2, that is, v ^* _m ≦ E _d / √3 (13). As can be seen by comparing the above equation (13) with the equation (7), this means that the limiting value of the amplitude of the basic sine wave in the phase voltage command value becomes 2 / √3 times the conventional value. In the above formulas (12a) to (12c), the amplitude of the third harmonic to be superimposed on the basic sine wave is 1/6 of the amplitude of the basic sine wave. This is because the peak value of the combined wave of the third harmonic and the third harmonic becomes the smallest.

The current control calculation unit in this embodiment has a configuration different from that of the first embodiment as follows based on the above principle.

As the amplitude limiting means in this embodiment
The first restriction processing unit 230 uses the above equation (13), that is, v^* _m
≤ E_dD-axis voltage command value v to satisfy / √3^* _dAnd q
Axis voltage command value v^* _qTo limit. In this respect, the present embodiment is
Formula (10) or v^* _m≤ E _dD-axis voltage to meet
Command value v^* _dAnd q-axis voltage command value v^* _qThe first to limit
It differs from the embodiment.

Further, the third harmonic generation in this embodiment is performed.
The minute calculation unit 202 determines each phase voltage command value v^* _u, V^* _v, V^* _wTo
Third harmonic component v to be superimposed on the fundamental wave component in
^* ₃Amplitude restriction by the first restriction processing unit 230 (equation (1
D-axis voltage command value v after processing 3)) ^* _dAnd q-axis voltage finger
Command v^* _qCalculate from Then, the adder 221 outputs dq
/ U-phase voltage obtained by the three-phase AC coordinate conversion unit 132
The third harmonic component v in the command value (fundamental wave component)^* ₃Add
The adder 222 calculates the dq / 3-phase AC coordinate conversion unit 1
To the v-phase voltage command value (fundamental wave component) obtained by 32
The third harmonic component v^* ₃Is added. Also, adder / subtractor
224 is the dq / 3-phase AC coordinate conversion unit 132.
Sign inversion of the obtained u-phase voltage command value and v-phase voltage command value
Then, the third harmonic component v is added to the added value.^* ₃
Is added. In this way, equations (12a) to (12
As shown in c), superposition of the 3rd harmonic component on the fundamental wave component
Each phase voltage command value v^* _u, V^* _v, V^* _wIs obtained. Na
The 3rd harmonic component is the 3rd harmonic component as described above.
Calculated by the calculation unit 202, but calculates the fundamental wave component
The means for performing the subtraction are 122, 124 and d-axis current PI control.
Section 126 and q-axis current PI control section 128 and dq / 3 phase crossing
By the flow coordinate conversion unit 132 and a part of the adder / subtractor 224,
Has been realized.

The second limit processing unit 136 determines that the absolute value of each phase voltage command value v ^* _u , v ^* _v , v ^* _w thus obtained is E.
Each phase voltage command value v ^* _u , v ^* _v , v ^* _w so as not to exceed _d / 2
A limiter process is applied to (E _d is the power supply voltage). However, in the present embodiment, the first restriction processing unit 230 causes the expression (1
Since the amplitude of the basic sine wave of each phase voltage command value is limited so as to satisfy 3), each phase voltage command value v ^* _u , v ^* _v , as long as the current control calculation unit operates normally. The absolute value of v ^* _w never exceeds E _d / 2.

The phase voltage command values v ^* _u , v ^* _v , v ^* _w after the limiter processing by the second limit processing section 136 are output from the current control calculation section (microcomputer 10). Motor drive unit comprising a like 3-phase PWM modulation portion 150 and the motor driving circuit 152, these phase voltage command values _{^{v * u, v * v,}} v * phase voltage corresponding to _{w v u, v v, v} w Is applied to the motor 6, which drives the motor 6.

<2.2 Current Control Operation> FIG. 6 is a flow chart showing the procedure of current control calculation by the microcomputer 10 in the present embodiment. When the microcomputer 10 executes this current control calculation, the part other than the motor drive part of the current control part 30 corresponding to the part surrounded by the dotted line in FIG. 5, that is, the current control calculation part is realized. Hereinafter, the operation (current control operation) of the current control calculation unit in this embodiment will be described with reference to FIGS. 5 and 6.

Also in the present embodiment, when the ECU 5 is activated, the microcomputer 10 in the ECU 5 repeatedly executes the current control calculation shown in FIG. In this current control calculation, the microcomputer 10 operates as follows.

The current control calculation routine shown in FIG. 6 is called.
Then, the microcomputer 10 is similar to that of the first embodiment.
Steps S12 to S20 are executed. This makes 3
Phase AC / dq coordinate conversion unit 138, subtractor 122 and
And 124, the d-axis current PI control unit 126, and the q-axis current P
The I control unit 128 is realized. These components are
Having similar functionality to the corresponding components of one embodiment,
From these, the d-axis voltage command value v represented by the following equation^* _dAnd
And q-axis voltage command value v^* _qIs obtained. v^* _d= K_p{E_d+ (1 / T_i) ∫e_ddt} v^* _q= K_p{E_q+ (1 / T_i) ∫e_qdt} e_d= I^* _d-I_d, E_q= I^* _q-I_q Here, Kp is a proportional gain, Ti is an integration time, and i
^* _dIs the d-axis current command value, i _dIs the d-axis current detection value, i^*
_qIs the q-axis current command value, i_qIs the q-axis current detection value.

After that, the microcomputer 10 determines the above-mentioned d-axis voltage.
Command value v^* _dAnd q-axis voltage command value v ^* _qEach phase corresponding to
Amplitude v of pressure command value^* _mIs equation (13) v^* _m≤ E_d/ √3 D-axis voltage command value v^* _dAnd q-axis voltage finger
Command v^* _qTo limit. Specifically, it is shown in equation (6d).
Sea urchin v^* _m= √ (2/3) √ {v^* _d ²+ (-V^* _q)²} Therefore, based on this equation, satisfy equation (13)
d-axis voltage command value v^* _dAnd q-axis voltage command value v^* _qLimit
(Step S32). Limiter of this step S32
The first restriction processing unit 230 is realized by the processing.

Next, the microcomputer 10 sets the d-axis voltage command value v ^* _d and the q-axis voltage command value v ^* _q after the limiter processing on the three-phase AC coordinates, that is, in the same manner as in the first embodiment. It is converted into each phase voltage command value (step S24). That is, the phase voltage command value consisting of only the fundamental wave component (hereinafter referred to as “fundamental wave phase voltage command value”) v ^* _uf , v ^* _vf ,
Calculate v ^* _wf . v ^* _uf = √ (2/3) {v ^* _d cos θ _re −v ^* _q sin θ _re } v ^* _vf = √ (2/3) {v ^* _d cos (θ _re −2π / 3) −v ^* _q sin (θ _re
−2π / 3)} v ^* _wf = −v ^* _uf− v ^* _vf

Next, the microcomputer 10 calculates the _third harmonic component v ^* ₃ from the d-axis voltage command value v ^* _d and the q-axis voltage command value v ^* _q after the above limiter processing by the following equation ( Step S34). _{^{v * 3 = (- 1/6)}} √ (2/3) √ {v * d 2 + (- v * q) 2} sin3 (θ re + φ) ... (16) where, v of ^* _q ≠ 0 Then, φ = tan ⁻¹ {v ^* _d / (− v ^* _q )}, and when v ^* _q = 0, φ = −π / 2.

Then, the microcomputer 10 calculates the above-mentioned _third harmonic component v ^* ₃ as the above-mentioned fundamental wave phase voltage command values v ^* _uf ,
By adding to v ^* _vf and v ^* _wf , that is, v ^* _u = v ^* _uf + v ^* ₃ v ^* _v = v ^* _vf + v ^* ₃ v ^* _w = v ^* _wf + v ^* ₃ , equation (12a) ~ (12c) to obtain each phase voltage command value v ^* _u , v ^* _v , v ^* _w (step S36).

The above steps S24, S34,
By S36, the dq / 3-phase AC coordinate converter 132, the third harmonic component calculator 202, the adders 221, 222,
And the adder / subtractor 224 is realized.

However, since the calculation of the square root and the arc tangent function included in the above equation (16) requires a large computing ability, the above equation (16) is used as it is to calculate the _third harmonic component v ^* ₃ . Is difficult to do. Therefore, in the present embodiment, the formula (16) is modified as follows using a trigonometric formula. _{^{v * 3 = - (1/6)}} √ (2/3) √ {v * d 2 + (- v * q) 2} sin3 (θ re + φ) = - (1/6) √ (2/3) ^{_{√ {v * d 2 + (}} - v * q) 2} × {sin (3θ re + φ) cos2φ + cos (3θ re + φ) sin2φ} ... (17) ^{where, φ = tan -1 {v *} d / (- v ^* from ^{_{q)}, sinφ = v *}} d / √ {v * d 2 + (- v * q) 2} ... (18a) cosφ = -v * q / √ {v * d 2 + (- v * since it is ^{_{q) 2} ... (18b)}} , sin2φ = 2sinφcosφ = -2v * d v * q / (v * d 2 + v * q 2) ... (19a) cos2φ = 1-2sin 2 φ = (- v * ^{_{d 2 + v * q 2)}} / (v * d 2 + v * q 2) ... a (19b). ^{_{Further, -√ {v * d 2 +}} (- v * q) 2} sin (3θ re + φ) = - v * q sin3θ re + v * d cos3θ re ... (20a) -√ {v * d 2 + (- v ^* _q ) ² } cos (3θ _re + φ) = − v ^* _q cos3θ _re −v ^* _d sin3θ _re (20b). Therefore, equations (19a) (19b) (20
By substituting a) (20b) into equation (17), the following equation is obtained. _{^{v * 3 = - (1/6)}} √ (2/3) [{- v * q + 4v * d 2 v * q / (v * d 2 + v * q 2)} sin3θ re + {- v * d + 4v ^{_{* d v * q 2 / (}} v * d 2 + v * q 2)} cos3θ re] ... (21) from the above equation, the reference of the third harmonic component v ^* _3, for example, four arithmetic operations and the sine wave ROM table Can be sought by. Therefore, it is possible to reduce the calculation load of the microcomputer 10 due to the calculation of the third harmonic component v ^* ₃ . In the present embodiment, the third-order harmonic component calculation unit 202 realized in step S34 uses the above equation (21).
Is used to calculate the _third harmonic component v ^* ₃ . Note that v ^* _d
= V when the ^* _q = 0 is v ^* ₃ = 0.

As described above, the third harmonic component v^* ₃But
Is calculated and its third harmonic component v ^* ₃Of each phase
Voltage command value v^* _u, V^* _v, V^* _wIs obtained, the microcomputer 1
0 indicates each phase voltage command value v^* _u, V^* _v, V^* _wAgainst
Limiter processing similar to that of the first embodiment is performed (step S
26). That is, each phase voltage command value v^* _u, V^* _v, V^* _wTo
Limit as in the following formula. -E_d/ 2≤v^* _u≤ E_d/ 2 -E_d/ 2≤v^* _v≤ E_d/ 2 -E_d/ 2≤v^* _w≤ E_d/ 2 Where E_dIs the power supply voltage. To this step S26
As a result, the second restriction processing unit 136 is realized. This second system
Each phase voltage command value after limiter processing by the limit processing unit 136
v^* _u, V^* _v, V^* _wIs output from the microcomputer 10 (step
(S28) The current control calculation for one time ends.

As described above, the above current control calculation is repeatedly executed, and the motor 6 is driven by the motor drive unit including the three-phase PWM modulation unit 150 and the motor drive circuit 152.
Each phase voltage command value v ^* _u output from the microcomputer 10,
_v ^* v, v ^* phase voltage according to the _{w v u, v v, v} w is applied.

<2.3 Actions and Effects> According to the second embodiment described above, the limitation on the amplitude v ^* _m of the fundamental sine wave at each phase voltage command value v ^* _u , v ^* _v , v ^* _w is v. ^* _m ≤ E _d / √
3 (equation (13)), the limit value is 2 / √3 times as large as the conventional value (equation (7)). Then, when the motor 6 is in a high-rotation and high-load state and the amplitude v ^* _m of the basic sine wave is maximum, the phase voltage command values v ^* _u , v ^* _v , and v ^* _w are _calculated by the equation (1)
2a) ~ from ^{(12c) (13) v *} u = - (E d / √3) {sin (θ re + φ) + (1/6) sin3 (θ re + φ)} v * v = - (E d / √3) {sin (θ _re −2π / 3 + φ) + (1/6) sin3 (θ _re −2π / 3 + φ)} = − (E _d / √3) {sin (θ _re −2π / 3 + φ) + (1/6) sin3 (θ _re + φ)} v ^* _w = − (E _d / √3) {sin (θ _re −4π / 3 + φ) + (1/6) sin3 (θ _re −4π / 3 + φ)} = − (E _d / √3) {sin (θ _re −4π / 3 + φ) + (1/6) sin3 (θ _re + φ)}, so the interphase voltage of each phase u, v, w (line voltage ) V
^* _uv , v ^* _vw , v ^* _wu are v ^* _uv = v ^* _u− v ^* _v = −E _d sin (θ _re + π / 6 + φ) (15a) v ^* _vw = v ^* _v− v ^* _w = −E _d sin (θ _re −π / 2 + φ) (15b) v ^* _wu = v ^* _w −v ^* _u = −E _d sin (θ _re + 5π / 6 + φ) (15c), and the interphase voltage v The amplitudes of ^* _uv , v ^* _vw , v ^* _wu are equal to the power supply voltage E _d . Therefore, according to this embodiment, the power supply voltage can be utilized to the maximum extent. However, the motor output in the present embodiment is larger than the conventional one, but is lower and smaller than that in the first embodiment (see FIG. 4).

Further, according to the present embodiment, the interphase voltages (line voltages) v ^* _uv , v ^* _vw , v ^* _wu only include the fundamental wave component and do not include the harmonic wave component (equations (15a) to (15a) 15
c)). As a result, the harmonic components contained in the motor current are suppressed, so that the motor output can be increased more than before without increasing the torque ripple and the vibration.

<2.4 Modification> As described above, when the amplitude of the third harmonic component to be superimposed in each phase voltage command value becomes 1/6 of the amplitude of the fundamental wave component, the phase voltage The limit value of the amplitude of the basic sine wave in the command value is the largest (the limit to the amplitude is the loosest), but the amplitude of the third harmonic to be superimposed is not necessarily 1/6 of the amplitude of the basic sine wave. There is no. For example, even if the amplitude of the third harmonic to be superimposed is 1/3 or 1/4 of the amplitude of the basic sine wave, the utilization rate of the power supply voltage in driving the brushless motor is improved as compared with the conventional case.

In the second embodiment described above, the third harmonic component is superimposed on the fundamental wave component of each phase voltage command value. However, if the third harmonic component is a multiple of 3 and an odd harmonic component, other The harmonic component of may be superimposed on the fundamental component. However, considering the feasibility of the device and the effects obtained, it is practical to superimpose only the third harmonic component.

Further, although the three-phase brushless motor is used in the second embodiment, generally, when the n-phase brushless motor is used, the nth harmonic is superimposed on the basic sine wave. By doing so, the same effect can be obtained.

[Brief description of drawings]

FIG. 1 is a schematic diagram showing a configuration of an electric power steering apparatus according to a first embodiment of the present invention together with a vehicle configuration related thereto.

FIG. 2 is a block diagram showing a configuration of an ECU that is a control device in the electric power steering device according to the first embodiment.

FIG. 3 is a flowchart showing a current control calculation executed by a microcomputer in the first embodiment.

FIG. 4 is a voltage waveform diagram showing a phase voltage command value for controlling the brushless motor in the first embodiment and an interphase voltage corresponding to the phase voltage command value.

FIG. 5 is a block diagram showing a configuration of an ECU (control device) in an electric power steering device according to a second embodiment of the present invention.

FIG. 6 is a flowchart showing a current control calculation executed by a microcomputer in the second embodiment.

FIG. 7 is a voltage waveform diagram showing a phase voltage command value for controlling the brushless motor in the second embodiment and an interphase voltage corresponding to the phase voltage command value.

FIG. 8 is a schematic diagram for explaining three-phase AC coordinates and dq coordinates in a three-phase brushless motor.

FIG. 9 is a voltage waveform diagram showing a phase voltage command value for controlling a conventional brushless motor and an interphase voltage corresponding to the phase voltage command value.

[Explanation of symbols]

3 ... Torque sensor 4 ... Vehicle speed sensor 5 ... Electronic control unit (ECU) 6 ... Brushless motor 8 ... Battery 11 ... Ball screw drive unit 12 ... Position detection sensor 10 ... Microcomputer (motor control unit, control calculation means) 20, 30 Current control unit 114 Target current calculation units 122 and 124 Subtractor 126 d axis current PI control unit 128 q axis current PI control units 130 and 230 First limit processing unit (amplitude limiting means) 132 ... d- q / 3-phase AC coordinate conversion unit 136 ... Second limit processing unit (limiter means) 138 ... 3-phase AC / dq coordinate conversion unit 150 ... 3-phase PWM modulation unit 152 ... Motor drive circuit 202 ... Third harmonic wave component calculating unit Ts ... steering torque signal i ^* _d ... d-axis current command value i ^* _q ... q-axis current command value i _d ... d-axis current detection value i _q ... q-axis current detection value e _d ... d-axis Flow deviation e _q ... q-axis current deviation v ^* _d ... d-axis voltage command value v ^* _q ... q-axis voltage command value v ^* _u ... u-phase voltage command value v ^* _v ... v-phase voltage command value v ^* _w ... w Phase voltage command value θ _re … electrical angle (rotor angle position)

Front page continuation (51) Int.Cl. ⁷ Identification code FI theme code (reference) B62D 137: 00 H02P 6/02 351A F term (reference) 3D032 CC50 DA15 DA23 DA63 DA64 DC01 DC02 DC17 DC29 DC34 DD02 DD06 DD10 DD17 EA01 EB11 EC23 GG01 3D033 CA13 CA16 CA20 CA21 5H560 AA08 BB04 BB12 DA02 DB20 DC03 DC12 DC13 EB01 RR04 SS02 TT15 XA02 XA04 XA11 XA12 XA13

Claims (5)

[Claims] 1. An electric motor for applying a steering assist force to a steering mechanism of a vehicle by applying a phase voltage to the brushless motor to drive the brushless motor according to a steering torque applied to an operating means for steering the vehicle. A power steering device, which represents a control calculation means for calculating a value of each phase voltage to be applied to the brushless motor as a sinusoidal phase voltage command value, and each phase voltage command value calculated by the control calculation means Amplitude limiting means for limiting the amplitude of the sine wave to a power supply voltage E _d or less for driving the brushless motor, and the phase voltages whose amplitude is limited by the amplitude limiting means based on the power supply voltage E _d Li performing command value, the limiter process of limiting the E _d / 2 or less in the range of neutral point -E _d / 2 or more when a 0 to the phase voltage command values Jitter means and an electric power steering apparatus characterized by comprising a motor driving means for applying to the brushless motor phase voltages corresponding to the phase voltage command value after the limiter process. 2. An electric motor for applying a steering assist force to a steering mechanism of a vehicle by applying a phase voltage to the brushless motor to drive the brushless motor according to a steering torque applied to an operating means for steering the vehicle. A power steering apparatus, wherein a control calculation unit calculates a value of each phase voltage to be applied to the brushless motor as a phase voltage command value of a waveform obtained by superimposing a third harmonic on a basic sine wave; An electric power steering apparatus, comprising: a motor drive unit that applies each phase voltage according to each phase voltage command value calculated by the unit to the brushless motor. 3. The fundamental wave component arithmetic means for calculating a fundamental wave component value in each phase voltage to be applied to the brushless motor as a fundamental wave phase voltage command value, and the fundamental wave phase voltage command. A harmonic component calculation means for calculating a third-order harmonic component that is a harmonic component of a sine wave representing a value and has an amplitude of 1/6 of the amplitude of the sine wave; The electric power steering apparatus according to claim 2, further comprising: an addition unit that calculates each phase voltage command value by adding a third harmonic component. 4. The fundamental wave component computing means introduces dq coordinates, which is a rotating coordinate system composed of a d-axis which is a magnetic flux direction of a rotating field in the brushless motor and a q-axis which is orthogonal to the d-axis. Then, the d-axis voltage command value and the q-axis voltage command value, which represent the value of the voltage to be applied to the brushless motor by the dq coordinates, flow through the brushless motor and the target current value determined according to the steering torque. D-axis and q-axis current control calculation means for calculating based on the deviation from the detected current value, and converting the d-axis voltage command value and the q-axis voltage command value into voltage values for each phase of the brushless motor Coordinate conversion means for generating the fundamental wave phase voltage command value is included, and the harmonic component calculation means calculates the third harmonic component v ^* ₃ by the following equation: Three
The electric power steering apparatus according _{^{to: v * 3 = - (1/6}} ) √ (2/3) [{- v * q + 4v * d 2 v * q / (v * d 2
^{_{+ V * q 2)} sin3θ}} re + {- v * d + 4v * d v * q 2 / (v * d 2 +
^{_{v * q 2)} cos3θ re}} ] Here, v ^* _d is the d-axis voltage command value, v ^* _q is the q-axis voltage command value, theta _re indicates the rotational position of the rotating field magnet It is an electrical angle. 5. The control calculation means further includes amplitude limiting means for limiting the amplitude of the sine wave representing the fundamental wave phase voltage command value to 1 / √3 or less of a power supply voltage for driving the brushless motor. The electric power steering apparatus according to claim 3, further comprising: