JP2003299356A - Dc-dc converter control method - Google Patents

Dc-dc converter control method

Info

Publication number
JP2003299356A
JP2003299356A JP2002098482A JP2002098482A JP2003299356A JP 2003299356 A JP2003299356 A JP 2003299356A JP 2002098482 A JP2002098482 A JP 2002098482A JP 2002098482 A JP2002098482 A JP 2002098482A JP 2003299356 A JP2003299356 A JP 2003299356A
Authority
JP
Grant status
Application
Patent type
Prior art keywords
dc
voltage
output
circuit
dc power
Prior art date
Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
Pending
Application number
JP2002098482A
Other languages
Japanese (ja)
Inventor
Kesanobu Kuwabara
Kiyoshi Moriya
Takashi Sonoda
崇 園田
今朝信 桑原
清志 森谷
Original Assignee
Fuji Electric Co Ltd
Nanao Corp
富士電機株式会社
株式会社ナナオ
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
Filing date
Publication date

Links

Classifications

    • YGENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02BCLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
    • Y02B70/00Technologies for an efficient end-user side electric power management and consumption
    • Y02B70/10Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion
    • Y02B70/16Efficient standby or energy saving modes, e.g. detecting absence of load or auto-off

Abstract

PROBLEM TO BE SOLVED: To provide a control method that attenuates the abnormal noise of a transformer without causing power consumption at a light load, because, if a DC power source voltage is higher than usual when switching elements are changed over in a DC-DC (direct current-direct current) converter, a rush current that flows into the transformer causes harsh grating noise.
SOLUTION: This DC-DC converter is the one that converts a DC power source voltage into a DC output of a constant voltage by on-off operations of switching elements, and that oscillates the switching elements intermittently by providing an oscillating period and a forced stop period at the time of a light load including no load. In this converter, by controlling the output of a comparator that compares the output of a function voltage generating circuit including a triangular-wave voltage with a command value according to the DC power source voltage, the on-range width of the switching elements is made to change so that the current that flows into the transformer does not increase as the DC power source voltage increases.
COPYRIGHT: (C)2004,JPO

Description

【発明の詳細な説明】 【0001】 【発明の属する技術分野】この発明は、直流電源の出力を、変圧器を介して任意の直流出力に変換するDC/D BACKGROUND OF THE INVENTION [0001] [Technical Field of the Invention The present invention, the output of the DC power source, via a transformer to convert any DC output DC / D
Cコンバータの制御方法、特に無負荷を含む軽負荷時における制御方法の改良に関する。 The method of C converter, to an improved control method in particular light load, including no-load. 【0002】 【従来の技術】図8に従来例の回路接続図を示す。 [0002] illustrates a circuit connection diagram of a conventional example of the Related Art FIG. 図8 Figure 8
の回路はフライバックコンバータの例で、直流電源1、 The circuit of the example of the fly-back converter, the DC power source 1,
スイッチ素子2、フライバックトランス3、整流器4及びコンデンサ5から電力回路が構成される。 Switching element 2, the fly-back transformer 3, the power circuit is composed of the rectifier 4 and capacitor 5. コンデンサ5の出力電圧Voは、出力電圧検出調整回路6の入力段である差動増幅器61に設定電圧Vrefと共に与えられ、その偏差に相当する信号がフォトカプラ62を介して指令値Vcとして出力される。 The output voltage Vo of the capacitor 5 is provided with a set voltage Vref in the differential amplifier 61 is the input stage of the output voltage detection adjusting circuit 6, a signal corresponding to the deviation is outputted as a command value Vc via the photocoupler 62 that. この指令値Vcは、コンパレータ7において関数電圧発生回路8の出力Vtと比較され、PWM(パルス幅変調)信号Vmを発生する。 The command value Vc is the comparator 7 is compared with the output Vt of the function voltage generating circuit 8, PWM (pulse width modulation) to generate the signal Vm. 関数電圧発生回路8は、三角波(鋸歯状波を含む) Function voltage generating circuit 8, a triangular wave (including sawtooth)
出力電圧を発生するものである。 It is intended to generate an output voltage. 【0003】このPWM信号Vmと、固定周波数かつ固定デューティ比の矩形波信号発生回路10の出力Vsとは、アンドゲート11において論理積をとられ,ゲート信号Vgとなってゲートドライブ回路9を介してスイッチ素子2をオンオフさせる。 [0003] and the PWM signal Vm, and the output Vs of the square wave signal generating circuit 10 of fixed frequency and fixed duty ratio, the AND gate 11 is ANDed, via a gate drive circuit 9 becomes the gate signal Vg turning on and off the switching element 2 Te. 【0004】図9は図8の従来回路の動作波形を示すもので、(1)は出力指令値Vcと関数電圧発生回路8の出力Vtとの関係を示しており、(2)は矩形波信号発生回路10の出力VsのLレベル領域を中心に描いたものである。 [0004] Figure 9 shows the operation waveforms of the conventional circuit of FIG. 8, (1) shows the relationship between the output Vt of the output command value Vc and functions voltage generating circuit 8, (2) the square wave those drawn around the L level region of the output Vs of the signal generating circuit 10. (3)に示すように、コンパレータ7の出力であるPWM信号Vmは、(1)の波形図で指令値Vc As shown in (3), PWM signal Vm is the output of the comparator 7, the command value Vc waveform diagram of (1)
が関数電圧発生回路8の出力電圧Vtを上回っている期間中、Hレベルとなるが、PWM信号Vmは矩形波信号発生回路10の出力VsがHレベルのときにのみアンドゲート11を通過できるので、アンドゲート11の出力信号Vgは(4)で示すような波形となる。 There during the period exceeds the output voltage Vt of the function voltage generating circuit 8, becomes a H level, the PWM signal Vm can pass only the AND gate 11 when the output Vs of the square wave signal generating circuit 10 is at H level , the output signal Vg of the aND gate 11 has a waveform as shown by (4). 【0005】このように、図9に示す従来回路は、矩形波信号発生回路10の出力VsがHレベルのときにスイッチ素子2がスイッチングを繰り返す発振期間となり、 [0005] Thus, the conventional circuit shown in FIG. 9, the switch element 2 output Vs of the square wave signal generating circuit 10 is at the H level becomes an oscillation period repeatedly switching,
出力VsがLレベルのときにスイッチ素子2がスイッチングを強制停止される強制停止期間となるようにして、 Output Vs is the switching element 2 when the L level as the suspended period is forcibly stops switching,
ゲートドライブ回路9を介してスイッチ素子2を間欠発振動作させるものである。 It is intended to intermittent oscillation operation of the switching element 2 via the gate drive circuit 9. 【0006】そして、上述の従来回路によれば、強制停止期間を設けることで単位時間当りのスイッチング回数を減少させ、スイッチング損失や導通損失を低減させることができる。 [0006] Then, according to the conventional circuit described above, it reduces the number of switching per unit time in providing a forced stop period, it is possible to reduce the switching losses and conduction losses. ただし、強制停止期間にはフライバックコンバータの出力にエネルギが供給されないことから、 However, since the energy is not supplied to the output of the flyback converter is suspended for the period,
出力電圧が多少脈動するため、軽負荷時に想定した最大負荷時における出力電圧の脈動を許容値内に抑えるよう、発振期間と強制停止期間を最適に設定することが必要である。 Since the output voltage is somewhat pulsating, to suppress the pulsation of the output voltage at the maximum load was assumed at the time of light load within the allowable value, it is necessary to optimally set the forced stop period and the oscillation period. 【0007】 【発明が解決しようとする課題】ところで、図8に示す従来例では、強制停止期間と発振期間の切り替わりのタイミングに、スイッチ素子のオン幅が急激に変化し、直流電源1の電圧が高い場合にはトランスに大電流が急激に流れて電気的なストレスが加わるため、矩形波信号発生回路10の発振周波数が可聴帯域内である場合には、 [0007] [SUMMARY OF THE INVENTION Incidentally, in the conventional example shown in FIG. 8, the timing of switching of the suspended period oscillation period, the ON width of the switching element changes rapidly, the DC power supply 1 voltage since the applied electrical stress by a large current flows rapidly in the transformer is higher, when the oscillation frequency of the square wave signal generating circuit 10 is within the audible band,
トランスから非常に耳障りな異常音が発生するという問題がある。 Very harsh abnormal sound from the transformer there is a problem that occurs. 【0008】すなわち、図9において、強制停止期間から発振期間に切り替わる時点toに着目すると、直流電源電圧が低い場合(A)にはトランスの1次電流Ipが同図の(5)に示すようにそれほど急峻でない勾配で立ちあがり、出力電圧も(6)に示すように緩やかな増減を繰り返すのであるが、直流電源電圧が高い場合(B) Namely, in FIG. 9, when attention is focused from the forced stop period in time to switch to the oscillation period, if the DC power supply voltage is low (A) has a primary current Ip of the transformer as shown in (5) in FIG. If the rise in slope is not so steep, the output voltage although repeating the gradual increase and decrease as shown in (6), the DC power supply voltage is high (B)
には、(7)に示すように電流が急激に流れ込み、出力電圧も急峻な立ちあがりを呈するため、トランスの巻線や鉄心には間欠的な強い電磁力が働き、結果として耳障りなノイズが発生することになる。 The, (7) current flows abruptly as shown in, for presenting an output voltage sharply rising, intermittent strong magnetic force acts on the winding and iron core of the transformer, resulting in unpleasant noise occurs It will be. 【0009】したがって、この発明の課題は、軽負荷時に間欠発振動作をさせて消費電力の低減を図るコンバータにおいて、如何なる入力電圧に対してもトランスからの異常音が発生しないようなDC/DCコンバータの制御方法を提供することにある。 Accordingly, an object of the present invention, in the converter to reduce the power consumption is not an intermittent oscillation operation at low load, DC / DC converter, such as abnormal sound from the transformer is not generated for any input voltage and to provide a control method. 【0010】 【課題を解決するための手段】この発明によれば、前記課題は、直流電源電圧をスイッチ素子のオンオフ動作により一定電圧の直流出力に変換するものであって、無負荷を含む軽負荷時に発振期間と強制停止期間を設けてスイッチ素子を間欠発振動作させるDC/DCコンバータにおいて、PWM信号の生成のために出力指令値と比較される三角波電圧の波高値を直流電源電圧の高低に応じて制御することにより、スイッチ素子のオン幅を変化させることによって達成できる。 [0010] [Means for Solving the Problems] According to the present invention, the object is a DC power supply voltage be one that converts the DC output of the constant voltage by the on-off operation of the switch element, light including no load in the DC / DC converter for intermittently oscillating operation of the switching element is provided a forced stop period and the oscillation period when the load, the height of the triangular wave voltage which is compared with the output command value for the generation of the PWM signal to the level of the DC power supply voltage by controlling in response can be achieved by changing the oN width of the switching element. 【0011】三角波電圧の波高値を定めるためには、定電流で充放電されるコンデンサの容量を切り替えるのが簡便である(請求項2記載の発明)。 [0011] In order to determine the height of the triangular wave voltage is to switch the capacitance of the capacitor is charged and discharged at a constant current is simple (the second aspect of the invention). 【0012】また、三角波電圧の波高値を定めるために、三角波電圧を生成するコンデンサへの充電電流を切り替えるようにすることも目的にかなっている(請求項3記載の発明)。 Further, in order to determine the height of the triangular wave voltage (invention described in claim 3) which expediently also possible to switch the charging current to the capacitor for generating a triangular wave voltage. 【0013】 【発明の実施の形態】図1は請求項1記載の発明における実施の形態を示す回路接続図で、この回路の動作波形を示す図2および図3を参照しつつ動作態様を説明する。 [0013] Figure 1 DETAILED DESCRIPTION OF THE INVENTION In the circuit connection diagram showing an embodiment of the first aspect of the present invention, illustrating the operation mode with reference to FIGS. 2 and 3 shows the operation waveform of the circuit to. この実施例は、フライバックコンバータを例にとったもので、図の上半部は図8と同様の構成であるので、 This example, which was taken flyback converter as an example, since the upper half of the figure is the same configuration as FIG. 8,
図8と同じ素子ないし回路には同一の符号を付してある。 The same elements or circuit of FIG. 8 are denoted by the same reference numerals. 図8との相違は関数電圧発生回路の構成にある。 The difference between FIG. 8 is in the configuration of the function voltage generating circuit. 【0014】図1における関数電圧発生回路18においては、コンパレータ7において出力電圧検出調整回路6 [0014] In the function voltage generating circuit 18 in FIG. 1, the output at the comparator 7 the voltage detection adjusting circuit 6
の出力指令値Vcと比較されるべき出力電圧Vtは、コンデンサ20と21の並列接続回路の端子電圧となっている。 The output voltage Vt to be compared with the output command value Vc has a terminal voltage of the parallel connection circuit of a capacitor 20 and 21. より正確にいえば、コンデンサ20は直接に、そしてコンデンサ21はスイッチ回路22を介して大地電位に接続されている。 More precisely, the capacitor 20 is directly, and capacitor 21 is connected to ground potential through the switch circuit 22. これらの二つのコンデンサは、矩形波信号発生回路23の出力によりオンオフされるスイッチ回路24および25ならびに定電流回路26および27により、一定電圧Vccから充放電される。 These two capacitors, the switch circuit 24 and 25 and constant current circuit 26 and 27 are turned on and off by the output of the square wave signal generating circuit 23, it is charged and discharged from the constant voltage Vcc. したがって、コンデンサ20および21の端子電圧は三角波状に変化する。 Therefore, the terminal voltage of the capacitor 20 and 21 is changed to a triangular waveform. 【0015】コンデンサ21に直列接続されたスイッチ回路22は、次のようにオンオフされる。 The switch circuit 22 connected in series to the capacitor 21 is turned on and off as follows. すなわち、直流電源1の電圧を抵抗28と抵抗29とで分圧した電圧Vdと基準電圧Vbとをコンパレータ30で比較し、分圧電圧Vdが基準電圧Vbより小さいと、コンパレータ30の出力によりコンデンサ21に直列接続されたスイッチ22がオンする。 Capacitor that is, the voltage Vd and the reference voltage Vb voltage divided by the resistors 28 and the resistance 29 of the DC power source 1 as compared with the comparator 30, and the divided voltage Vd is less than the reference voltage Vb, the output of the comparator 30 switch 22 connected in series are turned on to 21. この結果、三角波電圧を形成するためのコンデンサはコンデンサ20とコンデンサ21の並列接続となり、容量が増加するため三角波電圧の波高値は低くなる。 As a result, the capacitors for forming the triangular wave voltage becomes the parallel connection of the capacitor 20 and the capacitor 21, the peak value of the triangular wave voltage for capacitance increases decreases. したがってコンパレータ7の出力によるスイッチ素子2のオン時間は長くなる。 Thus on-time of the switching element 2 by the output of the comparator 7 becomes longer. 【0016】逆に、分圧電圧Vdが基準電圧Vbを超えると、コンパレータ30の出力によりコンデンサ21に直列接続されたスイッチ22がオフして、三角波を生成するコンデンサはコンデンサ20だけとなり、三角波電圧の波高値は高くなる。 [0016] Conversely, the divided voltage Vd exceeds the reference voltage Vb, a switch 22 connected in series to the capacitor 21 by the output of the comparator 30 is turned off, the capacitor for generating the triangular wave becomes only the capacitor 20, the triangular wave voltage wave height increases of. この結果、コンパレータ7の出力によるスイッチ素子2のオン時間は短くなる。 As a result, the ON time of the switching element 2 by the output of the comparator 7 becomes shorter. なお、 It should be noted that,
コンデンサ20とコンデンサ21の容量は1:2程度に選ぶのがよい。 Capacitance of the capacitor 20 and the capacitor 21 is 1: good to choose the order of 2. 【0017】これを、図2および図3を参照して説明すると、直流電源電圧が低く、従って図1に示す分圧電圧Vdが基準電圧Vbを超えないような場合には、三角波電圧Vtも相応して図2の(1)で示すように波高値が低く、出力指令値Vcが三角波電圧Vtを超える時間も長くなるので、図2の(3)に示すPWM信号Vm、したがって(4)に示すゲート信号Vgもある程度の幅を持つことになり、スイッチ素子2のオン時間も長くなる。 [0017] This is explained with reference to FIGS. 2 and 3, a DC power supply voltage is low, when the divided voltage Vd shown in FIG. 1 so as not to exceed the reference voltage Vb, therefore, the triangular wave voltage Vt even correspondingly low peak value as shown in FIG. 2 (1), the output command value Vc is time becomes longer exceeding the triangular wave voltage Vt, PWM signal Vm shown in FIG. 2 (3) Therefore, (4) will have a certain width gate signal Vg shown in, the longer the on-time is also the switching element 2. トランスの1次電流Ipも(5)に示すように低い波形となり、出力電圧Voも緩やかに変化する。 Transformer primary current Ip also becomes lower waveform as shown in (5), the output voltage Vo is also changed gradually. 【0018】これに対して直流電源電圧が高く、したがって図1の分圧電圧Vdが基準電圧Vbを超えるような場合には、三角波電圧Vtも相応して図3の(1)で示すように波高値が高く、出力指令値Vcが三角波電圧V The high DC power supply voltage to this, therefore when the divided voltage Vd of FIG. 1 that exceeds the reference voltage Vb, as shown triangular wave voltage Vt even correspondingly in (1) in FIG. 3 high peak value, the output command value Vc is the triangular wave voltage V
tを超える時間は短くなるので、図3の(3)に示すP Because time is short more than t, P shown in (3) in FIG. 3
WM信号Vm、したがって(4)に示すゲート信号Vg A gate signal Vg that indicates WM signal Vm, thus (4)
も相対的に幅が狭くなり、スイッチ素子2のオン時間はそれに応じて短くなる。 Even relatively width is narrow, the on time of the switching element 2 is shortened accordingly. この結果、トランスの1次電流Ipも(5)に示すように従来のような大きな値とはならず、出力電圧Vpも緩やかな変化を見せる。 As a result, not the large value as in the prior as shown in even primary current Ip of the transformer (5), the output voltage Vp also show gradual changes. 【0019】かくして、図1の実施例によれば、直流電源の電圧が変化してもトランスに流入する電流の変化は従来よりも軽減されることになり、したがってトランスの異常音の発生も少なくなるかレベルが低くなる。 [0019] Thus, according to the embodiment of FIG. 1, a change in current flowing into the transformer even when the voltage of the DC power source is changed would be alleviated than before, thus also generates less transformer abnormal sound made or level is low. 【0020】図4は、この発明の異なる実施例を示すもので、図5および図6はその動作波形図である。 [0020] Figure 4 shows a different embodiment of the invention, FIGS. 5 and 6 is an operation waveform diagram thereof. 図4において、ブロック18で簡単に示されている回路は、図1の関数電圧発生器18と同様のものである。 4, the circuit shown briefly in block 18 is the same as the function voltage generator 18 of FIG. 図4の実施例ではさらに図1の定電流充放電回路と同様な回路構成からなる回路素子33〜38を含む回路と、インピーダンス変換素子39とダイオード40とを備えている。 A circuit including a circuit element 33 to 38 made of the same circuit configuration as further constant current charging and discharging circuit of Figure 1 in the embodiment of FIG. 4, and a impedance conversion element 39 and the diode 40. 【0021】図4の回路においては、スイッチ素子2の発振期間と強制停止期間とを制御する矩形波信号発生回路33の出力Vsを用い、強制停止期間から発振期間に切り替わるタイミングでは、定電流回路36を介してコンデンサ38を定電流充電するようにスイッチ回路34 [0021] In the circuit of Figure 4, using the output Vs of the square wave signal generating circuit 33 for controlling the forcible stop period and the oscillation period of the switch element 2, at the timing of switching the oscillation period from the forced stop period, the constant current circuit the switch circuit 34 to the constant current charging the capacitor 38 through 36
をオンさせ、コンデンサ38の電圧が徐々に増加するようにする。 It was turned on so that the voltage of the capacitor 38 is gradually increased. コンデンサ38はスイッチ回路34に接続された電源電圧Vccまで充電される。 Capacitor 38 is charged up to the supply voltage Vcc that is connected to the switch circuit 34. 【0022】また、発振期間から強制停止期間に切り替わるタイミングでは、定電流回路37を介してコンデンサ38を定電流放電するようにスイッチ回路35をオンさせ、コンデンサ38の電圧が徐々に減少するようにする。 Further, at the timing of switching the suspended period from an oscillation period, the switch circuit 35 is turned on so that a constant current discharges the capacitor 38 through the constant current circuit 37, so that the voltage of the capacitor 38 gradually decreases to. コンデンサ38の電圧は、ゼロになるまで放電される。 Voltage of the capacitor 38 is discharged to zero. 【0023】さらに、コンデンサ38の電圧Vcsをインピーダンス変換素子39でインピーダンス変換した信号と、出力電圧検出調整回路6の出力信号Vcとを比較して、小さい方の値をコンパレータ7の入力信号とするために、ダイオード40が接続されている。 Furthermore, by comparing the signal obtained by impedance conversion by the impedance converter 39 the voltage Vcs of the capacitor 38, the output signal Vc of the output voltage detection adjusting circuit 6, a smaller value as the input signal of the comparator 7 for the diode 40 is connected. これにより、強制停止期間から発振期間に切り替わったタイミング以後は、コンデンサ38の充電スピードでスイッチ素子2のオンパルス幅が徐々に増加し、コンデンサ38の電圧よりも出力電圧検出調整回路6の出力Vcの方が小さくなると、出力電圧が一定となるようにスイッチ素子2のオンオフのタイミングがPWM制御される。 Thus, after the timing of switching to the oscillation period from the forced stop period, the switching element 2 in the charging speed of the capacitor 38-pulse width is gradually increased, the output Vc of the output voltage detection adjusting circuit 6 than the voltage of the capacitor 38 If it is smaller, the output voltage of a timing of on-off switching element 2 to be constant is PWM controlled. また、 Also,
発振期間から強制停止期間に切り替わったタイミング以後は、コンデンサ38の放電スピードでスイッチ素子2 Timing after switching to the suspended period from an oscillation period, the switching element 2 in the discharge speed of the capacitor 38
のオンパルス幅が叙叙に減少し、コンデンサ38の電圧がゼロになった時点で、スイッチ素子2が完全にオフ状態となる。 The pulse width is reduced to Jo - Jo, when the voltage of the capacitor 38 becomes zero, the switching element 2 is turned off completely. 【0024】この時、図1の実施例と同様に、関数電圧発生回路18は直流電源の電圧にしたがって波高値が切り替わり、直流電源1の電圧が変化してもトランス3に流入する電流の変化が軽減し、トランスの異常音の発生が低減される。 [0024] At this time, as in the embodiment of FIG 1, the function voltage generating circuit 18 is a peak value switches according to the voltage of the DC power supply, a change in current flowing in the transformer 3 even when the voltage of the DC power source 1 is changed but reduce the occurrence of the transformer of the abnormal sound is reduced. 【0025】図5は直流電源電圧が高いときの動作波形、図6は直流電源電圧が低いときの操作波形を示す。 FIG. 5 is operation waveforms when the DC power source voltage is high, FIG. 6 shows the operation waveforms when the DC power source voltage is low.
両図において、(1)は矩形波信号発生回路Vsの出力信号Vs、(2)はコンデンサ38の端子電圧Vcs、 In both figures, (1) the output signal Vs of the square wave signal generating circuit Vs, (2) the terminal voltage Vcs of the capacitor 38,
(3)は三角波電圧Vtと出力指令値Vcとの関係、 (3) the relationship between the output command value Vc with the triangular wave voltage Vt,
(4)は出力電圧Voと設定電圧Vref、(5)は三角波電圧Vtと出力指令値との関係、(6)はPWM信号Vm、(7)はトランス1次電流Ipをそれぞれ示す。 (4) the set voltage Vref and the output voltage Vo, (5) the relationship between the output command value and the triangular wave voltage Vt, (6) are respectively PWM signal Vm, (7) is a transformer primary current Ip. 【0026】図5と図6を比較すれば明らかなように、 [0026] As is clear from a comparison of FIG. 5 and FIG. 6,
直流電源電圧が低い場合には、図5に示すように三角波電圧Vtも低く、このため出力指令値Vcが三角波電圧Vtを超えている期間が相対的に長くなり、したがってPWM信号Vmの幅が長くなるので、スイッチ素子2のオン時間も相応して長くなる。 When the DC power supply voltage is low, the triangular wave voltage Vt as shown in Figure 5 is low and therefore the output command value Vc period that exceeds the triangular wave voltage Vt becomes relatively long, therefore the width of the PWM signal Vm since longer, it becomes longer correspondingly also the on-time of the switching element 2. 【0027】これに対して直流電源電圧が高い場合には、図6に示すように三角波電圧Vtも高く、このため出力指令値Vcが三角波電圧Vtを超えている期間が相対的に短くなり、したがってPWM信号Vmの幅が短くなるので、スイッチ素子2のオン時間も相応して短くなる。 [0027] When the DC power source voltage is high contrast, higher triangular wave voltage Vt as shown in FIG. 6, period. Thus the output command value Vc exceeds the triangular wave voltage Vt becomes relatively short, Thus the width of the PWM signal Vm is shortened, shortens correspondingly also the on-time of the switching element 2. 【0028】かくして、図4の実施例によれば、図1の実施例と同様に直流電源の電圧が変化してもトランスに流入する電流の変化は従来よりも軽減されることになり、したがってトランスの異常音の発生も少なくなるかレベルが低くなる。 [0028] Thus, according to the embodiment of FIG. 4, the change in current flowing into the transformer also vary the embodiment similarly to the voltage of the DC power supply of FIG. 1 will be reduced than conventionally, thus even fewer become one level occurrence of a transformer of the abnormal sound is lowered. 【0029】図7は、本発明の実施の際に用いられる関数電圧発生回路の異なる実施例を示す。 [0029] Figure 7 illustrates a different embodiment of the function voltage generating circuit used in the practice of the present invention. 直流電源1の電圧を、抵抗41と抵抗42とで分圧し、分圧電圧Vdと基準電圧Vbとをコンパレータ43で比較し、インバータ43Nを介してアンドゲート44の一方の入力に印加する。 The voltage of the DC power source 1, divided by the resistor 41 and the resistor 42, and a divided voltage Vd with the reference voltage Vb is compared by the comparator 43 is applied to one input of the AND gate 44 via the inverter 43N. アンドゲート44の他方の入力には矩形波信号発生回路45の出力が与えられる。 The other input of the AND gate 44 is given the output of the square wave signal generating circuit 45. この矩形波信号発生回路45の出力は、定電流回路48又は49と直列接続されたスイッチ回路46又は47に与えられ、これらのスイッチ回路を交互にオンオフし、コンデンサ52を所定の定電流で充電もしくは放電する。 The output of the square wave signal generating circuit 45 is supplied to the constant current circuit 48 or 49 and the switching circuit 46 or 47 which are connected in series, and off these switch circuits are alternately charges the capacitor 52 at a predetermined constant current or discharge. 前述のアンドゲート44の出力はスイッチ回路50に与えられ、このスイッチ回路50がオンすると、付加的な定電流回路51がコンデンサ52の充電に参加する。 The output of the aforementioned AND gate 44 is supplied to the switch circuit 50, the switch circuit 50 is turned on, additional constant current circuit 51 to participate in the charging of the capacitor 52. 定電流回路48と51 Constant current circuit 48 and 51
の供給電流比は1:2程度に選ぶのが良い。 Supply current ratio of 1: a good pick to about 2. 【0030】かかる構成の下で装置を運転した場合、矩形波信号発生回路45の出力により、コンデンサ52は定電流充電または定電流放電を繰り返し、三角波電圧V [0030] When the apparatus was operated under such a configuration, the output of the square wave signal generating circuit 45, the capacitor 52 is repeated constant current charging or constant current discharge, the triangular wave voltage V
tを生成するが、直流電源電圧が低く、分圧電圧Vdが基準電圧Vbより小さいと、コンパレータ43の反転出力により、定電流回路51に直列接続されたスイッチ5 Generating a t but, DC power supply voltage is low, and the divided voltage Vd is less than the reference voltage Vb, the inverted output of the comparator 43, the switch 5 to the constant current circuit 51 connected in series
0はオンせず、三角波形を生成するコンデンサへの充電電流は定電流回路48のみが貢献することになるので、 0 does not turn on, the charging current to the capacitor for generating a triangular waveform will contribute only the constant current circuit 48,
三角波電圧の波高値は低くなる。 The height of the triangular wave voltage is lower. このとき、コンパレータ7の出力によるスイッチ素子2のオン時間は長くなる。 At this time, the on time of the switching element 2 by the output of the comparator 7 becomes longer. 【0031】直流電源電圧が高くなり、分圧電圧Vdが基準電圧Vbを超えると、コンパレータ43の反転出力により定電流回路45に直列接続されたスイッチ44がオンして、新たな定電流回路51がコンデンサ52の充電のために投入されるので、三角波電圧の波高値は高くなる。 The DC power supply voltage is increased, the divided voltage Vd exceeds the reference voltage Vb, the switch 44 connected in series by inverting the output to the constant current circuit 45 of the comparator 43 is turned on, a new constant-current circuit 51 There therefore is introduced to charge the capacitor 52, the peak value of the triangular wave voltage is increased. この時コンパレータ7の出力によるスイッチ素子2のオン時間は短くなる。 ON time of the switching element 2 by the output of the time comparator 7 becomes shorter. 【0032】ここで、上記実施の形態において、コンデンサ20とコンデンサ21の容量比あるいは定電流回路48と定電流回路51の供給電流比の例として1:2をあげたが、これは直流電源1の電圧の想定される最低値と最大値の比が1:3程度の場合を例示したものであり、前記電圧の想定される最低値と最大値の比に基づいて適宜決定すればよい。 [0032] Here, in the above embodiment, 1 as an example of the supply current ratio of the capacitor 20 and the capacitance ratio or a constant current circuit 48 and the constant current circuit 51 of the capacitor 21: 2 has been mentioned, this is a direct current power source 1 the ratio of minimum and maximum values ​​that are assumed voltage 1: an illustration of a case of the order of 3, may be appropriately determined based on the ratio of minimum and maximum values ​​that are assumed in the voltage. 【0033】 【発明の効果】以上の通り、この発明によれば、直流電源電圧を検出してPWM演算用の関数電圧の波高値を変化させることによって、電源電圧の変化時に生じるトランスの異常音を低減させることができ、待機時など軽負荷時における消費電力を低減することが必要な用途においては、待機専用の特別なコンバータを省略することが可能となり、低価格の電源システムを提供することができる。 [0033] [Effect of the Invention] As described above, according to the present invention, by detecting the DC power supply voltage changes the peak value of the function voltage for PWM operation, transformer abnormal sound generated during changing of the power supply voltage can be reduced, in applications requiring it to reduce power consumption during standby, such as a light load, it is possible to omit a special converter standby only, to provide a low cost power supply system can.

【図面の簡単な説明】 【図1】本発明の第一の実施例における回路接続図【図2】第一の実施例の動作波形図(直流電源電圧が低い場合) 【図3】第一の実施例の動作波形図(直流電源電圧が高い場合) 【図4】本発明の第二の実施例における回路接続図【図5】第二の実施例の動作波形図(直流電源電圧が低い場合) 【図6】第二の実施例の動作波形図(直流電源電圧が高い場合) 【図7】本発明における関数電圧発生回路の異なる実施例の回路接続図【図8】従来の実施例の回路接続図【図9】従来回路の動作波形図【符号の説明】 1 直流電源2 スイッチ素子3 トランス4 整流器5 コンデンサ6 電圧検出調整回路7 コンパレータ9 ゲートドライブ回路10,23,33 矩形波信号発生回路11 アンドゲート18 関数電圧発生 BRIEF DESCRIPTION OF THE DRAWINGS circuit connection diagram in the first embodiment of the invention, FIG 2 shows operation waveforms of the first embodiment (when the DC supply voltage is low) [3] first example operation waveform diagram (when the DC supply voltage is high) circuit connection diagram in the second embodiment of the invention, FIG 5 shows an operation waveform diagram of a second embodiment (the DC power supply voltage is low If) 6 second operation waveform diagram of an embodiment of a (when the DC supply voltage is high) circuit connection diagram of different embodiments of the function voltage generating circuit in the present invention; FIG 8 conventional example circuit connection diagram of FIG. 9 operation waveform diagram of a conventional circuit [eXPLANATION oF sYMBOLS] 1 DC power source 2 switching element 3 transformer 4 rectifier 5 capacitor 6 voltage detection adjusting circuit 7 comparator 9 the gate drive circuit 10,23,33 square wave signal generating circuit 11 and the gate 18 function voltage generator 路39 インピーダンス変換回路 Road 39 impedance conversion circuit

───────────────────────────────────────────────────── フロントページの続き (72)発明者 森谷 清志 石川県松任市下柏野町153番地 株式会社 ナナオ内(72)発明者 桑原 今朝信 神奈川県川崎市川崎区田辺新田1番1号 富士電機株式会社内Fターム(参考) 5H730 AA02 BB43 DD01 EE02 EE07 EE59 FD01 FD11 FF03 FG03 FG05 ────────────────────────────────────────────────── ─── of the front page continued (72) inventor Kiyoshi Moriya Ishikawa Prefecture Matto City Shimokashiwano-cho, 153 address, Ltd. in Nanao (72) inventor Kuwahara this morning Shin Kawasaki City, Kanagawa Prefecture Kawasaki-ku, Tanabeshinden No. 1 No. 1 Fuji Electric Co., Ltd. in the F-term (reference) 5H730 AA02 BB43 DD01 EE02 EE07 EE59 FD01 FD11 FF03 FG03 FG05

Claims (1)

  1. 【特許請求の範囲】 【請求項1】 直流電源電圧をスイッチ素子のオンオフ動作により一定電圧の直流出力に変換するものであって、無負荷を含む軽負荷時に発振期間と強制停止期間を設けてスイッチ素子を間欠発振動作させるDC/DCコンバータにおいて、PWM信号の生成のために出力指令値と比較される三角波電圧の波高値を直流電源電圧の高低に応じて制御することにより、スイッチ素子のオン幅を変化させることを特徴とするDC/DCコンバータの制御方法。 A converts the Patent Claims 1. A DC power supply voltage to a DC output of a constant voltage by the on-off operation of the switch element by providing a forced stop period and the oscillation period at light load including a no-load in the DC / DC converter for intermittently oscillating operation of the switching element, by controlling in accordance with height of the triangular wave voltage which is compared with the output command value for the generation of the PWM signal to the level of the DC power supply voltage, on switching element DC / DC converter control method, characterized in that to change the width. 【請求項2】 三角波電圧の波高値を定めるために、定電流で充放電されるコンデンサの容量を切り替えることを特徴とする請求項1記載のDC/DCコンバータの制御方法。 2. To determine the height of the triangular wave voltage control method according to the DC / DC converter of claim 1, wherein the switching the capacitance of the capacitor is charged and discharged at a constant current. 【請求項3】 三角波電圧の波高値を定めるために、三角波電圧を生成するコンデンサへの充電電流を切り替えることを特徴とする請求項1記載のDC/DCコンバータの制御方法。 To 3. for determining the height of the triangular wave voltage control method according to claim 1 DC / DC converter, wherein the switching the charging current to the capacitor for generating a triangular wave voltage.
JP2002098482A 2002-04-01 2002-04-01 Dc-dc converter control method Pending JP2003299356A (en)

Priority Applications (1)

Application Number Priority Date Filing Date Title
JP2002098482A JP2003299356A (en) 2002-04-01 2002-04-01 Dc-dc converter control method

Applications Claiming Priority (1)

Application Number Priority Date Filing Date Title
JP2002098482A JP2003299356A (en) 2002-04-01 2002-04-01 Dc-dc converter control method

Publications (1)

Publication Number Publication Date
JP2003299356A true true JP2003299356A (en) 2003-10-17

Family

ID=29387955

Family Applications (1)

Application Number Title Priority Date Filing Date
JP2002098482A Pending JP2003299356A (en) 2002-04-01 2002-04-01 Dc-dc converter control method

Country Status (1)

Country Link
JP (1) JP2003299356A (en)

Cited By (9)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
JP2006174626A (en) * 2004-12-17 2006-06-29 Oki Electric Ind Co Ltd Switching regulator
JP2006246664A (en) * 2005-03-07 2006-09-14 Fuji Electric Device Technology Co Ltd Control method for switching power supply circuit
WO2007037033A1 (en) * 2005-09-27 2007-04-05 Mitsumi Electric Co., Ltd. Electric power supply device
EP1909382A2 (en) * 2006-10-04 2008-04-09 Power Integrations, Inc. Method and apparatus to reduce audio frequencies in a switching power supply
US7615984B2 (en) 2005-09-14 2009-11-10 Fuji Electric Device Technology Co., Ltd. DC-DC converter and method of controlling thereof
JP2012010439A (en) * 2010-06-22 2012-01-12 Canon Inc Switching power supply
CN103023333A (en) * 2011-09-23 2013-04-03 电力集成公司 Power supply controller with minimum-sum multi-cycle modulation
JP5285802B1 (en) * 2012-09-27 2013-09-11 Eizo株式会社 A power supply device and a display device
US9106148B2 (en) 2013-11-27 2015-08-11 Canon Kabushiki Kaisha Power supply apparatus and image forming apparatus

Cited By (23)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
JP4545576B2 (en) * 2004-12-17 2010-09-15 Okiセミコンダクタ株式会社 Switching regulator
JP2006174626A (en) * 2004-12-17 2006-06-29 Oki Electric Ind Co Ltd Switching regulator
JP2006246664A (en) * 2005-03-07 2006-09-14 Fuji Electric Device Technology Co Ltd Control method for switching power supply circuit
US7242596B2 (en) 2005-03-07 2007-07-10 Fuji Electric Device Technology Co., Ltd. Control method for switching power supply circuit
JP4617931B2 (en) * 2005-03-07 2011-01-26 富士電機システムズ株式会社 Control method of the switching power supply circuit
US7615984B2 (en) 2005-09-14 2009-11-10 Fuji Electric Device Technology Co., Ltd. DC-DC converter and method of controlling thereof
WO2007037033A1 (en) * 2005-09-27 2007-04-05 Mitsumi Electric Co., Ltd. Electric power supply device
US8537570B2 (en) 2006-10-04 2013-09-17 Power Integrations, Inc. Method and apparatus to reduce audio frequencies in a switching power supply
JP2008092793A (en) * 2006-10-04 2008-04-17 Power Integrations Inc Method and apparatus to reduce audio frequency in switching power supply
JP2011109913A (en) * 2006-10-04 2011-06-02 Power Integrations Inc Controller used for power regulator
US7965524B2 (en) 2006-10-04 2011-06-21 Power Integrations, Inc. Method and apparatus to reduce audio frequencies in a switching power supply
EP1909382A2 (en) * 2006-10-04 2008-04-09 Power Integrations, Inc. Method and apparatus to reduce audio frequencies in a switching power supply
US8194423B2 (en) 2006-10-04 2012-06-05 Power Integrations, Inc. Method and apparatus to reduce audio frequencies in a switching power supply
US9219418B2 (en) 2006-10-04 2015-12-22 Power Integrations, Inc. Method and apparatus to reduce audio frequencies in a switching power supply
EP1909382A3 (en) * 2006-10-04 2014-07-09 Power Integrations, Inc. Method and apparatus to reduce audio frequencies in a switching power supply
US8774669B2 (en) 2010-06-22 2014-07-08 Canon Kabushiki Kaisha Switching power source and image forming apparatus including the same
JP2012010439A (en) * 2010-06-22 2012-01-12 Canon Inc Switching power supply
US8693217B2 (en) 2011-09-23 2014-04-08 Power Integrations, Inc. Power supply controller with minimum-sum multi-cycle modulation
CN103023333A (en) * 2011-09-23 2013-04-03 电力集成公司 Power supply controller with minimum-sum multi-cycle modulation
US9531279B2 (en) 2011-09-23 2016-12-27 Power Integrations, Inc. Power supply controller with minimum-sum multi-cycle modulation
WO2014050165A1 (en) * 2012-09-27 2014-04-03 Eizo株式会社 Power source device and display device
JP5285802B1 (en) * 2012-09-27 2013-09-11 Eizo株式会社 A power supply device and a display device
US9106148B2 (en) 2013-11-27 2015-08-11 Canon Kabushiki Kaisha Power supply apparatus and image forming apparatus

Similar Documents

Publication Publication Date Title
US7116090B1 (en) Switching control circuit for discontinuous mode PFC converters
US6130831A (en) Positive-negative pulse type high frequency switching power supply unit
US4864547A (en) Regulated ultrasonic generator
US6323623B1 (en) Charging device and charging method thereof
US6670779B2 (en) High power factor electronic ballast with lossless switching
US6442047B1 (en) Power conversion apparatus and methods with reduced current and voltage switching
US20020071295A1 (en) Method of controlling DC/DC converter
US20040196669A1 (en) Switching power supply
US20070164720A1 (en) Switch-mode power supply controllers
US6548966B2 (en) Discharge lamp lighting device
US6097614A (en) Asymmetrical pulse width modulated resonant DC-DC converter with compensating circuitry
US5686797A (en) Electronluminescent lamp inverter
JPH06335241A (en) Transformer-coupled secondary dc power-supply forming device
JP2003333839A (en) Method for controlling power supply and power supply controller
WO1985001400A1 (en) Minimization of harmonic contents for mains operated solid state inverters driving gas discharge lamps
US7394232B2 (en) Interleaved switching converters in ring configuration
US20060171179A1 (en) Low audible noise power supply method and controller therefor
JP2005094827A (en) High voltage pulse power supply
JPH08227790A (en) High-frequency heating device
JP2006204044A (en) Resonance switching power supply
JP2007013916A (en) Signal generator
JP2001204170A (en) Capacitor charging device
US20110176335A1 (en) Resonant converters and burst mode control method thereof
US5140510A (en) Constant frequency power converter
JP2004236461A (en) Switching controller and switching power supply equipped with the switching controller

Legal Events

Date Code Title Description
A621 Written request for application examination

Free format text: JAPANESE INTERMEDIATE CODE: A621

Effective date: 20050324

A131 Notification of reasons for refusal

Free format text: JAPANESE INTERMEDIATE CODE: A131

Effective date: 20071206

A02 Decision of refusal

Free format text: JAPANESE INTERMEDIATE CODE: A02

Effective date: 20080327