JP2003219635A - Semiconductor integrated circuit for controlling power source and power source device - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To solve the problem that since a conventional overcurrent detector in a level of an excess power to be detected in response to a level of an AC power source, an element having a high yield strength so as to sufficiently endure against the excess power as the element such as a switching MOSFET or the like is used, or it is necessary to separately constitute the circuit for detecting the excess power by an outside attached circuit so that the cost of power source system is raised. <P>SOLUTION: The IC for controlling the power source for constituting a switching power source device comprises a converter (34, 37) for changing a comparison reference voltage (Vth) or an apparent threshold value (VTH) of a comparator (31) for detecting the excess power in response to an input voltage (Vin), or more particularly changing a threshold value of the comparator in proportion to an inverse number of the input voltage. <P>COPYRIGHT: (C)2003,JPO

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a power supply control device and a technique effectively applied to a power supply device using the power supply control device,
In particular, the present invention relates to a technique for preventing overpower in an AC-DC (alternating current-direct current) conversion type power supply device, for example, a technique effectively applied to a PFC (power factor correction) type power supply device.

[0002]

2. Description of the Related Art Conventionally, a method of obtaining a DC power source from a commercial AC power source is used to rectify an alternating current from a 100 V AC power source by a rectifier circuit composed of a diode bridge to charge a smoothing capacitor, Charge voltage B +
Is supplied to the switching regulator as the primary voltage to generate a desired DC voltage by PWM (pulse width modulation) control or the like. With this method,
When the charging voltage B + of the smoothing capacitor becomes lower than the rectified voltage from the rectifier circuit, a current flows from the rectifier circuit to the smoothing capacitor to charge the capacitor. Therefore, a large current Iac is temporarily flowed to the smoothing capacitor from the AC power supply via the rectifier circuit. Then, there is a problem that a harmonic current is generated by this momentary current and harmonic noise is carried on the AC power supply.

Therefore, in order to reduce this harmonic noise, a booster circuit is provided between the rectifier circuit and the switching regulator, the voltage rectified by the rectifier circuit is boosted by the booster circuit, and the boosted voltage B ++ thus formed is switched. PFC to be used as the primary voltage of the regulator
There is a (power factor correction) type power supply device, and a PFC control semiconductor integrated circuit (hereinafter, also referred to as PFC control IC) for forming a booster circuit of the power supply device has been put into practical use. In the power supply device using the PFC control IC,
The smoothing capacitor is not provided at the output of the rectifier circuit, but is provided at the output of the booster having the coil and the switch in the booster circuit. IC for PFC control is boosted voltage B +
While the PWM control of the switch is performed so that + becomes a desired voltage such as 385V, the level of the output voltage B + of the rectifier circuit is detected and the PWM control of the switch is corrected to instantaneously change the current Iac. By suppressing the fluctuation, the harmonic current is suppressed. Note that the PFC control IC corrects the PWM control of the switch so that the phase of the current Iac fetched from the rectifier circuit to the booster circuit is similar to that of the output voltage from the rectifier circuit. With this, the power factor is improved.

[0004]

SUMMARY OF THE INVENTION In the power supply device as described above, a coil forming a booster in the booster circuit,
If the overcurrent continues to flow to the switch forming the booster for some reason such as the smoothing capacitor being short-circuited, the power supply device may not operate normally. Therefore, the switching power supply device includes a detection circuit that detects an overcurrent flowing through the switch, and has a function of shutting off the switch when the overcurrent is detected. However, in an actual power supply device, overpower is a problem rather than overcurrent flowing through the switch, and it is necessary to suppress power below a predetermined level. However, the overcurrent detection circuit of the conventional switching power supply device is composed of a resistor for converting a current into a voltage and a comparator for comparing the converted voltage with a reference voltage of a predetermined level, and detects the overcurrent flowing through the switch. It was something to do.

Therefore, in the conventional overcurrent detection circuit, the overcurrent detection level Vth is constant as indicated by the symbol b in FIG. 9 even when the level of the AC power supply, that is, the input voltage Vin of the power supply device is different. As a result, the conventional overcurrent detection circuit has a characteristic that, from the viewpoint of electric power, as shown by a symbol B in FIG. 12, as the input voltage Vin becomes higher, the overpower detection level becomes higher accordingly. Was. This is because the limiter level Plm of the overpower is expressed by the following equation Plm = Vin × | Vth | / (√2 × Rcs) (1), where Rcs is the value of the resistance for overcurrent detection. is there.

By the way, it is desirable that the power supply control IC constituting the power supply device has high versatility, that is, the same IC can be used even if the voltage of the AC power supply is different because the cost can be reduced. However, the conventional overcurrent detection circuit has no problem when it is used for a planned AC power supply, but when it is used for a different AC power supply as described above, the level of overpower to be detected is It will change.
Therefore, conventionally, it has been necessary to use an element having a high withstand voltage that can sufficiently withstand overpower as the switch, or to separately configure a circuit for detecting overpower with an external circuit. door there was a problem that rises.

An object of the present invention is to provide an overpower detection circuit suitable for a power supply control IC used in a switching type power supply device. Another object of the present invention is to provide a versatile switching power supply control IC capable of detecting a predetermined level of overpower even when used for different AC power supplies. Regarding the above and other objects and novel features of the present invention,
It will be apparent from the description of this specification and the accompanying drawings.

[0008]

The typical ones of the inventions disclosed in the present application will be outlined below. That is, the present invention changes the comparison reference voltage or the apparent threshold value of the comparator for detecting overpower in the power supply control IC that constitutes the switching type power supply device according to the input voltage . Specifically, a conversion circuit for changing the threshold value of the comparator in proportion to the reciprocal of the input voltage is provided. Here, the conversion circuit that changes the apparent threshold value of the comparator in accordance with the input voltage may be one that directly changes the threshold value, or the threshold value may be fixed and the detected voltage (monitor current I- It is also possible to change the V-converted voltage). According to the above-mentioned means, it is possible to detect overpower at substantially the same detection level even if the input voltage, that is, the level of the AC power supply is different, and thereby a highly reliable switching power supply system can be configured. The versatility of the power control IC used in the power system can be improved.

[0009]

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS Preferred embodiments of the present invention will be described below with reference to the drawings. FIG. 1 shows a schematic configuration of a PFC control IC according to the present invention and a power supply device using the same. In the figure, 11 is a rectifier circuit composed of a diode bridge for converting AC from the AC power supply 10 into DC, and 20 is a PF.
A C control IC, 16 is a switch MOSFET that is on / off controlled by a PFC control IC 20, 17 is a step-up coil, 18 is a step-up diode, and 12 is a smoothing capacitor. When the switch MOSFET 16 is turned on, a coil is formed. When the current from the AC power supply 10 is supplied to 17 and energy is accumulated and the switch MOSFET 16 is turned off, a current is discharged by the energy accumulated in the coil 17 and the smoothing capacitor 12 is charged through the diode 18. A DC voltage VRB sufficiently higher than an AC voltage such as 385V is generated.

In the PFC control IC 20 of this embodiment,
A power supply voltage terminal VCC to which a DC voltage such as DC12V is applied, a ground terminal GND to which a ground potential is applied, a monitor terminal IAC for the input current Iac, a monitor terminal VAC for the input voltage Vin, and a boosted output voltage. VRB monitor terminal PFC-FB and drive terminal PFC- which outputs a control pulse for turning on / off the switch MOSFET 16.
DRIVE, terminal PFC-CS for overpower monitoring and current feedback, and CR circuits 41, 42 for phase compensation
A terminal PFC-EO to which an external element constituting the
PFC-CAO is provided. In this embodiment, the DC voltage VRB boosted as described above is applied to the resistor R.
It is divided by 3 and R4 and fed back to the output voltage monitor terminal PFC-FB of the PFC control IC 20. The input current Iac obtained by rectifying the current from the AC power supply 10 by the diode bridge 11 is input to the input current monitor terminal IAC of the PFC control IC 20 via the resistor R0. The voltage Vin obtained by rectifying the alternating current supplied from the AC power source 10 by the rectifier circuit 11 is divided by the resistors R1 and R2, and is input as the input voltage Vin ′ to the PFC control IC 2
0 is applied to the input voltage monitor terminal VAC. Further, an overpower detection resistor Rcs is connected between one node n1 of the rectifier circuit 11 and the ground point GND, and a terminal of the resistor Rcs on the opposite side of the ground point GND is connected to the overpower of the PFC control IC 20. It is connected to the monitor terminal PFC-CS.

The PFC control IC 20 of this embodiment is connected to the input voltage monitor terminal VAC and has an input voltage Vin '.
Is connected to the output voltage monitor terminal PFC-FB and the potential difference between the feedback voltage of the DC voltage VRB and the reference voltage (2.5V). An error amplifier 22 for detecting and outputting a voltage VEO according to the potential difference; a multiplication circuit 23 connected to the input current monitor terminal IAC for outputting a current according to the input current Iac and the output voltage VEO of the error amplifier 22; A gain selection circuit 24 that selects the gain K of the multiplication circuit 23 according to the voltage detected by the power supply level detection circuit 21, a resistor Rmo that converts the output current Imo of the multiplication circuit 23 into a voltage, and the converted voltage. And an error amplifier 25 that outputs a voltage according to the difference between the ground potential and a switch MOSFET based on the output voltage of the error amplifier 25. 6, a drive pulse control circuit 26 that determines the duty of the pulse that drives 6, a oscillating circuit 27 that generates a clock with a frequency of 100 kHz that is the source of drive pulse generation, a driver 28 that outputs a drive pulse, and the like. There is.

The output terminal of the error amplifier 25 is the external terminal PF.
The phase compensating CR circuit 42 connected to the C-CAO and connected to the external terminal prevents the error amplifier 25 from oscillating in response to feedback at the PFC control switching frequency (100 kHz). In addition, since the output of the error amplifier 25 is supplied to the drive pulse control circuit 26, the switch MOSFET 1
The duty of the control pulse for driving 6 is the above-mentioned external terminal P
It becomes proportional to the voltage of FC-CAO. Further, the output terminal of the error amplifier 22 is also connected to the external terminal PFC-EO to which the CR circuit 41 for phase compensation is connected, and the error amplifier 22 at the frequency (50 Hz) of the input voltage Vin.
Is prevented from oscillating in response to feedback

The gain selection circuit 24 switches the gain K of the multiplication circuit 23 according to the voltage level detected by the power supply level detection circuit 21. As a result, the power supply capability of the power supply device can be made substantially the same when boosting AC100V and boosting AC220V, for example. Specifically, the gain K of the multiplication circuit 23 when the voltage from the AC 220V power source is input is set to be about 1/5 of the gain when the voltage from the AC 100V power source is input. It is switched by the selection circuit 24. The input voltage is AC
Gain K when detecting 200V or AC100V
About 1/4, and the input voltage is AC250V or AC100
When detecting V, the gain K may be switched to about 1/6. In this specification, A
C200V, AC220V, AC250V, etc. are collectively referred to as a 200V system. Similarly, AC100V and AC12
0V and the like are collectively referred to as 100V system.

The PFC control IC 20 of this embodiment is also used.
Is an overshoot monitoring circuit 2 which is connected to the output voltage monitor terminal PFC-FB and monitors the feedback voltage of the DC voltage VRB to detect the overshoot of the DC voltage VRB.
9 and a current control circuit 30 for controlling the operating current of the error amplifier 25 at the start (when the power is turned on) based on the detection signal of the overshoot detection circuit 29 and the detection signal of the power supply level detection circuit 21, Power monitor terminal PF
An overpower detection comparator 31 connected to the C-CS,
A power supply monitoring circuit 32 that is connected to the power supply voltage terminal VCC and that detects the rising of the power supply voltage to generate the reset signal RES, a reference voltage generation circuit 33 that generates the reference voltage Vref, and the like are provided.

The multiplication circuit 23 has an input voltage Vin, an output voltage VEO of the error amplifier 22 and a gain selection circuit 24.
A current Imo determined by the following equation Imo = K {Iac * (VEO-1)} is output from the gain K selected in step S1. Therefore, the current I
The waveform of mo becomes a waveform similar to the waveform of the input voltage Vin of (A), as shown in FIG. 2 (B). This current Imo
One terminal (node n2) of the resistor Rmo through which the
Error amplifier 2 whose inverting input terminal is connected to ground GND
5 is connected to the non-inverting input terminal of the amplifier 5, so that the potential of the node n2 is virtually grounded in the steady state.
Works. In addition, the output voltage monitor terminal PFC-F
The error amplifier 22 having the inverting input terminal connected to B has a terminal P
It operates so that the potential of FC-FB becomes 2.5 V, which is the same as the voltage applied to the non-inverting input terminal. as a result,
Switch MOSFET 1 by PFC control IC 20
The feedback system operates so that the DC voltage VRB generated by the ON / OFF operation of 6 becomes a voltage such as 385V determined by the ratio (710: 4.7) of the resistors R3 and R4.

The current I output from the multiplication circuit 23
In the steady state, the potential of the external terminal PFC-CS to which the other terminal of the resistor Rmo to which mo flows is connected is shown in FIG.
As shown in, the waveform 0 of the current Imo in (B) is vertically symmetric
It is set to a negative potential such as ~ -0.3V. Therefore, Isum = Imo (Rmo / Rc
A current as determined by s) flows. That is, the current Isu
The waveform is similar to the waveform of the waveform current Imo of m. However, if the coil 17 and the smoothing capacitor 12 are short-circuited and current continues to flow, the external terminal PF
C-CS is made a negative potential deeper than -0.3V. The PFC control IC 20 of this embodiment has an overpower detection comparator 31 having an inverting input terminal connected to the external terminal PFC-CS, and -0.5V or − applied to the non-inverting input terminal of the comparator 31. Since the threshold value switching circuit 34 for generating the comparison reference voltage (logical threshold value) Vth such as 0.25 V is provided, the resistance Rc
overcurrent flows to s and external terminal PFC-CS is -0.5V
Alternatively, when it becomes −0.25 V or less, this is detected and the driver 28 is notified. The driver 28 turns off the switch MOSFET 16 to cut off the current when it knows that an overcurrent has flown from the detection signal from the overpower detection comparator 31. As a result, it is possible to prevent the overcurrent from flowing continuously.

Comparison reference voltage Vth of the comparator 31
The switching of the comparison voltage switching circuit 34 for generating is performed based on the detection signal of the power supply level detection circuit 21. Since the detection signal of the comparator 31 is output as a pulse, a latch circuit 35 including a flip-flop is provided. When the output of the latch circuit 35 is input to the NOR gate 36 provided between the drive pulse control circuit 26 and the driver 28, the comparator 31 detects overpower and the output changes to high level. Latched by the latch circuit 35,
The NOR gate 36 is controlled so that the drive pulse output from the drive pulse control circuit 26 is not supplied to the driver 28. The latch circuit 35 includes the drive pulse control circuit 2
When the reset signal RS supplied from 6 resets every cycle and the comparator 31 no longer detects the overpower state, the output of the latch circuit 35 is changed to the low level, and the NOR gate 36 supplies the drive pulse to the driver 28. Then, the switch MOSFET 16 is switched back to the normal operating state.

The error amplifier 22-multiplication circuit 23-
Since only the control system including the error amplifier 25 and the drive pulse control circuit 26 may cause an excessive stress to the related elements by operating so as to rapidly increase the boosted voltage at the rise of the power supply, In the PFC control IC 20 of the embodiment, when the voltage detected by the power supply level detection circuit 21 is low, the current control circuit 30 controls the operating current of the error amplifier 25 so that the DC voltage VRB rises at a desired speed. (Soft start)
It is configured to be able to. As a result, the stress applied to the element can be reduced, and the DC voltage VRB is less likely to overshoot. Further, in this embodiment, the DC voltage is controlled by the overshoot monitoring circuit 29.
When it is detected that VRB has reached a desired voltage, the soft start process by the current control circuit 30 is terminated to prevent overshoot. Here, the method of controlling the operating current of the error amplifier 25 may be the method of controlling the current value itself, or the method of controlling the total time for flowing the current by causing the current to flow intermittently.

The power supply monitor circuit 32 functions to stop the operation of the entire IC by stopping the operation of the reference voltage generation circuit 33 when the power supply terminal VCC is lower than a predetermined voltage such as 8V. At the same time, it has a function of generating a power-on reset signal RES when the power supply rises. In order to operate the IC until the power switch is turned on and the power supply device operates to supply a desired DC voltage to the power supply terminal VCC, the power supply terminal VCC
Is a starting resistance Rst for supplying a voltage from the AC power supply 10.
Are connected.

FIG. 3 shows a more specific configuration example of the power supply level detection circuit 21 and the threshold value switching circuit 34. The power supply level detection circuit 21 is connected to an external terminal VAC to which a voltage Vin ′ that is proportional to the input voltage Vin is input, and determines the input voltage Vin with two comparison voltages Vcp such as 2.55V and 2.95V. It is composed of a comparator. When the input voltage Vin 'is higher than 2.95V (Vin is about 160V), it is judged as 200V system, and when the input voltage Vin' is lower than 2.55V, it is judged as 100V system. It is configured. Two determination levels are provided in order to provide a hysteresis characteristic and prevent malfunction due to power supply fluctuation.

The threshold value switching circuit 34 includes resistors R11, R12, R13 connected in series between the positive power source voltage Vcc and the negative power source voltage Vee, and a MOSFET Q0 connected in parallel with the resistor R11. It consists of and. A
Power supply level detection circuit 2 when the C power supply 10 is a 200V system
The output of 1 is set to low level, the MOSFET Q0 is turned on, and the threshold value switching circuit 34 sets the resistors R12 and R13.
The voltage (Vcc-Vee) is divided by the ratio of, for example, -0.25
A voltage such as V is supplied to the overpower detection comparator 31 as the comparison reference voltage Vth. On the other hand, the AC power source 10 is 1
In the case of the 00V system, the output of the power supply level detection circuit 21 is set to the high level, the MOSFET Q0 is turned off, and the threshold value switching circuit 34 sets the sum of the resistances R11 and R12 and the resistance R1.
A voltage such as -0.5 V obtained by dividing the voltage (Vcc-Vee) by the ratio of 3 is supplied to the overpower detection comparator 31 as the comparison reference voltage Vth. As a result, the comparison reference voltage Vth
In the conventional method of overpower detection that does not switch the
Although the overpower limiter level for stopping the switching control is increased in proportion to the power supply voltage Vin as indicated by the ▲ mark, in the overpower detection of the circuit of this embodiment for switching the comparison reference voltage Vth, As indicated by ■, 160
It gradually increases up to near V, but the power supply level detection circuit 21
The overpower limiter level is about 60 when the output of
The operation is performed from 0 W to 300 W at once, and gradually increases again. As a result, 100V system and 2
It becomes possible to avoid an extreme difference in the overpower limiter level from the 00V system.

By the way, the threshold value switching circuit 34 of the embodiment of FIG. 3 is an example of a circuit for generating the negative comparison reference voltage Vth. A negative voltage generating circuit such as a pump is required separately. Therefore, in FIG. 4, the comparison reference voltage Vth is set to a positive fixed voltage such as 1 V, and instead the detected voltage is level-shifted according to the AC power supply, thereby eliminating the need for a negative voltage generation circuit. Here is an example: In this embodiment, the overpower detection comparator 31
1V as a comparison reference voltage Vth to the non- inverting input terminal of
A fixed voltage such as the above is applied, and on the inverting input terminal side of the comparator 31, resistors R11 to R11 connected in series between the power supply voltage Vcc and the external terminal PFC-CS for overvoltage monitoring.
MOSFET connected in parallel with R13 and resistor R11
A level shift circuit 37 composed of Q0 and Q0 is connected, and a voltage divided by the ratio of the resistors forming the level shift circuit 37 is input. Then, the resistor R11
The output of a power supply level detection circuit 21 for detecting whether the AC power supply 10 is 100V system or 200V system is applied to the gate terminal of a MOSFET Q0 connected in parallel with
Power supply level detection circuit 2 when the C power supply 10 is a 100 V system
The output of 1 is set to low level to turn on the MOSFET Q0, and when the AC power supply 10 is 200 V system, the output of the power supply level detection circuit 21 is set to high level to set the MOSFET to 0.
Q0 is turned off.

As a result, even if the potential of the external terminal PFC-CS is the same, the AC power source 10 has a voltage level of
The output voltage of the level shift circuit 37 is different when the V system is used (when the AC power supply 10 is the 200 V system, the voltage is lower than when the AC power supply 10 is the 100 V system). That is, the same result as when the comparison reference voltage Vth of the overpower detection comparator 31 is relatively changed is obtained. Specifically, A
The relative comparison reference voltage Vth is made lower when the C power supply 10 is a 200V system than when it is a 100V system. As a result, the overpower limiter level becomes almost the same level in the 100V system and the 200V system, and it is possible to prevent the overpower limiter level from becoming high when the AC power source is the 200V system as in the conventional case.

Next, a second embodiment of the present invention will be described. In the second embodiment, the comparison reference voltage Vth is continuously changed according to the level of the AC power source so that the overpower limiter level Plm becomes constant. FIG. 6 (A),
A schematic configuration of the second embodiment is shown in (B). Of these, FIG. 6A uses the same negative voltage as the previous example of the first embodiment, and FIG. 6B uses the same positive voltage as the latter example of the first embodiment. This is the configuration of the case. FIG. 6 (A)
Is the comparison reference voltage V of the overpower detection comparator 31 based on the input voltage Vin ′ of the input voltage monitor terminal VAC.
A threshold variable circuit 34 'that changes th according to the level of the input voltage Vin' is provided. Specifically, since the limiter level Plm of the overpower is represented by the equation (1) as described above, the following equation obtained by modifying the equation (1) | Vth | = Plm × (√2 × Rcs) / Vin ′ The threshold variable circuit 34 'is configured to generate the threshold Vth (see the symbol a in FIG. 9) that changes according to (2). Here, since the resistance value Rcs of the detection resistor RCS is constant, the overpower limiter level P is calculated from the equation (2).
To keep lm constant, the comparison reference voltage Vth is set to the input voltage Vi.
It can be seen that it should be changed in inverse proportion to n '. That is, the threshold variable circuit 34 'can be configured by a division circuit that divides a constant C by the input voltage Vin'.

On the other hand, in FIG. 6B, the comparison reference voltage Vth of the overpower detection comparator 31 is fixed (for example, 1 V).
Then, the detected voltage (voltage of the overpower monitor terminal PFC-CS) is changed to the input voltage Vi of the input voltage monitor terminal VAC.
A level conversion circuit 37 'for converting according to the level of n'is provided to convert the converted voltage into an overpower detection comparator 31.
It is designed to be input to the inverting input terminal of. The level conversion circuit 37 ′ includes a voltage-current conversion circuit 371 that converts the input voltage Vin ′ of the input voltage monitor terminal VAC into a current Iin, and a division circuit 372 that divides a constant (constant current) by the converted current Iin. , A current-voltage conversion circuit 37 for converting the operation result (Iout) of the division circuit into a voltage
3 and 3. The reason why the input voltage Vin ′ is once converted into the current and then converted into the voltage again is that the current divider circuit is easier to design than the voltage divider circuit.

FIG. 7 shows the level conversion circuit 3 of FIG. 6 (B).
7'shows a concrete circuit example. Voltage-current conversion circuit 37
Reference numeral 1 denotes a pnp transistor Q1 having an input voltage Vin 'of an input voltage monitor terminal VAC applied to its base, a constant current source I6 connected to an emitter of the transistor Q1, and a Q.
Npn emitter follower transistor Q2 with its base connected to the emitter of 1 and its emitter resistor R3
And a transistor Q3 and a current mirror transistor Q4 connected to the collector of the transistor Q2,
Transistor Q5 connected in series with transistor Q4
And a current mirror transistor Q6, and a current Iin (= Vin '/ R3) proportional to the input voltage Vin' flows through the transistor Q6.

The division circuit 372 includes transistors Q7 to Q.
14 and constant current sources I1, I2, I3 and a resistor R4. Of these, the base of the transistor Q7 is connected to the collector of the transistor Q6, and the transistor Q8 is connected in series with the transistor Q6. Transistors Q7 and Q10 are emitter-coupled and connected to a constant current source I3, and a collector of Q7 is a pnp transistor Q1.
3 is connected, and the emitter voltage of the transistor Q9 connected to the collector of Q10 is applied to the base of the transistor Q8. A bias voltage generated by the emitter follower transistors Q11 and Q12 is applied to the base of the transistor Q10, the constant current sources I1 and I2 are connected to the emitters of Q11 and Q12, and the bases of the transistors Q9 and Q12 are connected to the reference. The voltage Vref is applied. The transistors Q13 and Q14 are current-mirror connected, and the resistor R4 is connected between the collector of Q14 and the contact point.

In the division circuit 372 having such a configuration,
Base-emitter voltage V of transistors Q7 to Q12
BE has the following equation: VBE7 + VBE8 + VBE9 = VBE10 + VBE11 + VBE12 (3). By the way, in the bipolar transistor circuit, the base-emitter voltage VBE has a property that it can be converted into a multiplication of the collector current Ic. (4) Here, Ic7 to Ic12 are Ic7 = Iou
t, Ic8 = Iin, Ic9 = Ic10, Ic11 = I
1 and Ic12 = I2. Then, by substituting these into the equation (4) and transforming, the following equation (5) Iout = I1 · I2 / Iin (5) is obtained. From this, it can be seen that Iout is inversely proportional to Iin.

The current-voltage conversion circuit 373 includes a pnp transistor Q15 whose base is connected to the collector of the transistor Q14, a constant current source I4 connected between the base of the transistor Q15 and the power supply voltage Vcc, and a transistor Q15. Constant current source I connected between the emitter and the power supply voltage Vcc
5, an emitter follower transistor Q16 to which the emitter voltage of Q15 is applied to the base, and resistors R1 and R2 connected in series between the emitter of Q16 and the overpower monitor terminal PFC-CS. . The constant current source I4 is for giving a predetermined offset to the output voltage Vout corresponding to the comparison reference voltage Vth such as 1V applied to the non- inverting input terminal of the comparator 31. Here, the emitter voltage of the transistor Q16 is VE
When it is set to 16, VE16 = R4 × (Iout + I4) (6)

When the apparent threshold value of the comparator 31 viewed from the overpower monitor terminal PFC-CS is VTH, VTH is given by the following equation: VTH = {Vth (R1 + R2) -VE16R1} / R2 ... It is represented by (7). Substituting equations (5) and (6) and Iin = Vin '/ R3 into equation (7), VTH = {Vth. (R1 + R2) -R4. (I1.I2.R3 / Vin' + I4) .R1} / R2 = Vth * (R1 + R2) / R2-R4 * R1 / R2 (I1 * I2 * R3 / Vin '+ I4) is obtained. From this equation, it can be seen that the apparent threshold value VTH of the comparator 31 is inversely proportional to the input voltage Vin '.

FIG. 10 shows how the apparent threshold value VTH of the comparator 31 changes as a result of the simulation of the overpower detection circuit when the level conversion circuit of the embodiment of FIG. 7 is applied. 11, the change of the overcurrent detection level is shown by a black square in FIG. 11, and the change of the overvoltage limit level is shown by a black triangle in FIG. Comparing FIG. 10 and FIG. 9, the change in the apparent threshold value VTH of the comparator 31 in the circuit of FIG.
It can be seen that the variation of the threshold value Vth generated by the threshold value variable circuit 34 'of FIG. Further, it can be seen from FIG. 12 that by applying the circuit of the embodiment of FIG. 7, the overvoltage limit level of the overpower detection circuit can be made substantially constant regardless of the change of the input voltage Vin, as indicated by the symbol A.

FIG. 8 shows a modification of the level conversion circuit of the embodiment of FIG. In this modification, in the circuit of FIG.
A transistor Q17 whose base is biased by the reference voltage Vref is provided between the transistors Q7 and Q13 that form the divider circuit 372. In the circuit of FIG. 7 in which the transistor Q17 is not provided, the collector voltages of the transistors Q7 and Q10 that operate differentially are different, so that the respective emitter currents of Q7 and Q10 are different due to the Early effect. By providing the transistor Q17 as described above, the collector voltages of Q7 and Q10 become the same (Vref-VBE), and Q7 and Q10
You can cancel the early effect with 10.
As a result, the advantage that the power supply voltage dependency of the output voltage Vout and thus the threshold value VTH of the comparator 31 can be improved is obtained.

As another modification, all the transistors forming the division circuit in the circuits of FIGS. 7 and 8 are p-type.
It is also possible to use an np transistor. Then, in this case, the collector of the transistor corresponding to Q7 in FIG. 7 is directly connected to the base of the transistor corresponding to Q15, so that Q6, Q
It is possible to realize a circuit having a small number of constituent elements which does not require transistors corresponding to 8, Q13 and Q14.

Although the invention made by the present inventor has been specifically described based on the embodiments, the invention is not limited to the above embodiments, and various modifications can be made without departing from the scope of the invention. Needless to say. For example, in the second embodiment, an external terminal VAC to which a voltage proportional to the voltage output from the rectifier circuit is applied is provided, and the overcurrent detection comparator 31 of the overcurrent detection comparator 31 is based on the voltage Vin ′ input to the external terminal. The threshold variable circuit 34 'or the level conversion circuit 37' for changing the threshold is provided. However, when an external terminal to which a current proportional to the current output from the rectifier circuit is input, The threshold value of the overpower detection comparator 31 may be changed according to the input current of the terminal.

In the embodiment, the NOR gate 36 is controlled by the output of the overpower detection comparator 31 to cut off the control pulse supplied from the drive pulse control circuit 26 to the switch MOSFET 16. The output of the detection comparator 31 may be directly supplied to the drive pulse control circuit 26 so that the control pulse supplied to the switch MOSFET 16 is not generated. The position at which the overcurrent detection resistor Rcs is provided is not limited to the example, and may be another position. Furthermore, in the embodiment, the present invention has been described as applied to an IC having a PFC control function, but it can also be applied to an overpower detection circuit in a PWM control IC that constitutes a switching power supply circuit. Moreover, the present invention may be applied to a booster circuit. In this case, it is possible to apply a booster circuit having a substantially constant power to different input voltages. Of course, the booster circuit may be built in the IC.

[0036]

The effects obtained by the representative one of the inventions disclosed in the present application will be briefly described as follows. That is, according to the present invention, overpower exceeding a desired limit level can be detected in a power supply control IC used in a switching power supply device, and a highly reliable power supply device can be realized. Further, by applying the present invention, it is possible to realize a switching power supply control IC having high versatility that can detect a predetermined level of overpower even when used for different AC power supplies.

[Brief description of drawings]

FIG. 1 is a block diagram showing an example of a PFC control IC suitable for application of an overpower detection circuit according to the present invention and a schematic configuration of a power supply device using the same.

FIG. 2 is a waveform diagram showing a voltage waveform after rectification in the power supply device of the embodiment, an output current waveform from a multiplication circuit in the PFC control IC, and a voltage waveform at an overpower detection external terminal.

FIG. 3 is a circuit configuration diagram showing a first embodiment of an overpower detection circuit in a PFC control IC.

FIG. 4 is a circuit configuration diagram showing another configuration example of the overpower detection circuit.

FIG. 5 is a characteristic diagram showing a difference in overpower detection level between the overpower detection circuit of the first embodiment and the conventional overcurrent detection circuit.

FIG. 6 is a circuit configuration diagram showing a second embodiment of an overpower detection circuit.

FIG. 7 is a circuit diagram showing a specific example of an overpower detection circuit according to a second embodiment.

FIG. 8 is a circuit diagram showing a modification of the overpower detection circuit of the second embodiment.

FIG. 9 is a graph showing the input voltage dependency of the threshold value of the comparator in the overpower detection circuit of the embodiment of FIG. 6 (A).

FIG. 10 is a graph showing the input voltage dependence of the apparent threshold value of the comparator in the overpower detection circuit of the embodiment of FIG. 6 (B).

FIG. 11 is a graph showing the input voltage dependence of the overcurrent detection level in the overpower detection circuit of the second embodiment.

FIG. 12 is a characteristic diagram showing a difference in overpower detection level between the overpower detection circuit of the first embodiment and the conventional overcurrent detection circuit.

[Explanation of symbols]

11 Rectifier circuit 12 Smoothing capacitor 13 Switching regulator 14 Booster circuit 16 Switching element (switch MOSFET) 17 Booster coil 18 Boost diode 20 PFC control IC 21 Power supply level detection circuit 22 1st differential amplifier (error amplifier) 23 multiplication circuit 24 Gain selection circuit 25 Second differential amplifier (error amplifier) 26 Drive pulse control circuit 27 Oscillation circuit 28 pulse driver 31 Overpower detection comparator 34 Threshold switching circuit 37 Level conversion circuit 371 voltage-current conversion circuit 372 division circuit 373 current-voltage conversion circuit

─────────────────────────────────────────────────── ─── Continuation of front page (51) Int.Cl. ⁷ identification code FI theme code (reference) H02H 7/20 H02H 7/20 D H02M 7/12 H02M 7/12 Q (72) Inventor Makoto Chiba Tokyo 5-20-1 Kamimizuhoncho, Kodaira-shi F-Term in the semiconductor group, Hitachi, Ltd. Semiconductor group (reference) 5G004 AA04 AB02 BA04 DA02 DC03 DC04 EA01 5G053 AA01 BA01 BA04 CA01 DA01 EA09 EB05 EC03 5H006 AA02 AA05 CA02 CA07 CB01 CB08 DA04 DC02 DC05 FA02 5H730 AA18 AA20 AS01 AS04 BB14 BB57 CC01 DD04 FD01 FD11 FG05 FG25 FG26 XX04 XX16 XX26 XX35 XX43

Claims (17)

[Claims] 1. A power supply device for switching a current flowing through a coil connected to a rectifier circuit receiving an AC power source to convert and outputting a voltage output from the rectifier circuit, and turning on the current flowing through the coil. , A semiconductor integrated circuit for power supply control that outputs a control pulse of a switching element to be turned off, and compares the voltage converted by the resistance for overcurrent detection with a reference voltage and When the converted voltage exceeds the reference voltage, a comparator for determining that an overcurrent has flown and outputting a signal for stopping the control pulse supplied to the switching element, and a threshold value for the determination of the comparator are A semiconductor device for power supply control, comprising: a conversion unit that performs conversion so as to be changed according to a voltage output from the rectifier circuit. Circuit. 2. A voltage discriminating circuit for discriminating whether the AC power supply is a first voltage system or a second voltage system higher than the first voltage system, and the converting means is provided by the voltage discriminating circuit. The threshold value of the comparator is configured to be switched to a first threshold value or a second threshold value higher than the first threshold value based on a signal. Semiconductor integrated circuit for power supply control. 3. A voltage discriminating circuit for discriminating whether the AC power supply is a first voltage system or a second voltage system higher than the first voltage system, wherein the converting means is a circuit for detecting the voltage from the voltage discriminating circuit. 2. The semiconductor integrated device for power supply control according to claim 1, wherein the voltage converted by the resistor for detecting the overcurrent is converted based on a signal so that the threshold value of the comparator relatively changes. circuit. 4. A first external terminal to which a voltage proportional to the voltage rectified by the rectifying circuit is applied, wherein the converting means is inversely proportional to the voltage input to the first external terminal. 2. The semiconductor integrated circuit for power supply control according to claim 1, wherein the threshold value of the comparator is continuously changed. 5. A first external terminal to which a voltage proportional to a voltage output from the rectifier circuit is applied, wherein the conversion means converts the voltage converted by the overcurrent detection resistor into the first external terminal. 2. The power source control device according to claim 1, wherein the threshold value of the comparator is continuously converted so as to relatively change in accordance with a voltage input to an external terminal. Semiconductor integrated circuit. 6. The voltage converting means converts the voltage input to the first external terminal into a current, and the dividing means outputs a current inversely proportional to the output current of the voltage converting current circuit. 6. The semiconductor integrated circuit for power supply control according to claim 4, comprising a circuit and a current-voltage conversion circuit for converting an output current of the division circuit into a voltage. 7. A second external terminal to which a current proportional to the current output from the rectifier circuit is input, a third external terminal to which a feedback voltage of an output converted voltage is input, and an input to the third external terminal. A differential amplifier that outputs a voltage according to a voltage, a multiplication circuit that outputs a current according to an input current of the second external terminal and an output voltage of the differential amplifier, and an output current of the multiplication circuit 7. A fourth external terminal is provided, and the pulse control circuit is configured to control the pulse width of the control pulse supplied to the switching element according to the output current of the multiplication circuit.
The semiconductor integrated circuit for power supply control according to any one of claims 1 to 4, wherein the comparator is configured to receive the voltage of the fourth external terminal as one input. 8. A rectifier circuit for converting an alternating current supplied from an alternating current power source into a direct current, and a coil connected to the rectifier circuit.
A switching element that turns on and off a current flowing through the coil, and the power supply control semiconductor integrated circuit according to claim 1 that controls the switching element. A power supply device characterized in that it is configured to boost and output the voltage with the coil. 9. The fourth external terminal is connected to the rectifier circuit, and the overcurrent detection resistor is connected as an external element between the fourth external terminal and a ground potential. The power supply device according to claim 8. 10. A PFC used in a power supply device for converting a rectified voltage by a switching operation of a coil and a switch element coupled to the coil to charge a capacitor.
A semiconductor integrated circuit for control, comprising means for inhibiting the switching operation of the switch element when the product of the current supplied to the coil and the rectified voltage exceeds a predetermined value. Characteristic PFC control semiconductor integrated circuit. 11. A PFC used in a power supply device for converting a rectified voltage by a switching operation of a coil and a switch element coupled to the coil to charge a capacitor.
A semiconductor integrated circuit for control, comprising: a detection circuit that detects the value of the rectified voltage; and a voltage according to the current supplied to the coil, which is compared with a threshold value that changes according to the detection output of the detection circuit. However, the PF has means for inhibiting the switching operation of the switch element when the voltage is equal to or higher than a threshold value.
C control semiconductor integrated circuit. 12. The PFC according to claim 11, wherein in the comparison in the means, the change in the threshold value due to the detection output is a change in the reference value to be compared.
Control semiconductor integrated circuit. 13. The PFC control according to claim 11, wherein, in the comparison in the means, the change in the threshold value due to the detection output is a change in the voltage according to the current supplied to the coil. Semiconductor integrated circuit. 14. A semiconductor integrated circuit used in a voltage conversion circuit for converting a voltage by a switching operation of a coil and a switch element coupled to the coil, the detection circuit detecting a value of the voltage, and the coil. A voltage corresponding to the current supplied to the comparator is compared with a threshold value that changes according to the detection output of the detection circuit, and when the voltage is equal to or higher than the threshold value, the switching element is prohibited from switching operation. A semiconductor integrated circuit characterized by the above. 15. The semiconductor integrated circuit according to claim 14, wherein in the comparison in the means, the change in the threshold value due to the detection output is a change in the reference value to be compared. 16. The semiconductor integrated circuit according to claim 14, wherein in the comparison in the means, the change in the threshold value due to the detection output is a change in the voltage according to the current supplied to the coil. . 17. A semiconductor integrated circuit used in a voltage conversion circuit for converting a voltage by a switching operation of a coil and a switch element coupled to the coil, the product being a product of a current supplied to the coil and the voltage. However, the semiconductor integrated circuit has means for prohibiting the switching operation of the switch element when the value exceeds a predetermined value.