JP2003125597A - Controller for electric vehicle - Google Patents

Abstract

PROBLEM TO BE SOLVED: To eliminate variation in the tone of magnetic noise in a two level inverter for electric vehicle and to control the output voltage substantially continuously over the entire region thereof. SOLUTION: The controller for electric vehicle comprises means for causing an electric vehicle to make a transition of a plurality of control modes, i.e., a pulse width modulation control mode, an asynchronous overmodulation control mode and a synchronous single pulse control mode, sequentially in this order when the electric vehicle is operated from a low speed region to a high speed region. In the asynchronous overmodulation control mode between the pulse width modulation control mode and the single pulse control mode, the voltage pulse period in the vicinity of the peak of an output voltage basic wave becomes longer than that in the vicinity of zero-cross as the output voltage increases and a plurality of voltage pulses are outputted asynchronously with the period of the output voltage basic wave. In the asynchronous overmodulation control mode, switching frequency Fsw of the switching element in a power converter decreases gradually from a predetermined constant frequency Fc in the pulse width modulation control mode to the starting frequency of the single pulse control mode.

Description

Description: BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a control device for an electric vehicle, and more particularly to a control of a PWM (pulse width modulation) inverter for supplying an AC output to a motor driving the electric vehicle. About technology. 2. Description of the Related Art "Electric Vehicle Science" 199, published by Electric Vehicle Research Society
An example of an inverter modulation method is described in FIG. 1, page 14 of the April, March issue article, "Evaluating Recent Inverter Control Technologies", page 14. As shown in FIG. 2, the inverter of the electric railway vehicle performs control to keep the ratio between the magnitude of the output voltage and the fundamental wave frequency constant when the output voltage fundamental wave frequency is low. When the output voltage fundamental wave frequency increases and the magnitude of the output voltage becomes maximum, the frequency control is performed while maintaining the maximum value voltage (the region in which this control is performed is a constant voltage variable region). Let's call it the frequency domain). In the variable voltage variable frequency region, in order to adjust the output voltage by pulse width modulation control, a multi-pulse mode in which a half cycle of the output voltage is composed of a plurality of voltage pulses is used. On the other hand, in the constant voltage variable frequency region, a one-pulse mode in which a half cycle of the output voltage is constituted by a single pulse is used in order to maximize the voltage utilization rate and reduce the size of the device. [0003] A conventional inverter using a GTO thyristor as a switching element (hereinafter, referred to as a switching element).
As shown in FIG. 3, a multi-pulse mode of a pulse number switching system in which the number of pulses included in one cycle is switched and gradually reduced as the output voltage fundamental frequency increases, as shown in FIG. Was. This is because the upper limit of the switching frequency of the GTO thyristor is several hundred Hz. In this method, since the switching frequency becomes discontinuous when the number of pulses is switched, there is a problem that the timbre of the magnetic noise is generated due to the switching of the number of pulses, which is annoying. Further, in the GTO inverter, between the output voltage in the three-pulse mode including three voltage pulses in a half cycle of the output voltage and the output voltage in the one-pulse mode, about 10% depending on the limitation of the minimum off time of the GTO thyristor. There is a jump, and there is a problem that the torque generated by the motor fluctuates when switching between the three-pulse mode and the one-pulse mode. An object of the present invention is to control a motor for driving an electric vehicle using an AC output of a two-level inverter that controls the magnitude of an output voltage from zero to a maximum voltage by a combination of a multi-pulse mode and a one-pulse mode. In this case, the discontinuity of the switching frequency of the two-level inverter is eliminated, the timbre of the annoying magnetic noise is eliminated, the gap between the output voltage in the multi-pulse mode and the output voltage in the one-pulse mode is reduced, and the entire output voltage is almost continuous. Control. In order to solve the above problems, when an electric vehicle is operated from a low speed range to a high speed range, a pulse width modulation control mode as a plurality of control modes and an asynchronous over modulation control Mode and synchronous one-pulse control mode,
In this order, the palace width modulation control mode outputs a plurality of voltage pulses at a constant cycle in a half cycle of the output voltage fundamental wave asynchronously with the cycle of the output voltage fundamental wave. In the power converter, the switching frequency of the switching element is set to a predetermined constant frequency, and in the one-pulse control mode, a single voltage pulse is applied to a half cycle of the output voltage fundamental wave in synchronization with the cycle of the output voltage fundamental wave. Output, the switching frequency of the switching element of the power converter in the mode is set to be the frequency of the output voltage fundamental wave, and the asynchronous overmodulation control mode interposed between the palace width modulation control mode and the one-pulse control mode includes: As the output voltage increases, the voltage pulse period near the peak of the output voltage fundamental wave is longer than near the zero cross,
In addition, a plurality of voltage pulses are output asynchronously with the cycle of the output voltage fundamental wave, and the switching frequency of the switching element of the power converter in the mode is changed by one pulse from the predetermined constant frequency in the pulse width modulation control mode. In the control mode, the frequency is gradually decreased to the frequency at which the one-pulse control mode starts. FIG. 1 is a block diagram showing an embodiment of the present invention; FIG.
7 will be described. The configuration of the inverter of the present invention in the PWM mode is as shown in FIG. It operates in the bipolar mode (pulse width modulation control mode) in the low output voltage range, the overmodulation mode (overmodulation control mode) in the high output voltage range, and the one-pulse mode (one-pulse control mode) in the maximum output voltage range. Here, the multi-pulse mode has a bipolar mode and an overmodulation mode. FIG. 1 is a configuration diagram showing an embodiment of the present invention, and is an example in which a voltage type two-level inverter is used as a control converter of an induction motor for driving an electric vehicle. In the figure, 6 is an induction motor, 5 is a two-level three-phase PWM inverter for driving it, 9 is a DC overhead wire serving as a power supply for the inverter, and 7 and 8 are filter reactors and capacitors on the inverter DC input side. The multi-pulse generating means 2, one-pulse generating means 3, and PWM mode selecting means 4 shown in FIG. 1 are configured to integrate the output voltage command E * of the inverter and the frequency command Fi * of each phase obtained by integrating with the integrator 1. The control signal for the inverter is generated based on the phase θx of the output voltage fundamental wave (the subscript x is a general term for the subscript representing the phase, that is, any one of u, v, and w). Of the control signals of the inverter, S1x, S2x, S
x is called a switching function, and is defined as 1 when the positive arm of the inverter is on and 0 when the negative arm is on. First, a method of generating a control signal for an inverter will be described. FIG. 5 shows an example (one phase) of the multi-pulse generating means 2 of FIG. Here, the switching functions of the bipolar mode and the overmodulation mode are generated by the same means. The output voltage command → modulation rate conversion means 21 obtains the modulation rate A, that is, the amplitude of the modulated wave from the output voltage command E *. Assuming that the carrier amplitude is 1, 0 ≦ A ≦ in the bipolar mode.
1, A> 1 in the overmodulation mode. In order to make the magnitude of the output voltage fundamental wave coincide with the voltage command, E * and A are made to correspond to each other in the bipolar mode (Equation 1) and in the overmodulation mode (Equation 2). (Equation 1) (Equation 2) In the function y = sin (x) 22, sin of the phase of the output voltage fundamental wave (equivalent to the phase of the modulated wave) θx is obtained. A product obtained by multiplying this by the modulation rate A is a modulated wave ax. The modulated wave ax and the carrier frequency (equivalent to the switching frequency in the bipolar mode) Fc are given to the switching function calculating means 24 to obtain the switching function S1x. The switching function calculating means 24 generates a carrier wave, which is a triangular wave having an amplitude of 1 and a frequency Fc, and compares it with the value of the modulated wave to generate a switching function. Further, the switching function may be obtained by calculation from the modulated wave ax and the pulse interval without using the triangular wave. FIGS. 6 and 7 show examples of the waveforms of the switching function in the bipolar mode and the overmodulation mode obtained by the triangular wave comparison. In the inverter device of the present invention, a device capable of switching at several kHz, such as an IGBT or a large-capacity power transistor, is used as a switching element (hereinbelow, collectively referred to as an IGBT inverter). And the carrier are asynchronous. In the bipolar mode shown in FIG. 6, since 0 ≦ A ≦ 1, the carrier 242 corresponds to the switching function 243, and the carrier 242 and the modulation wave 241 are not synchronized. Further, since A> 1 in the overmodulation mode shown in FIG. 7, a switching function 246 of a wide pulse is obtained in a portion where A exceeds 1, and a comparison between the carrier 245 and the modulation wave 244 is performed in other portions. A switching function 246 is obtained. Also in this overmodulation mode, the carrier 245 and the modulated wave 244 are generated asynchronously. As described above, in the figure, a method of obtaining a switching function by comparing a carrier wave and a modulated wave is shown for the sake of understanding, but the switching function can be obtained by calculation from the modulated wave ax and the pulse interval. As a result, the switching frequency becomes constant in the bipolar mode, and can gradually approach the switching frequency in the one-pulse mode described below in the overmodulation mode. In the multi-pulse mode, since the modulation wave and the carrier are asynchronous, the carrier frequency needs to be sufficiently higher than the modulation wave frequency.
It is desirable to be higher than about 0 times. FIG. 8 shows an example of the waveform of the switching function generated by the one-pulse generating means of FIG. The value of the switching function S2x is 1 when the sign of the fundamental wave 31 (of any amplitude) of the output voltage is positive, and S when the sign is negative.
The value of 2x is 0. Next, the combination of the multi-pulse mode and the one-pulse mode for controlling the high output voltage range will be described. No.1 Proceedings of the National Convention of the Institute of Electrical Engineers of Japan in 1991.
06 “Voltage-type three-phase PWM inverter overmodulation control method”
There is. According to this, the operation of the 6-step inverter, ie, 1
It is stated that this is a pulse mode operation. However, when the one-pulse mode is realized as an extension of the overmodulation mode (the one-pulse mode is realized by making the modulation rate extremely large), the following inconvenience occurs. First, the point at which the overmodulation mode and the one-pulse mode are switched depends on the switching frequency and cannot be set arbitrarily. Second, when the modulated wave in the overmodulation mode and the carrier wave are asynchronous (hereinafter referred to as asynchronous PWM), the modulated wave near the boundary between the overmodulation mode and the one-pulse mode is affected by the turn-on and turn-off times of the element. A pulse near the zero cross may or may not appear. As a result, an imbalance occurs between the positive and negative of the output voltage, and a beat phenomenon occurs in which low-frequency pulsation is superimposed on the load current of the inverter.
Third, in the overmodulation mode, as shown in FIG. 7, the output voltage waveform (equivalent to a switching function waveform described later) has a modulated wave (equivalent to an output voltage fundamental wave) with a uniform pulse interval near the zero cross. That is, the pulse is divided into a portion where the pulse generation period is uniform (equal interval pulse) and a portion of the wide pulse centering on the peak of the modulated wave. Switching can occur. In this case, the load current of the inverter is disturbed,
The switching element may be destroyed due to the overcurrent, and the generated torque of the motor may fluctuate significantly. In order to solve these problems, a voltage for switching between the overmodulation mode and the one-pulse mode (hereinafter, referred to as a transition voltage) and a phase of the output voltage fundamental wave (hereinafter, referred to as a transition phase). Manage. First, management of the transition voltage will be described. When the transition voltage is set and the overmodulation mode and the one-pulse mode are switched at the boundary, it is desirable that the set value of the transition voltage be as close as possible to the output voltage of the one-pulse mode, that is, a value close to 100%. This is because the smaller the difference from the maximum value of the output voltage in the overmodulation mode, the smaller the fluctuation of the generated torque of the motor at the time of switching. However, in the asynchronous PWM, the width of each voltage pulse included in one cycle of the fundamental wave of the output voltage is different in each cycle, and the zero crossing of the fundamental wave of the output voltage as the output voltage approaches 100% in the overmodulation mode. When the number of pulses in the vicinity decreases, this effect becomes apparent, an imbalance occurs between the positive and negative of the output voltage, and a beat phenomenon occurs in the load current of the inverter. FIG. 9 shows an example of this state. FIG. 10 shows an example of the relationship between the average number of pulses near the zero cross of the output voltage fundamental wave and the current pulsation due to the beat phenomenon. As shown in FIG. 7, the absolute value of the modulated wave is 1.0.
Since the following portions correspond to equally spaced pulses, the average number of pulses is given by the equation shown in (Equation 3). The current pulsation rate was defined by (Equation 4). [Equation 3] (Equation 4) From FIG. 10, if at least one pulse is not secured near the zero cross of the output voltage fundamental wave, the low frequency pulsation of the load current of the inverter due to the beat phenomenon becomes extremely large. Therefore, the set value of the transition voltage is set to a value that secures at least one or more voltage pulses near the output voltage fundamental wave zero cross. Since this value depends on the output voltage fundamental wave frequency Fi * and the carrier frequency Fc in the multi-pulse mode, means for calculating these values by calculation may be provided.
It may be set by calculating in advance from the upper limit of the output voltage fundamental wave frequency Fi *. Next, management of the transition phase will be described. Depending on the phase of the output voltage fundamental wave at the time of switching between the overmodulation mode and the one-pulse mode, the state of the transient fluctuation of the load current of the inverter and the torque generated by the motor immediately after the switching differs. FIG. 11 shows an example of the current fluctuation. FIG.
FIG. 12 shows a case where all three phases are switched at 0 ° with the phase of the U-phase output voltage fundamental wave as shown in FIG. 12, and a transient fluctuation is observed in the current immediately after the switching. On the other hand, FIG.
As shown in FIG. 13, the phase of the U-phase output voltage fundamental wave is 9
This is a case where the three phases are switched at once at 0 °, and there is almost no transient fluctuation of the current. FIG. 14 shows an example of the relationship between the phase of the output voltage fundamental wave (U-phase reference) and the transient current fluctuation when switching from overmodulation mode to one-pulse mode collectively in three phases.
Here, the current fluctuation rate is defined by (Equation 5). (Equation 5) In FIG. 14, the current fluctuation rate increases every 60 ° in the phase of the output voltage fundamental wave. This is the case where the overmodulation mode and the one-pulse mode are switched when any one of the three phases is an equally-modulated pulse in the overmodulation mode. Since the unbalance increases, the transient current fluctuation also increases.
Therefore, as shown in FIG. 15, by setting the transition phase to a portion where all phases become wide pulses in the overmodulation mode, transient current and torque fluctuations can be suppressed. Here, in order to switch overmodulation mode and one-pulse mode collectively in three phases, there must be a section in which the output voltage of the overmodulation mode in all three phases becomes a wide pulse. For this purpose, the intersection (U
30 °, 90 °, 150 ° with respect to the phase of the phase modulated wave
{, 210}, 270 °, 330 °), the absolute value of the modulated wave must be greater than 1.0. Considering the case of 30 °, au = Asin30 ゜> 1.0 and A> 2. In the overmodulation mode, the relationship between the modulation factor A and the output voltage E * is given by (Equation 2), so E *> 95. Must be 6%. Therefore, in order to switch overmodulation mode and one-pulse mode in a three-phase mode, the transition voltage is greater than 95.6%, and at least one voltage pulse is secured near the zero cross of the output voltage fundamental wave by overmodulation. It becomes. FIG. 16 shows an example of the configuration of the PWM mode selecting means 3 for realizing the management of the transition voltage and the transition phase. The mode selection command generation means 32 compares the transition voltage Ec set in the transition voltage means 31 with the voltage command E *, and generates a mode selection command Mc indicating whether to select the multi-pulse mode or the one-pulse mode. Here, the mode selection command Mc is determined based on the output voltage command E *. However, since the output voltage command E * uniquely corresponds to the modulation factor A, the modulation factor Ac corresponding to the transition voltage is set in advance. Alternatively, the mode selection command Mc may be generated by comparing this with the modulation factor A. In the variable voltage variable frequency region, the output voltage command and the output voltage fundamental frequency also uniquely correspond to each other. Therefore, an output voltage fundamental wave frequency Fic corresponding to the transition voltage is set in advance, and this and the frequency command Fi * are set.
May be compared to generate a mode selection command Mc. The transition phase management unit 44 refers to Mc, and when the mode needs to be switched, compares the phase θx of the output voltage fundamental wave with the transition phase θc set in the transition phase setting unit 43, and determines that θx is θ
If it has reached c, the mode selection signal M is switched. The mode selection switches 45, 46, and 47 select either the output S1x of the multi-pulse generation means or the output S2x of the one-pulse generation means according to the mode selection signal M, and determine the switching function Sx. The transition phase may be managed by the following method. Take the absolute value of the modulated wave of each phase,
If all three phases are greater than 1.0, then all phases are at the point of the overmodulated wide pulse.
Therefore, at such time, the outputs of the multi-pulse generating means and the one-pulse generating means are switched. As described above, the gap between the output voltages in the multi-pulse mode and the one-pulse mode is reduced from about 10% in the conventional GTO inverter to about 1-2%, and the magnitude of the output voltage is reduced from zero to the maximum voltage. Control almost continuously until
Further, it is possible to configure a two-level inverter capable of smoothly performing switching without changing the current or the generated torque of the electric motor when switching between the multi-pulse mode and the one-pulse mode. The relationship between the output voltage fundamental wave frequency and the switching frequency in the present invention is as shown in FIG.
There is no large discontinuity as in the conventional modulation method of the inverter in FIG. 3, and discontinuous timbre change of magnetic noise can be eliminated. As described above, according to the present invention,
Discontinuous magnetic noise of a two-level inverter when controlling a motor driving an electric vehicle using the AC output of an inverter that controls the magnitude of the output voltage from zero to the maximum voltage by a combination of the multi-pulse mode and the one-pulse mode In addition to the above, it is possible to eliminate a significant change and to control the entire output voltage range almost continuously. Further, by adopting the overmodulation mode as the control mode, there is no change in the current or the generated torque of the electric motor driving the electric vehicle at the time of switching between the multi-pulse mode and the one-pulse mode. A level inverter can be configured.

BRIEF DESCRIPTION OF THE DRAWINGS FIG. 1 is a configuration diagram showing an embodiment of the present invention. FIG. 2 is a diagram showing operating characteristics of a vehicle inverter. FIG. 3 is a diagram showing an example of a conventional inverter modulation method. FIG. 4 is a diagram showing operating characteristics of the inverter according to the present invention. FIG. 5 is a diagram showing an example of a configuration of a multi-pulse generation unit. FIG. 6 is a diagram showing a modulation wave, a carrier wave, and a switching function in a bipolar mode. FIG. 7 is a diagram showing a modulated wave, a carrier wave, and a switching function in an overmodulation mode. FIG. 8 is a view showing a fundamental function of an output voltage and a switching function in a one-pulse mode. FIG. 9 is a diagram showing how a beat phenomenon occurs. FIG. 10 is a diagram showing a relationship between an average number of pulses near an output voltage fundamental wave zero cross and current pulsation. FIG. 11 is a view showing that the state of transient current fluctuation immediately after mode switching differs depending on the transition phase; FIG. 12 is a diagram showing the switching timing of FIG. FIG. 13 is a diagram showing the switching timing of FIG. 11 (b). FIG. 14 is a diagram showing a relationship between a transition phase and a transient current fluctuation immediately after mode switching. FIG. 15 is a diagram showing a transition phase settable section. FIG. 16 is a diagram showing an example of a configuration of a PWM mode selection unit. FIG. 17 is a diagram showing a relationship between an output voltage fundamental wave frequency and a switching frequency of the inverter according to the present invention. [Description of Signs] 1 ... Integrator, 2 ... Multi-pulse generating means, 3 ... 1-pulse generating means, 4 ... PWM mode selecting means, 5 ... 2-level three-phase P
WM inverter, 6 induction motor, 7 filter reactor, 8 smoothing capacitor, 9 DC overhead wire, 21 frequency command → modulated wave amplitude reference conversion means, 22 function y = sin
(x), 23: switching frequency, 24: switching function calculating means, 241, 244: modulated wave ax, 24
2, 245 carrier wave c, 243, 246 switching function S1x, 31 output voltage fundamental wave, 41 transition voltage setting means, 42 mode selection command generating means, 43 transition phase setting means, 44 transition phase management means , 45, 46, 4
7… Mode selection switch

Continuation of front page    (72) Inventor Yuto Suzuki             7-1-1, Omika-cho, Hitachi City, Ibaraki Prefecture             Hitachi, Ltd., Hitachi Laboratory (72) Inventor Mutsuhiro Terunuma             1070 Ma, Katsuta-shi, Ibaraki Pref.Hitachi, Ltd.             Inside Mito Factory F-term (reference) 5H007 BB06 CA01 CB02 CB05 CC01                       CC03 DB02 EA10 EA15                 5H115 PA05 PC02 PG01 PI03 PI29                       PU09 PV09 RB23 SE03                 5H576 AA01 BB04 CC01 DD04 EE13                       HA04 HB02 JJ01

Claims (1)

Claims: 1. A plurality of switching elements are controlled in accordance with a plurality of control modes to convert a DC voltage into a three-phase AC phase voltage having a variable voltage and a variable frequency. A two-level power converter that outputs a pulse and an AC output supplied from the power converter, and a control device for an electric vehicle including an electric motor that drives an electric vehicle. When operating up to, the pulse width modulation control mode as the plurality of control modes,
Means for sequentially shifting between the asynchronous overmodulation control mode and the synchronous one-pulse control mode in this order, wherein the Palace width modulation control mode is asynchronous with the cycle of the output voltage fundamental wave and is half of the output voltage fundamental wave. A plurality of voltage pulses are output at a constant cycle, and the switching frequency of the switching element of the power converter in the mode is set to a predetermined constant frequency. The one-pulse control mode is a mode of the output voltage fundamental wave. A single voltage pulse is output in a half cycle of the output voltage fundamental wave in synchronization with the cycle, so that the switching frequency of the switching element of the power converter in the mode becomes the frequency of the output voltage fundamental wave, In the asynchronous overmodulation control mode interposed between the palace width modulation control mode and the one-pulse control mode, the output voltage is large. The voltage pulse period near the peak of the output voltage fundamental wave is longer than the vicinity of the zero cross, and a plurality of voltage pulses are output asynchronously with the period of the output voltage fundamental wave. The switching frequency of the switching element of the pulse width modulation control mode, the predetermined constant frequency in the control mode, the one-pulse control mode in the one-pulse control mode the frequency of the start of the one pulse control mode is to be gradually reduced. Characteristic electric vehicle control device.