JP2002354807A - Switching power supply circuit - Google Patents

Abstract

PROBLEM TO BE SOLVED: To reduce dimensions and weight of a power supply circuit and to improve an AC-DC power conversion efficiency. SOLUTION: In a flyback transformer to which a switching output of a primary voltage resonance type converter is transmitted, a step-up winding NHV is wound so as to form a tight coupling state with a high voltage primary winding N0, and a low voltage secondary winding N2 is wound so as to form a loose coupling state with a low voltage primary winding N1. In the secondary side of the flyback transformer FBT, an AC voltage obtained in the step-up winding NHV is utilized to generate a DC high voltage, and an AC voltage obtained in the low voltage secondary winding N2 is utilized to generate a DC low voltage. In the flyback transformer, a coil comprising the low voltage primary winding is connected in series with a series circuit of the high voltage primary winding and a primary series resonance capacitor, and the series- connection circuit is connected in parallel with the low voltage primary winding.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a switching power supply circuit suitable for a large color television receiver having a high resolution and a cathode ray tube display device having a cathode ray tube as a projector device. is there.

[0002]

2. Description of the Related Art A cathode ray tube (hereinafter referred to as a CRT (Cathode-Ray Tu
For example, as a cathode ray tube display device having a high definition and a high image quality corresponding to a high definition television broadcast called HDTV (High Definition Television) or a digital television broadcast, the cathode ray tube display device having the same. . Among these devices, those corresponding to HDTV have a horizontal synchronizing signal frequency twice as high as that of a normal television receiver in order to realize a high resolution. Becomes Those corresponding to digital television broadcasting are NT
Under the SC system, the horizontal synchronization signal frequency is specified as 33.75 KHz. The high-voltage anode voltage supplied to the anode electrode of the CRT in such a video device is set to 30 KV or more.

As described above, as the cathode ray tube display device, a higher resolution has been promoted, and a device with a larger screen has been widely used. For this reason, for example, as a television receiver, in the case of the NTSC system, the horizontal synchronizing signal frequency is set to 31.5 KHz (= 1.
5.75 KHz × 2) and convert to double speed mode.
DTVs that are designed to be able to receive DTVs are also widely used. In a television receiver as described above, when a high-voltage DC output voltage is applied to the anode electrode of a CRT, for example, a horizontal synchronization signal frequency of 31.5
The high-voltage direct-current output voltage fluctuates between KHz and 33.75 KHz, so that the brightness and the raster size of the screen displayed on the CRT change. For this reason, it is indispensable to stabilize the power supply circuit for generating the anode voltage.

[0004] Against this background, the present applicant has proposed various switching circuits suitable for application to various cathode ray tube display devices. Therefore, a switching power supply circuit for video equipment configured based on the switching power supply circuit filed by the present applicant is shown in the circuit diagram of FIG.

In the power supply circuit shown in FIG. 1, first, a rectifier diode Di is applied to a commercial AC power supply AC.
1, Di2, smoothing capacitors Ci1, Ci2] in a connection configuration shown in the figure to form a voltage doubler rectifier circuit. This voltage doubler rectifier circuit generates a rectified smoothed voltage Ei (DC input voltage) corresponding to twice the AC input voltage VAC, at both ends of the smoothing capacitors Ci1 and Ci2 connected in series.

The switching converter which inputs and outputs the DC input voltage intermittently comprises a voltage resonance type converter having a single main switching element Q1 and performing a switching operation by a so-called single-ended system by a self-excited system. In this case, the main switching element Q
For 1, a high breakdown voltage bipolar transistor (BJT; junction type transistor) is used.

[0007] A series resonance circuit for driving self-oscillation, which is formed by a series connection circuit of a drive winding NB, a resonance capacitor CB and a base current limiting resistor RB, is connected to the base of the main switching element Q1. A clamp diode D inserted between the base of the main switching element Q1 and the negative electrode (primary ground) of the smoothing capacitor Ci2.
By D, a path for a clamp current flowing through the base-collector of the main switching element Q1 during the turn-on start period is formed. The collector of the main switching element Q1 is connected to one end of a primary winding N1 formed on the primary side of the insulating converter transformer PIT, and the emitter is grounded.

A primary side parallel resonance capacitor Cr is connected in parallel between the collector and the emitter of the main switching element Q1. The primary-side parallel resonance capacitor Cr has its own capacitance and the primary winding N1.
The primary side parallel resonance circuit of the voltage resonance type converter is formed by the leakage inductance L1 on the side. And
Although detailed description is omitted here, when the main switching element Q1 is turned off, the voltage V1 across the primary side parallel resonance capacitor Cr due to the operation of the primary side parallel resonance circuit is actually a sinusoidal pulse waveform. Thus, a voltage resonance type operation can be obtained.

The orthogonal control transformer PRT is a saturable reactor on which a resonance current detecting winding ND, a driving winding NB, and a control winding NC are wound. This orthogonal control transformer P
RT is provided for driving the main switching element Q1 and for stabilizing a DC high voltage obtained on the secondary side of the flyback transformer FBT. Although the illustration is omitted as the structure of the orthogonal control transformer PRT,
A three-dimensional core is formed by joining the ends of the magnetic legs of two double U-shaped cores having four magnetic legs. Then, the resonance current detecting winding ND and the driving winding NB are wound in the same winding direction with respect to two predetermined magnetic legs of the three-dimensional core.
And the control winding NC is wound in a direction perpendicular to the resonance current detection winding ND and the drive winding NB.

In this case, the resonance current detection winding ND of the orthogonal control transformer PRT is inserted in series between the positive electrode (DC input voltage line) of the smoothing capacitor Ci1 and the primary winding N1. Thereby, the main switching element Q
The switching output 1 is transmitted to the resonance current detection winding ND via the primary winding N1. Orthogonal control transformer PRT
In, the switching output obtained in the resonance current detection winding ND is induced in the drive winding NB via a transformer coupling, so that an alternating voltage as a drive voltage is generated in the drive winding NB. This drive voltage is output to the base of the main switching element Q1 as a drive current from a series resonance circuit (NB-CB) forming a self-excited oscillation drive circuit via a base current limiting resistor RB. Thus, the main switching element Q1 is connected to the series resonance circuit (NB
The switching operation is performed at the switching frequency determined by the resonance frequency of -CB). At the time of startup, the main switching element Q1 starts switching operation by a startup current flowing from the rectified smoothed voltage Ei to the base via the startup resistor Rs.

[0011] The insulating converter transformer PIT transmits the switching output of the main switching element Q1 to the secondary side. As shown in FIG. 6, the structure of the insulating converter transformer PIT includes an EE-type core in which E-type cores CR11 and CR12 made of, for example, a ferrite material are combined so that their magnetic legs face each other. For the center magnetic leg, the primary winding N
1 and the secondary winding N2 are wound separately. As shown in the figure, the gap G
Is formed. Thereby, loose coupling with a required coupling coefficient is obtained. The gap G can be formed by making the central magnetic legs of the E-shaped cores CR11 and CR12 shorter than the two outer magnetic legs. In addition, as the coupling coefficient k, for example, a loosely coupled state of k ≒ 0.8 is obtained, so that a saturated state is hardly obtained.

The winding start end of the primary winding N1 of the insulating converter transformer PIT is connected to the collector of the main switching element Q1 as shown in FIG. 4, and the winding end end is connected in series with the resonance current detection winding ND. Connected to the positive electrode of the smoothing capacitor Ci. A secondary winding N is provided on the secondary side of the insulating converter transformer PIT as a secondary winding.
2 are wound.

In this case, the winding start end of the secondary winding N2 is connected to the secondary side ground, and the winding end end is connected to the anode of the rectifier diode DO1. The voltage level of the half-wave rectifying / smoothing circuit including the rectifying diode DO1 and the smoothing capacitor CO1 is 110V to 140V.
(For example, 135 V) to obtain a DC output voltage EO1 for a horizontal deflection circuit.

In this case, a tap is provided for the secondary winding N2 as shown, and a half-wave rectifier composed of a rectifier diode DO2 and a smoothing capacitor CO2 is provided for this tap output as shown. By connecting the circuit, the secondary side DC output voltage EO2 of a required level (for example, 15V) lower than the secondary side DC output voltage EO1
Is also getting to get.

A secondary-side parallel resonance capacitor C2 is connected in parallel to the secondary winding N2. In this case, a secondary parallel resonance circuit is formed by the leakage inductance L2 of the secondary winding N2 and the capacitance of the secondary parallel resonance capacitor C2. Thus, the alternating voltage induced on the secondary side of the insulation converter transformer PIT becomes a resonance voltage, and a voltage resonance operation is obtained on the secondary side of the insulation converter transformer PIT.

That is, in the power supply circuit shown in FIG. 4, the primary side of the insulated converter transformer PIT is provided with a parallel resonance circuit for making the switching operation a voltage resonance type. Are provided. In the present specification, such a switching converter configured to operate with a resonance circuit provided for the primary side and the secondary side is also referred to as a “composite resonance type switching converter”.

An active clamp circuit 20 is provided on the secondary side of the insulating converter transformer PIT. This active clamp circuit 20 includes an auxiliary switching element Q2, a clamp capacitor CCL, and a clamp diode DD2. As the clamp diode DD2, a so-called body diode incorporated as a component in the switching element Q2 which is a MOS-FET is used. The drive circuit system for driving the auxiliary switching element Q2 includes a drive winding Ng, a capacitor Cg, and a resistor Rg.

In this case, a clamp diode DD2 is provided between the drain and the source of the auxiliary switching element Q2.
Are connected in parallel. In addition, the auxiliary switching element Q2
Is connected to the winding start end of the secondary winding N2 via a clamp capacitor CCL. The source of the auxiliary switching element Q2 is connected to the winding end of the secondary winding N2. That is, the active clamp circuit 20 is configured by connecting a clamp capacitor CCL in series to a switching circuit in which the auxiliary switching element Q2 // clamp diode DD2 is connected in parallel. The circuit thus formed is connected in parallel to the secondary winding N2 of the insulating converter transformer PIT. In this case, since the secondary parallel resonance circuit is formed by the secondary winding N2 and the secondary parallel resonance capacitor C2, the active clamp circuit 20 is connected to the secondary parallel resonance circuit. It can be seen that they are connected in parallel.

As a driving circuit system for the auxiliary switching element Q2, a series connection circuit of a resistor Rg, a capacitor Cg and a driving winding Ng is connected to the gate of the auxiliary switching element Q2, as shown in the figure. This series connection circuit forms a self-excited oscillation drive circuit for the auxiliary switching element Q2. Here, the drive winding Ng is formed so as to wind up the winding end end side of the primary winding N1 in the insulating converter transformer PIT.

The gate of the auxiliary switching element Q2 is also connected to the first control circuit 1A. In this case, the secondary DC output voltage EO1 is input to the first control circuit 1A as the detection voltage. The first control circuit 1A applies a control voltage having a variable level according to a change in the level of the input secondary-side DC output voltage EO1. Thus, the gate threshold voltage (bias) of the auxiliary switching element Q2 is varied, so that the ON period (conduction angle) within one switching cycle is variably controlled, that is, the PWM control is performed. Thus, the switching operation is performed. The current to be charged in the secondary side parallel resonance capacitor C2 forming the secondary side parallel resonance circuit is divided and flows through the clamp capacitor CCL, but the conduction angle of the auxiliary switching element Q2 is variably controlled. Then, the amount of current flowing through the clamp capacitor CCL2 changes, and accordingly, the amount of charging current to the secondary-side parallel resonance capacitor C2 changes. By changing the amount of charging current to the secondary parallel resonance capacitor C2 in this way, the level of the alternating voltage (parallel resonance voltage) obtained in the secondary parallel resonance circuit also changes. Then, by changing the parallel resonance voltage, the level of the secondary side DC output voltage EO1 is also variably controlled. In this way, the DC output voltage obtained on the secondary side of the insulating converter transformer PIT is stabilized.

A high-voltage generating circuit 40 surrounded by a dashed line.
Is composed of a flyback transformer FBT and a high-voltage rectifier circuit, and utilizes, for example, a CR
A DC high voltage corresponding to the anode voltage level of T is generated. For this reason, on the secondary side of the flyback transformer FBT, four to five sets of step-up windings NVH are divided and wound by so-called slit winding or interlayer winding as described later. In this case, the primary winding N0 and the boost winding NHV are wound so as to be tightly coupled. In addition,
In this case, the coupling coefficient k between the primary winding N0 and the boost winding NHV is set to k ≧ 0.95. On the secondary side of the flyback transformer FBT, the winding voltage V3 generated in the primary winding N0 is applied to the winding ratio (N0) between the boost winding NHV and the primary winding N0.
HV / N0).

In the case of the power supply circuit shown in FIG. 4, five sets of boost windings NHV are provided on the secondary side of the flyback transformer FBT.
1, NHV2, NHV3, NHV4, and NHV5 are wound independently of each other, and high-voltage rectifier diodes DHV1, DH
The anodes of V2, DHV3, DHV4 and DHV5 are connected. Then, the cathode of the high voltage rectifier diode DHV1 is connected to the positive terminal of the smoothing capacitor COHV, and the respective cathodes of the remaining high voltage rectifier diodes DHV2 to DHV5 are connected to the winding start ends of the step-up windings NVH1 to NHV4, respectively.

That is, on the secondary side of the flyback transformer FBT, [the boost winding NHV1, the high-voltage rectifier diode DHV
1], [boost winding NHV2, high voltage rectifier diode DHV2],
[Step-up winding NHV3, high-voltage rectifier diode DHV3], [Step-up winding NHV4, high-voltage rectifier diode DHV4], [Step-up winding N
HV5, high-voltage rectifier diode DHV5], that is, a so-called multisingle-type half-wave rectifier circuit in which five sets of half-wave rectifier circuits are connected in series.

Therefore, on the secondary side of the flyback transformer FBT, five sets of half-wave rectifier circuits
~ NHV5 rectifies the current induced and smoothes capacitor CO
An operation of charging the HV is performed, and a DC high voltage (anode voltage) EHV having a level corresponding to about five times the induced voltage induced in each of the boost windings NVH1 to NVH5 is provided across the smoothing capacitor COHV. (For example, 31.5 KV).

A series circuit composed of a resistor R1 and a resistor R2 is connected in parallel to both ends of the smoothing capacitor COHV. A DC high voltage EHV divided by these resistors R1 and R2 is supplied to a second control circuit. 1B. The second control circuit 1B is constituted by, for example, an error amplifier or the like, and controls the flow to the control winding NC of the orthogonal control transformer PRT in accordance with the level change of the DC high voltage EHV divided as described above. By changing the current (DC current) level, the drive winding N wound around the orthogonal control transformer PRT is changed.
The inductance LB of B is variably controlled. Thereby, the resonance condition of the series resonance circuit in the self-excited oscillation drive circuit for the main switching element Q1 formed including the inductance LB of the drive winding NB changes, and the switching frequency of the main switching element Q1 is varied. Operation.
By this operation, the DC output high voltage EHV output from the secondary side of the flyback transformer FBT is stabilized.

When an orthogonal control transformer PRT for variably controlling the inductance LB of the drive winding NB is provided as in the power supply circuit shown in FIG. 4, the main switching element Q1 is used to change the switching frequency. The period during which is turned off is kept constant, and the period during which it is turned on is variably controlled. That is, FIG.
In the power supply circuit shown in (1), as a constant voltage control operation, the switching impedance is variably controlled to perform resonance impedance control on the switching output, and at the same time,
Main switching element Q1 in switching cycle
It can be seen that the conduction angle control (PWM control) is also performed. The composite control operation is realized by a set of control circuit systems. In this specification,
Such a complex control is also called a “complex control method”.

As described above, in the power supply circuit shown in FIG. 4, by controlling the conduction angle of the auxiliary switching element Q2 of the active clamp circuit 20 according to the voltage level of the DC output voltage EO1, the secondary converter of the isolated converter transformer PIT is controlled. The DC output voltage on the secondary side is made constant. Further, the DC high voltage EHV is made to have a constant voltage by a complex control method for simultaneously controlling the switching frequency of the main switching element Q1 and the conduction angle according to the voltage level of the DC high voltage EHV.

In the switching power supply circuit shown in FIG. 4, an isolated converter transformer PIT is connected to a primary side voltage resonance type converter which operates by inputting commercial AC power.
And the flyback transformer FBT in the high voltage generation circuit 40
To obtain a low-voltage secondary-side DC output voltage EO1, EO2 and a DC high voltage EHV. For this reason, for example, as compared with a multi-stage high-voltage generation circuit that has been known for a long time, it is superior in terms of power conversion efficiency, circuit size reduction, and the like.

Here, an example of the structure of the flyback transformer FBT will be described with reference to the cross-sectional views of FIG. 7 and FIG. First, in the flyback transformer FBT shown in FIG. 7, for example, two U-shaped cores CR1 and CR2 made of a ferrite material are used.
The U-U-shaped core CR is formed by combining the respective magnetic legs so as to face each other. Then, gaps G1 and G2 are provided at portions where the magnetic leg ends of the U-shaped core CR1 and the magnetic leg ends of the U-shaped core CR2 face each other. Then, as shown in the figure, the low-voltage winding bobbin LB on which the primary winding N0 is wound is attached so as to penetrate one magnetic leg of the UU-shaped core CR. Then, the boost winding N is provided further outside the low-voltage winding bobbin LB.
The high-voltage bobbin HB on which HV (1 to 5) is wound is attached so as to penetrate. As a result, a structure in which the primary winding N0 and the boost windings NHV (1 to 5) are separately wound can be obtained.

Here, as the boost winding NHV wound around the high-voltage winding bobbin HB, for example, a plurality of boost windings NHV (1 to
It is necessary to wind each of 5) in an insulated state. For this reason, the winding method of the boost windings NVH is determined by each boost winding NVH (1 to
5) is a so-called interlayer winding in which an interlayer film F is interposed for each winding layer obtained by winding a predetermined number of times.
Then, after winding the boost windings NVH (1 to 5) as described above, the boost windings NVH (1 to 5) are applied to the circuit so that the mode shown in FIG. On the other hand, high voltage rectifier diodes DHV (1 to 5) are connected and attached. In practice, the structure shown in FIG. 7 is housed in a case, filled with a filler such as a high-molecular epoxy resin or the like, and then molded to ensure insulation.

In order to obtain an insulated state for each of the step-up windings NHV (1 to 5), in addition to the structure shown in FIG. 7, as shown in FIG. (Slit winding) can also be adopted. In FIG. 8, the same parts as those in FIG. In the case where the boost winding NHV is wound by split winding, as shown in FIG.
A partition plate DV is formed integrally with the inside of B1. As a result, a plurality of slits S, which are winding regions, are formed between adjacent partition plates DV. Then, the boosting windings NHV are wound around the respective slits S so as to obtain insulation between the boosting windings NHV. Then, depending on the structure of the flyback transformer FBT shown in FIG. 7 or FIG. By being wound by “coaxial winding”, a tightly coupled state can be obtained as a mutually coupled state. For example, in practice, the coupling coefficient k = 0.
A close coupling of about 98 is obtained.

The waveform diagram of FIG. 5 shows the operation of the main part in the power supply circuit having the configuration shown in FIG. The parallel resonance voltage V1 obtained at both ends of the primary side parallel resonance capacitor Cr is, as shown in FIG. 5A, a period during which a voltage resonance pulse is obtained during a period TOFF1 during which the main switching element Q1 is off and a period during which it is on. At TON1, a waveform having a level of 0 is obtained, which is a waveform corresponding to the switching timing of the main switching element Q1 of the primary-side voltage resonance type converter. AC input voltage V
In the case of AC = 100V system, the voltage doubler rectifier circuit (Di1, Di
When a DC input voltage is obtained in (2, Ci1, Ci2), in this case, the peak value of the voltage resonance pulse voltage V1 generated at both ends of the switching element Q1 is about 1200V.

At this time, as shown in FIG. 5D, the collector current IQ1 flowing through the switching element Q1 first has a negative clamp current flowing through the clamp diode DD at the start of the period TON1. , And a waveform flowing to the collector-emitter inverted to the positive level is obtained. FIG. 5 shows the winding current I1 obtained in the primary winding N1 by the switching operation on the primary side.
As shown in (b), a substantially sinusoidal waveform that is inverted positive / negative for each switching cycle is obtained.

A sinusoidal winding current I3 flows through the primary winding N0 of the flyback transformer FBT as shown in FIG. 5 (c). This winding current I3 is
A waveform substantially corresponding to the winding current I1 shown in FIG. In response to this, the rectified current Io flowing through the rectifier circuit on the secondary side of the flyback transformer FBT is shown in FIG.
Flows according to the waveform shown in FIG. That is, a waveform that flows in a sinusoidal manner depending on the positive polarity direction is obtained within the period TOFF1.

The secondary parallel resonance voltage V2 obtained at both ends of the secondary parallel resonance capacitor C2 provided on the secondary side of the insulating converter transformer PIT is shown in FIG.
As shown in (e), during the period DON during which the rectifier diode DO1 is turned on, the secondary-side DC output voltage EO1
During the OFF period DOFF, the waveform has a sine wave peak in the positive direction of the negative electrode. Then, the rectifier diode DO1 is supplied from the secondary winding N2.
As shown in FIG. 5 (f), the winding current I2 which flows into the PDP is maintained at a substantially predetermined constant level in the positive direction during the period DON, and reverses from positive to negative during the period DOFF. Thus, a waveform having a peak in the positive direction of the negative electrode is obtained.

The operation of the active clamp circuit 20 provided on the secondary side of the insulating converter transformer PIT is shown as a clamp current IQ2 in FIG. That is, in the first half of the period TON2 in which the switching circuit (Q2 // DD2) is turned on by conducting, the current flows through the path of the clamp diode DD2 → the clamp capacitor CCL → the secondary winding N2, and thus the clamp current IQ2 As a sawtooth wave due to negative polarity, and in the latter half period,
The flow of the current is reversed to become positive, and the secondary winding N
It is made to flow in the path of 2 → Q2 drain → Q2 source. In a period TOFF2 during which the switching circuit (Q2 // DD2) is turned off, the waveform has a waveform that maintains the 0 level. In addition, depending on the control by the first control circuit 1A, this period TON2 is varied by controlling the conduction angle of the auxiliary switching element Q2.

[0037]

By the way, the power supply circuit having the configuration shown in FIG. 4 does not need to be combined with a plurality of switching converters for a cathode ray tube display unlike the conventionally known configuration. From that
It can be said that it is advantageous for miniaturization. If further miniaturization is achieved, it is possible to sufficiently respond to recent demands for miniaturization of electronic devices, and it is possible to provide more useful power supply circuits.

However, it is difficult to effectively reduce the size of the power supply circuit shown in FIG. The most problematic factors that hinder miniaturization include the following. In the power supply circuit shown in FIG. 4, an insulating converter transformer PIT, a quadrature control transformer PRT, and a flyback transformer FBT are provided as transformers. It has a relatively large size.
That is, in the circuit shown in FIG. 4, since two sets of large transformers are provided, a considerable mounting area on the board is required.

Specifically, the insulation converter transformer PI
The core cross-sectional area of T is 1.23 square centimeters and the core cross-sectional area of the flyback transformer FBT is 2.01 square centimeters, whereas the load power of the secondary side DC output voltage EO1 on the isolation converter transformer PIT side is 1
The load power is about 50 W, and the load power of the DC high voltage EHV on the secondary side of the flyback transformer FBT is about 70 W. That is, although the load power of the DC high voltage EHV is about 1/2 of the secondary side DC output voltage EO1, when comparing the core cross-sectional area, the flyback transformer FBT is 63% larger than the insulating converter transformer PIT. Has become something. This means that in terms of the utilization factor of the flyback transformer FBT core, the isolated converter transformer PIT
Is significantly inferior to the utilization rate, indicating that the size is not efficient. Therefore, in order to promote the miniaturization of the power supply circuit based on the configuration shown in FIG. 4, it is necessary to solve the problem of the transformer.

The switching element Q is provided from the insulating converter transformer PIT and the flyback transformer FBT.
A leakage magnetic flux having a frequency corresponding to the switching frequency of No. 1 is generated. For example, the frequency of the leakage magnetic flux ranges from 80 KHz to 95 KHz by controlling the switching frequency fs.

On the other hand, the frequency fh of the horizontal synchronization signal of the television receiver is fh = 15.75K in the NTSC system.
Hz, fh = 33.75KHz in the high-vision system,
In the double speed system of the NTSC system, fh = 31.5 KHz, 3
The double speed system differs depending on various television broadcasting systems such as fh = 47.25 KHz. With respect to a horizontal deflection circuit operating in synchronization with the horizontal synchronization signal, a deflection yoke is provided at the neck of a cathode ray tube (CRT), and a horizontal linearity correction coil, a dynamic focus transformer, and the like are provided on a printed circuit board. Many reactors and inductors are mounted. Then, when the leakage magnetic flux of the above-mentioned insulating converter transformer PIT and flyback transformer FBT is coupled to these components of the horizontal deflection circuit,
A power beat with oblique stripes is generated on the surface of the CRT due to interference between the horizontal synchronization frequency fh and the switching frequency fs.

For this purpose, the insulation converter transformer PI
Each transformer (PIT, FBT) must be provided with magnetic shielding measures for the T and flyback transformer FBT, and the material cost of the copper plate is required for the shielding process and the necessity of the mounting / soldering process is required. However, there is a problem in that the manufacturing process becomes complicated and cost increases.

[0043]

In view of the above-mentioned problems, the present invention is configured as a switching power supply circuit as follows. That is, a switching means formed with a main switching element for intermittently outputting a DC input voltage, a U-U-shaped core in which two sets of U-shaped cores are joined via a gap, and a U-shaped core. A high voltage primary winding wound around one of the magnetic legs of the U-shaped magnetic core and to which the switching output of the switching means is transmitted; and a high voltage primary winding wound coaxially with the high voltage primary winding. A secondary step-up winding which is provided with a required degree of coupling tightly coupled to the winding; a switching output of the switching means which is wound around the other magnetic leg of the UU-shaped core; And the low voltage primary winding is wound around the same magnetic leg as the low voltage primary winding so as to obtain a required degree of coupling that is loosely coupled to the low voltage primary winding. And a low voltage secondary winding, and A high-voltage generating transformer in which a winding winding on which the low-voltage primary winding is wound is connected in series with the high-voltage primary winding, and the low-voltage primary winding in order to make the operation of the switching means a voltage resonance type. A primary-side parallel resonance circuit having a primary-side parallel resonance capacitor connected in parallel with the wire, a primary-side series resonance circuit including a primary-side series resonance capacitor connected in series to the high-voltage primary winding, and a low-voltage secondary A secondary parallel resonance circuit formed by connecting a secondary parallel resonance capacitor to the winding in parallel, and the low-voltage secondary winding formed including the secondary parallel resonance circuit and including the secondary parallel resonance circuit DC low voltage generating means configured to obtain a DC low voltage by performing a rectification operation on the obtained alternating voltage, and DC high voltage by performing a rectification operation on the high voltage obtained in the secondary side boost winding. DC high-voltage generating means for obtaining a voltage, and a series connection circuit of a secondary-side clamp capacitor and a secondary-side auxiliary switching element, and this series connection circuit is parallel to the secondary-side parallel resonance circuit. The secondary-side active clamp means formed so as to be connected to, and by controlling the conduction angle of the secondary-side auxiliary switching element according to the level of the DC low voltage, to perform constant voltage control The first
, A series connection circuit of a primary side clamp capacitor and a primary side auxiliary switching element, and the series connection circuit is formed so as to be connected in parallel to the low voltage primary winding. A primary-side active clamp unit and a second constant-voltage control unit configured to perform a constant-voltage control by performing a conduction angle control of the primary-side auxiliary switching element according to a level of the DC high voltage. Be prepared. Furthermore, the main switching element may be provided with a synchronizing means for executing a switching operation in synchronization with the horizontal synchronizing signal frequency based on a signal synchronizing with the horizontal synchronizing signal used in the cathode ray tube display device.

According to the above configuration, the high-voltage primary winding and the boosting winding are wound around the high-voltage generating transformer in a tightly coupled state, and the low-voltage primary winding and the low-voltage secondary winding are wound. The wires are wound so that they are loosely coupled. Therefore, the overall configuration of the switching power supply circuit includes a primary-side voltage resonance type switching converter, a high-voltage generating transformer that transmits the switching output to the secondary side, and a DC high-voltage formed on the secondary side of the high-voltage generating transformer. A rectifying circuit system as a generating means and a rectifying circuit system as a DC low voltage generating means are provided. Here, in the rectification circuit system of the DC low-voltage generating means, a secondary-side parallel resonance circuit is formed, so that a composite resonance switching converter is formed as the whole power supply circuit. With such a circuit configuration, in order to obtain a high DC voltage and a low DC voltage with a set of switching power supply circuits,
Power transmission from the primary side to the secondary side is performed by a set of high-voltage generating transformers. In other words, in other words, there is no need to provide two sets of large transformers, a high-voltage generating transformer and an insulating converter transformer, and the insulating converter transformer is omitted.

In the high-voltage generating transformer, a hoist winding wound around the low-voltage primary winding is connected in series with the high-voltage primary winding, and the high-voltage primary winding is connected in series with a primary-side series resonance capacitor. Connected. Then, the hoisting winding, the high-voltage primary winding, and the primary-side series resonance capacitor connected in series are connected in parallel with the low-voltage primary winding. For this reason, a voltage waveform similar to the voltage resonance pulse voltage generated at both ends of the main switching element is induced in the winding, and the low voltage primary winding and the voltage resonance pulse voltage generated in the winding are converted to the high voltage primary winding. Is applied to This has the effect of increasing the DC high voltage from the voltage value originally intended to be obtained. Conversely, even if the peak of the voltage resonance pulse voltage generated at both ends of the main switching element is reduced, the DC voltage which should be originally obtained is obtained. It means that a high voltage can be obtained.

Further, the synchronizing means can synchronize the operation of the composite resonance type switching converter with the horizontal synchronizing frequency, thereby preventing the power beat from being generated due to the interference between the leakage magnetic flux of the transformer and the horizontal synchronizing frequency.

[0047]

DESCRIPTION OF THE PREFERRED EMBODIMENTS Hereinafter, a switching power supply circuit according to an embodiment of the present invention will be described. The switching power supply circuit according to the present embodiment described below adopts a basic configuration as a complex resonance type switching converter including a voltage resonance type converter on the primary side and a parallel resonance circuit on the secondary side. The power supply circuit shown in this figure is mounted on a television receiver, a monitor display device, a projector device, or the like having a CRT (cathode ray tube) as a display device.

FIG. 1 shows a configuration example of a switching power supply circuit according to a first embodiment of the present invention. In the power supply circuit shown in FIG.
By connecting [rectifier diodes Di1, Di2, smoothing capacitors Ci1, Ci2] to C in the connection configuration shown in the figure, a voltage doubler rectifier circuit is formed. The voltage doubler rectifier circuit includes a series-connected smoothing capacitor Ci1−
A rectified smoothed voltage Ei (DC input voltage) corresponding to twice the AC input voltage VAC is generated at both ends of Ci2. In this embodiment, the reason for obtaining the rectified smoothed voltage Ei corresponding to twice the AC input voltage VAC in this manner is as follows.
As will be described later, it is necessary to obtain a required level of the DC high voltage EHV by the high voltage generating circuit 40. To improve the efficiency, the peak level of the primary side parallel resonance voltage V1 needs to be about 1000 V. A correspondingly high input voltage level to the resonant converter is required.

The switching converter which inputs and outputs the DC input voltage Ei intermittently is provided with a single main switching element Q1 and a so-called single-ended type self-excited voltage resonant converter which performs a switching operation. . In this case, a high withstand voltage bipolar transistor (BJ) is connected to the main switching element Q1.
T: junction type transistor).

The collector of the main switching element Q1 is connected to the positive terminal of the smoothing capacitor Ci, and the emitter is the low voltage primary winding N of the flyback transformer FBT.
1 connected. The base of the main switching element Q1 is connected to a series resonance circuit for self-excited oscillation driving, which includes a series connection circuit of a drive winding NB, a resonance capacitor CB, an inductor LB, and a base current limiting resistor RB. In this case, an external trigger pulse as the horizontal synchronizing signal frequency fH is input to the base of the main switching element Q1 via the insulating capacitor Ct. This external trigger pulse is extracted from a horizontal deflection circuit system, which is a load of a DC high voltage EHV described later.

A clamp diode DD is connected in parallel between the base and the emitter of the main switching element Q1 to form a path for a clamp current flowing when the main switching element Q1 is turned off.

Here, the drive winding NB forming the self-excited oscillation drive circuit of the main switching element Q1 is formed so as to wind up the winding end end of the low-voltage primary winding N1 of the flyback transformer FBT. I have. As a result, an alternating voltage is generated in the drive winding NB as a drive voltage transmitted from the low voltage primary winding N1. This drive voltage is output as a drive current from the series resonance circuit (NB, CB, LB) forming the self-excited oscillation drive circuit to the base of the main switching element Q1 via the base current limiting resistor RB. That is, a drive signal having the resonance frequency of the series resonance circuit is supplied to the main switching element Q1.

Further, in the present embodiment, as described above, an external trigger pulse having the horizontal synchronizing signal frequency fH is input to the base of the main switching element Q1. The horizontal resonance signal frequency fH is used as the series resonance circuit (NB, CB, LB).
A predetermined resonance frequency, which is slightly lower than the resonance frequency, is set, and the main switching element Q1 is driven by a driving signal based on the resonance frequency. At this time, an external trigger pulse is input. As a result, the main switching element Q1
Performs a switching operation in synchronization with the timing of the external trigger pulse. That is,
The switching operation is performed in synchronization with the horizontal deflection frequency of the beam current driving the CRT for display. Specifically, for example, in the case of supporting the HDTV of the NTSC system, the switching timing is synchronized with the horizontal synchronizing signal frequency fH = 31.5 KHz. In the case of supporting the digital television broadcasting, the horizontal synchronizing signal frequency fH =
The switching timing is synchronized with 33.75 KHz or fH = 45 KHz. At the time of startup,
The main switching element Q1 starts a switching operation by a starting current flowing from the rectified smoothed voltage Ei to the base via the starting resistor Rs.

A primary side parallel resonance capacitor C is provided between the emitter of the main switching element Q1 and the primary side ground.
r is connected. The primary side parallel resonance capacitor Cr is in parallel with the low voltage primary winding N1 of the flyback transformer FBT. The primary side parallel resonance capacitor Cr forms a primary side parallel resonance circuit of the voltage resonance type converter by its own capacitance and the leakage inductance L1 of the low voltage primary winding N1. Although not described in detail here, when the main switching element Q1 is turned off, the voltage V1 across the primary side parallel resonance capacitor Cr due to the operation of the primary side parallel resonance circuit is actually a sinusoidal waveform. A voltage resonance operation is obtained in the form of a pulse waveform.

The primary side active clamp circuit 10 is provided on the primary side of the power supply circuit shown in FIG. The primary side active clamp circuit 10 includes an auxiliary switching element Q2, a clamp capacitor CCL2, and a clamp diode DD2. In this case, the auxiliary switching element Q
For 2, a MOS-FET is selected. Further, a body diode incorporated in the auxiliary switching element Q2 which is a MOS-FET can be used as the clamp diode DD2.

The drain of the auxiliary switching element Q2 is connected via a clamp capacitor CCL2 to the connection point between the emitter of the main switching element Q1 and the end of the low voltage primary winding N1. The source of the auxiliary switching element Q2 is the primary side ground (the low voltage primary winding N1).
(The winding start end side). The clamp diode DD2 has an anode connected to the source of the auxiliary switching element Q2 and a cathode connected to the drain of the auxiliary switching element Q2. As described above, the primary side active clamp circuit 10 of the present embodiment is configured by connecting the clamp capacitor CCL2 in series to the switching circuit including the auxiliary switching element Q2 and the clamp diode DD2.
The circuit thus formed is connected in parallel to the low voltage primary winding N1 of the flyback transformer FBT.

As a drive circuit system for the auxiliary switching element Q2, an LCR series resonance circuit formed by a series connection circuit of a resistor Rg, a capacitor Cg and a drive winding Ng is connected to the gate of the auxiliary switching element Q2. Is done. As the resonance frequency of the LCR series resonance circuit (Cg-Rg-Ng), a series resonance circuit (RB-CB-N) forming a self-excited oscillation drive circuit of the main switching element Q1 is used.
It is set to be equivalent to B). That is, it is set so as to be substantially equal to the switching frequency of the main switching element Q1.

The drive winding Ng is formed so as to wind up the winding start end of the low-voltage primary winding N1 in the flyback transformer FBT. As a result, an alternating voltage excited by the alternating voltage which is the switching output of the main switching element Q1 obtained in the low voltage primary winding N1 is generated in the drive winding Ng. Further, in this case, due to the relationship of the winding direction, an alternating voltage of the opposite polarity is obtained between the low voltage primary winding N1 and the driving winding Ng. The alternating voltage obtained in this manner causes the LCR series resonance circuit to perform a resonance operation, and applies its output to the gate of the auxiliary switching element Q2.
The auxiliary switching element Q2 driven in this manner performs a switching operation at a switching cycle similar to that of the main switching element Q1 and at substantially alternate on / off timings.

The primary side active clamp circuit 10 is connected to both ends of the parallel circuit of the main switching element Q1 // parallel resonance capacitor Cr as described later by the switching operation of the switching circuit (Q2 // DD2). It operates so as to suppress the peak level of the generated resonance voltage V1. A control voltage is applied from a second control circuit 1B, which will be described later, to the gate of the auxiliary switching element Q2. According to the level of the control voltage, PWM control (conduction angle control) for the ON period of the auxiliary switching element Q2 is performed. ) Is performed. As a result, the DC high voltage EHV described later is stabilized.

The flyback transformer FBT originally transmits the switching output of the main switching element Q1 from the primary side to the secondary side, and generates a high voltage alternating voltage for obtaining a high voltage DC voltage for the anode voltage on the secondary side. Prepared for. However, in the present embodiment, the secondary side of the flyback transformer FBT is also configured to generate, for example, a low-voltage alternating voltage for obtaining a low-voltage secondary-side DC output voltage for various circuits.

For this reason, in the flyback transformer FBT of this embodiment, as the primary winding wound on the primary side, a low-voltage primary winding N1 is provided in addition to the high-voltage primary winding N0. Further, as described above, the winding start side of the low-voltage primary winding N1 is wound up to provide the drive winding Ng, and the winding end side of the low-voltage primary winding N1 is further wound up to provide the drive winding NB. The winding end side of the low-voltage primary winding N1 is further wound up following the driving winding NB to provide a winding winding N4.

The hoist winding N4 is connected in series to the winding start side of the high voltage primary winding N0. A primary-side series resonance capacitor C5 is connected in series to the winding end side of the high-voltage primary winding N0. The other end of the primary side series resonance capacitor C5 is connected to the primary side ground. This high voltage primary winding N0
And the capacitance of the primary-side series resonance capacitor C5 forms a primary-side series resonance circuit. This primary side series resonance circuit is provided in parallel with the primary side parallel resonance circuit including the low voltage primary winding N1 and the primary side parallel resonance capacitor Cr. As a secondary winding of the flyback transformer FBT, a low-voltage secondary winding N2 is provided in addition to the boost winding NHV (1 to 5).

Since the low voltage primary winding N1 is inserted between the emitter of the main switching element Q1 and the primary side ground, the switching output of the main switching element Q1 is transmitted. The high voltage primary winding N0 is also inserted between the emitter of the main switching element Q1 and the primary side ground (via the series resonance capacitor C5) via the winding winding N4, so that the main switching is performed. The switching output of element Q1 is transmitted.

As a circuit configuration on the secondary side of the flyback transformer FBT, the configuration on the high voltage generating circuit 40 side provided with the boost windings NVH (1 to 5) is as follows.
In the figure, a high-voltage generation circuit 40 surrounded by a dashed line is
It is composed of a flyback transformer FBT and a high-voltage rectifier circuit, and utilizes an alternating voltage (switching output) obtained in a high-voltage primary winding N0 of the flyback transformer FBT, for example, to adjust a DC voltage corresponding to an anode voltage level of a CRT. Generate voltage. Therefore, on the secondary side of the flyback transformer FBT, four to five sets of boost windings NVH
However, it is divided and wound by so-called slit winding or interlayer winding as described later. In this case, the low-voltage primary winding N0 and the boost winding NHV are wound so as to be tightly coupled. In this case, the primary winding N
The coupling coefficient k between 0 and the step-up winding NVH is k ≧ 0.95
It has been. On the secondary side of the flyback transformer FBT, the winding voltage V3 generated in the primary winding N0
A boosted voltage is obtained in accordance with the turn ratio between the HV and the primary winding N0 (NHV / N0).

In the case of the power supply circuit shown in this figure, five sets of boost windings NHV are provided on the secondary side of the flyback transformer FBT.
1, NHV2, NHV3, NHV4, and NHV5 are wound independently of each other, and high-voltage rectifier diodes DHV1, DH
The anodes of V2, DHV3, DHV4 and DHV5 are connected. Then, the cathode of the high voltage rectifier diode DHV1 is connected to the positive terminal of the smoothing capacitor COHV, and the respective cathodes of the remaining high voltage rectifier diodes DHV2 to DHV5 are connected to the winding start ends of the step-up windings NVH1 to NHV4, respectively.

That is, on the secondary side of the flyback transformer FBT, [boost winding NHV1, high voltage rectifier diode DHV
1], [boost winding NHV2, high voltage rectifier diode DHV2],
[Step-up winding NHV3, high-voltage rectifier diode DHV3], [Step-up winding NHV4, high-voltage rectifier diode DHV4], [Step-up winding N
HV5, high-voltage rectifier diode DHV5], that is, a so-called multisingle-type half-wave rectifier circuit in which five sets of half-wave rectifier circuits are connected in series.

Therefore, on the secondary side of the flyback transformer FBT, five sets of half-wave rectifier circuits
~ NHV5 rectifies the current induced and smoothes capacitor CO
An operation of charging the HV is performed, and a high DC voltage E of a level corresponding to about five times the induced voltage induced in each of the boost windings NVH1 to NHV5 is provided across the smoothing capacitor COHV.
HV will be obtained. This DC high voltage EHV is CRT
It is used as the anode voltage of The signal is also branched and input to the second control circuit 1B for constant voltage control.

The configuration of the low voltage secondary winding N2 of the flyback transformer FBT is as follows. In this case, the winding end of the low voltage secondary winding N2 is connected to the anode of the rectifier diode DO1, and the winding start end is connected to the secondary side ground. Then, a secondary-side DC output voltage EO1 is obtained by a half-wave rectifying / smoothing circuit including the rectifying diode DO1 and the smoothing capacitor CO1.
The secondary side DC output voltage EO1 is set to, for example, 135 V and used as a horizontal deflection circuit system. Further, the detection voltage is branched and supplied to the first control circuit 1A.

In this case, a tap is provided for the secondary winding N2 as shown in the figure, and a half-wave rectifier composed of a rectifier diode DO2 and a smoothing capacitor CO2 is provided for the tap output as shown in the figure. By connecting the circuit, the voltage is made lower than the secondary side DC output voltage EO1.
For example, a secondary DC output voltage EO2 of 15 V is generated. This secondary side DC output voltage EO2 is used, for example, in a vertical deflection circuit system.

In practice, a low-voltage secondary-side DC output voltage of a required level to be supplied to other various circuit systems may be generated. For example, a video output circuit system ( 200 V), a secondary side DC output voltage for a CRT heater circuit system (7.5 V), an audio output circuit system (24 V), and the like.

A secondary parallel resonance capacitor C2 is connected in parallel to the low voltage secondary winding N2. In this case, a leakage inductance L2 of the low voltage secondary winding N2 and a capacitance of the secondary parallel resonance capacitor C2 form a secondary parallel resonance circuit. Thus, the alternating voltage induced on the secondary side becomes a resonance voltage, and a voltage resonance operation is obtained on the secondary side.

That is, the power supply circuit of this embodiment also
A composite resonance type switching in which the primary side of the flyback transformer FBT is provided with a parallel resonance circuit for performing a voltage resonance type switching operation, and the secondary side is provided with a parallel resonance circuit for obtaining a voltage resonance type operation. It forms a converter.

Here, as the flyback transformer FBT of the present embodiment, in order to obtain a flyback operation for obtaining a high DC voltage, the high-voltage primary winding N0 and the boost windings NVH1 to NHV5 are tightly coupled. On the other hand, in order to obtain the operation as the above-described composite resonance type switching converter, the low voltage primary winding N1 and the low voltage secondary winding N2 need to be loosely coupled. . Therefore,
As the flyback transformer FBT of the present embodiment,
A structure in which the primary winding N0 for high voltage and the boost winding NHV (1 to 5) are tightly coupled, and the primary winding N1 for low voltage and the secondary winding N2 for low voltage are loosely coupled. It is assumed to have. The structure of the flyback transformer FBT will be described later.

A secondary active clamp circuit 11 is provided for the secondary parallel resonance circuit. The secondary side active clamp circuit 11 includes an auxiliary switching element Q3, a clamp capacitor CCL3, and a clamp diode DD3. A MOS-FET is selected for the auxiliary switching element Q3, and the clamp diode D
A body diode is used for D3.

The drive circuit system for driving the auxiliary switching element Q3 includes a capacitor Cg3− It is formed by connecting an LCR series resonance circuit to which a resistor Rg3 is connected.

Also in this case, since the alternating voltage of the opposite polarity is obtained between the low voltage secondary winding N2 and the driving winding Ng3 due to the winding direction, the auxiliary switching element Q3 and the secondary rectifier are connected. The diode performs a switching operation at substantially alternating on / off timings. The switching frequency of the secondary-side auxiliary switching element Q3 and the secondary-side rectifier diode is determined by the relationship that the switching output of the primary-side voltage resonance type converter is transmitted to the secondary side via the flyback transformer FBT. , Corresponding to the switching frequency of the main switching element Q1.

The drain of the auxiliary switching element Q3 is connected to the winding end of the secondary winding N2 via the clamp capacitor CCL3. The source of the auxiliary switching element Q3 is grounded with respect to the secondary side ground. The clamp diode DD3 has an anode connected to the source of the auxiliary switching element Q3 and a cathode connected to the drain of the auxiliary switching element Q3, so that a path of a clamp current flowing during a period in which the auxiliary switching element Q3 is turned off. Is formed. Thus, as the secondary side active clamp circuit 11 of the present embodiment,
A switching circuit comprising the auxiliary switching element Q3 and the clamp diode DD3 is connected to a clamp capacitor CCL in series. The circuit thus formed is further connected in parallel to a secondary parallel resonance circuit comprising a secondary winding N2 // secondary parallel resonance capacitor C2.

The stabilizing operation in the power supply circuit according to the present embodiment is as follows. The first control circuit 1A outputs a DC control voltage of a level that is varied according to a change in the secondary-side DC output voltage level EO1. Depending on the control voltage, the gate threshold voltage (bias) of the auxiliary switching element Q3 in the secondary-side active clamp circuit 11 is varied. In this case, the ON period of the auxiliary switching element Q3 is reduced. Will be variable. That is, PWM control for the conduction angle is performed. In the case of the circuit configuration shown in FIG. 1, when the switching circuit (Q3 // DD3) conducts during the ON period and a current flows through the clamp capacitor CCL3, the current should flow into the secondary parallel resonance capacitor C2 and be charged. A current flows through the clamp capacitor CCL3. By this operation, the peak level of the secondary parallel resonance voltage V2 obtained at both ends of the secondary parallel resonance capacitor C2 is suppressed and clamped. Therefore, the auxiliary switching element Q
If the conduction angle of the capacitor 3 is variably controlled and the amount of current flowing through the clamp capacitor CCL3 is varied, the amount of charging current in the secondary parallel resonance capacitor C2 is varied, and the clamp of the secondary parallel resonance voltage V2 is performed. The level also changes. By changing the level of the secondary side parallel resonance voltage V2 in this manner, the level of the rectified current flowing into the smoothing capacitor CO1 changes, and as a result, the level of the secondary side DC output voltage EO1 is changed. Can be obtained. By such an operation, the low-voltage secondary-side DC output voltage is stabilized.

Further, a series connection circuit of voltage dividing resistors R1-R2 is provided in parallel with respect to the smoothing capacitor COHV from which the DC high voltage EHV is obtained. The voltage dividing resistor R1−
The voltage dividing point of R2 is connected to the second control circuit 1B. That is, in the present embodiment, the second control circuit 1B
, A voltage level obtained by dividing the DC high voltage EHV by the voltage dividing resistors R1-R2 is input as the detection voltage. The second control circuit 1B includes a DC high voltage EHV
And outputs a DC control voltage of a level that is varied according to the level change of. This control voltage changes the bias (gate threshold voltage) to be applied to the gate of the auxiliary switching element Q2 in the primary-side active clamp circuit 10. By varying the bias of the auxiliary switching element Q2 in this manner, the ON period of the auxiliary switching element Q2 within one switching cycle is varied. That is, PWM control for the conduction angle is performed. As described above, depending on the switching operation of the auxiliary switching element Q2, the resonance voltage V1 generated at both ends of the primary-side parallel resonance capacitor Cr is generated.
Can be obtained by clamping the switching element.
By performing the M control, the voltage level of the resonance voltage V1 is variably controlled according to the fluctuation of the DC high voltage level. Here, the switching output of the main switching element Q1, the level of which is varied according to the level change of the resonance voltage V1, is supplied to the high voltage primary winding N0. For this reason, the current level of the winding current flowing through the primary winding N0 of the flyback transformer FBT also changes,
Along with this, the level of the winding voltage generated at both ends of the primary winding N0 is also varied. As a result, the level of the induced voltage induced on the secondary side of the flyback transformer FBT is varied, and the DC high voltage EHV output from the high voltage generation circuit 40 is stabilized.

FIG. 3 is a sectional view showing a structural example of a flyback transformer FBT provided in the power supply circuit shown in FIG. In the flyback transformer FBT shown in this figure, for example, two U-shaped cores CR made of a ferrite material are used.
The U-U-shaped core CR is formed by combining the magnetic legs 1 and CR2 so as to face each other. Then, gaps G1 and G2 are provided at portions where the magnetic leg ends of the U-shaped core CR1 and the magnetic leg ends of the U-shaped core CR2 face each other. Then, as shown in the figure, the low-voltage winding bobbin LB on which the high-voltage primary winding N0 is wound is attached so as to penetrate one of the magnetic legs of the U-U-shaped core CR. Then, the high-voltage winding bobbin HB on which the boost windings NHV (1 to 5) are wound is attached to the outside of the low-voltage winding bobbin LB so as to pass therethrough. As a result, a structure in which the primary winding N0 for high voltage and the boost winding NHV (1 to 5) are divided and wound is obtained. And depending on this structure, the high voltage primary winding N0 and the secondary side boosting winding NHV
Regarding (1-5), the same magnetic leg is wound by so-called “coaxial winding”. For this reason, a tightly coupled state is obtained as a mutually coupled state. For example, in practice, the coupling coefficient k = 0.95
The above tightly coupled state can be obtained.

Here, as the boost winding NHV wound around the high-voltage winding bobbin HB, for example, a plurality of boost windings NVH (1 to
It is necessary to wind each of 5) in an insulated state. For this reason, the winding method of the boost windings NVH is determined by each boost winding NVH (1 to
5) is a so-called interlayer winding in which an interlayer film F is interposed for each winding layer obtained by winding a predetermined number of times.
Then, after winding the step-up windings NVH (1 to 5) as described above, each of the step-up windings NVH (1 to 5) is provided with a circuit so that the mode shown in FIG. 1 is obtained. On the other hand, high voltage rectifier diodes DHV (1 to 5) are connected and attached.

The step-up winding NVH may have a split winding (slit winding) structure as shown in FIG.

Then, the primary winding N0 and the boost winding NHV (1 to
5) For the other magnetic leg not wound, another
The two low-voltage winding bobbins LB-1 are attached so as to penetrate the magnetic legs. In this case, the low-voltage winding bobbin LB-1 is a so-called divided bobbin in which a winding portion for two windings is divided by providing one partition as illustrated. . Then, with respect to the low-voltage winding bobbin LB-1, the low-voltage primary winding N1 and the hoisting winding N4 (and the driving windings Ng and NB) and the low-voltage secondary winding N2 (and the driving winding Ng3). Are divided and wound around different winding portions to ensure insulation from each other. With such a winding structure, a loose coupling state with a coupling coefficient k of about 0.71 can be obtained for the low voltage primary winding N1 and the low voltage secondary winding N2. Thus, an operation as a complex resonance type switching converter can be obtained. Incidentally, the degree of coupling of the low-voltage primary winding N1 or the low-voltage secondary winding N2 to the high-voltage primary winding N0 is set to a coupling coefficient k of about 0.55. The number of turns of the winding N0 can be increased.

In such a configuration, as a power transmission correspondence in the flyback transformer FBT, the high-voltage primary winding N0 and the boost winding NHV correspond to each other, and the low-voltage primary winding N1 and the low-voltage secondary winding N1. N2 will correspond. That is, as described above, the alternating voltage is excited in the boost winding NHV on the secondary side by the primary-side switching output obtained from the high-voltage primary winding N0. Further, depending on the switching output obtained from the low voltage primary winding N1, an alternating voltage is excited in the low voltage secondary winding N2.

FIG. 2 shows operation waveforms of main parts of the power supply circuit having the configuration shown in FIG. First, in the operation of the primary side, as described above, the switching timing of the main switching element Q1 is such that one switching cycle of the period TON1 + the period TOFF1 is synchronized with the horizontal synchronization signal frequency fH. The primary side parallel resonance voltage V1 obtained at both ends of the primary side parallel resonance capacitor Cr is, as shown in FIG. 2A, the main switching element Q of the primary side voltage resonance type converter.
It has a waveform corresponding to the switching timing of 1 and the period TOFF during which the main switching element Q1 is off.
At 1, a voltage resonance pulse is obtained. In the period TON1 during which the main switching element Q1 is turned on, the level is offset because the low-voltage primary winding N1 is connected, so that the constant level is maintained by the negative polarity instead of the 0 level. Is done.

The collector current IQ1 flowing through the switching element Q1 is, as shown in FIG.
In the period TON1, at the start of the period, a clamp current flows in the direction of negative polarity via the clamp diode DD → Q1 base → Q1 collector, and thereafter, the waveform is inverted to a positive level and flows to the collector-emitter. In the period TOFF1, it is at the 0 level. As shown in FIG. 2 (e), the winding current I3 flowing through the high-voltage primary winding N0 due to the switching operation on the primary side is substantially saw-tooth inverted to positive or negative in each switching cycle. An alternating waveform in the shape of is obtained. The winding current I1 flowing through the low-voltage primary winding N1, which is connected in parallel with the high-voltage primary winding N0, is shown in FIG. 2D, and is inverted from positive to negative during the period TOFF1. And period TON
In 1, a smooth waveform in which a predetermined level due to the positive polarity is maintained is obtained.

The primary side active clamp circuit 10
2 is shown as the clamp current IQ2 in FIG. 2 (c).
Flows through a sawtooth wave that reverses from negative polarity to positive polarity, and becomes 0 level in the period TOFF2. Here, the period TON2 is a period during which the switching circuit including the auxiliary switching element Q2 and the clamp diode DD2 conducts. In the first half of the period, the clamp diode DD2 →
Flows to the primary winding N0 through the clamp capacitor CCL2,
In the latter half period, the current flows from the primary winding N0 through the path of the clamp capacitor CCL2 → Q2 drain → Q2 source. In this case, the period TON2 is set to be within the period TOFF1. On the other hand, the period TOFF2 is a period in which the switching circuit is turned off. Here, as can be seen by comparing the [period TON1, TOFF1], which is the ON / OFF timing of the main switching element Q1, with the [period TON2, TOFF2], which is the ON / OFF timing of the auxiliary switching element Q2, Element Q1
The on / off operation is performed between the auxiliary switching element Q2 and the auxiliary switching element Q2 at substantially alternate timings. Then, as described above, the switching circuit (Q2 // DD2) in the primary-side active clamp circuit 10 performs an on / off operation, so that the voltage across the switching circuit (Q2 // DD2) is increased during the period TON2. At the 0 level, a waveform maintained at a constant positive polarity level in the period TOFF2 is obtained.

By the way, the period TOFF1 is a period during which the switching output current originally obtained on the primary side flows into the primary side parallel resonance capacitor Cr. As indicated by the current IQ2, the switching output current is caused to flow to the primary side active clamp circuit 10 in the period TON2. Accordingly, the switching output current flows through the primary side parallel resonance capacitor Cr only during the period other than the period TON2 in the period TOFF1, and the charging current amount is reduced accordingly. Thus, the primary side parallel resonance voltage V1 has a waveform whose peak level is clamped and suppressed in the period TOFF1 as shown in FIG. 2A.

The operation on the secondary side of the flyback transformer FBT is shown in FIGS. 2 (g) to 2 (i).
A secondary parallel resonance voltage V2, which is a voltage across the secondary parallel resonance capacitor C2 forming a secondary parallel resonance circuit with the low voltage secondary winding N2, is obtained as shown in FIG.
A waveform having a peak level in the negative polarity direction during a period DON in which the secondary side rectifier diode DO1 conducts and a rectified current flows and is clamped at the level of the secondary side DC output voltage EO1 during a period DOFF in which the secondary side rectifier diode DO1 conducts and a rectified current flows. Become.

The operation of the switching circuit (Q3 // DD3) of the secondary active clamp circuit 11 connected in parallel to the secondary parallel resonance circuit is performed as the clamp current IQ3 of FIG. Is shown. The switching circuit (Q3 // DD3) conducts during the period TON3 within the period DOFF, and turns off during the other period TOFF2. Further, the period TOFF2 is a period including the period DON. Therefore, the switching circuit (Q3 // DD3) of the secondary side active clamp circuit 11 and the rectifier diode DO1
Are turned on / off at substantially alternate timings. In the first half of the period TON3, a current flows through the path of the clamp diode DD3 → the clamp capacitor CCL3 → the low voltage secondary winding N2, so that a sawtooth wave with a negative polarity is obtained as the clamp current IQ3. , The current flow is inverted to have a positive polarity and flow through the low voltage secondary winding N2 → Q3 drain → Q3 source. Then, in a period TOFF3 in which the switching circuit (Q3 // DD3) is off,
This is a waveform in which the 0 level is maintained.

For example, by setting the clamp current IQ3 to flow in the period TON3 in this manner, the current to be charged in the secondary parallel resonance capacitor C2 is charged and discharged to the clamp capacitor CCL3. As a result, the alternating voltage V2 across the secondary side parallel resonance capacitor C2 shown in FIG.
The peak level of the negative polarity is clamped. As described above, when the level of the secondary side DC output voltage EO1 fluctuates due to the fluctuation of the load of the AC input voltage VAC or the secondary side DC output voltage EO1, the control circuit 1A controls the auxiliary switching element Q3. Conduction angle control is performed,
The period TON3 will be variable. Accordingly, the conduction angle of the rectifier diode DO1 is also controlled, and as a result, the level of the secondary DC output voltage EO1 is controlled to be stabilized.

The rectified current Io flowing in the high voltage generating circuit 40 on the secondary side of the flyback transformer FBT is:
It flows according to the waveform shown in FIG. That is, the period TOF
Within F1, a waveform that flows sinusoidally in the positive polarity direction is obtained.

Also, in this embodiment, as described above, the primary side of the low-voltage primary winding N
1 is wound up to provide a winding winding N4. The hoist winding N4 is connected in series to the high voltage primary winding N0, and the series resonance capacitor C5 is further connected in series. Moreover, between the emitter of the main switching element Q1 and the primary side ground, the low voltage primary winding N1 and the winding winding N
4. The series circuit of the high voltage primary winding N0 and the series resonance capacitor C5 is connected in parallel. For this reason, as shown in FIG. 2 (j), a voltage V4 having a waveform similar to the voltage resonance pulse voltage V1 is induced at both ends of the hoist winding N4 due to the turn ratio with respect to the low voltage primary winding N1. You. Then, the voltage resonance pulse voltage generated in the low voltage primary winding N1 and the hoisting winding N4 is applied to a primary side series resonance circuit composed of the high voltage primary winding N0 and the series resonance capacitor C5. The voltage V3 across the high-voltage primary winding N0 is as shown in FIG.

In the case of such a configuration having the hoist winding N4, as compared with, for example, the prior art circuit shown in FIG. 4, unless the various constants are changed, the DC high voltage EHV is increased. Become. Therefore, in order to obtain a desired voltage value (for example, 31.5 KV) as the value of the DC high voltage EHV, the number of turns of the five sets of the boost windings NHV (1 to 5) is reduced or the primary winding for high voltage is used. It is necessary to increase NO (that is, change the turns ratio). However, the boost winding NHV (1 to
5) If the peak value of the voltage resonance pulse voltage V1 generated at both ends of the main switching element Q1 is reduced without changing the number of turns of the primary winding N0 for high voltage,
The desired voltage value can be obtained as the value of HV.

That is, in the case of the present embodiment, the addition of the hoist winding N4 and the reduction of the peak value of the voltage resonance pulse voltage V1 generated at both ends of the main switching element Q1 with the addition thereof increase the primary winding for high voltage. The current flowing through N0 is reduced. As a result, the switching loss of the main switching element Q1 and the copper loss of the high voltage primary winding N0 are reduced, and the AC / DC power conversion efficiency is also improved.

Here, the specifications of the main components selected for the power supply circuit shown in FIG. 1 will be shown by comparison with the prior art power supply circuit shown in FIG. It is assumed that the load Po on the DC low voltage output side is 150 W to 100 W and the load PHV on the DC high voltage side is 31.5 KV × 2.15 mA = 68 W.

In the circuit shown in FIG. 4, the insulation converter transformer PIT uses an EE type core called EE-40 type, a gap length G = 1 mm, a primary winding N1 = 130T, and a secondary winding N2 = It was 100T.
The flyback transformer FBT has a gap length =
0.4mm × 2, primary winding N0 = 70T, boost winding NVH
(1-5) = 530T. Further, the primary side parallel resonance capacitor Cr was set to 2200 pF. In that case, the AC / DC power conversion efficiency was 90.1%.

On the other hand, in the power supply circuit according to the present embodiment shown in FIG.
Will be omitted. As for the flyback transformer FBT, the primary winding N0 for high voltage is used.
= 80T, low voltage primary winding N1 = 100T, low voltage secondary winding N2 = 70T, boost winding NHV = 530T, and winding winding N4 = 30T. Further, the primary side parallel resonance capacitor Cr = 5600 pF, the clamp capacitor CCL2 = 0.
15 μF, primary side series resonance capacitor C5 = 0.033
μF. In this case, the AC / DC power conversion efficiency was 91.1%. This also allowed the AC input power to be reduced by 3.2 W.

When the AC input voltage VAC is 100 V, the resonance voltage V1, the collector current IQ1, and the current I3 shown in FIGS.
The peak is reduced as compared with the case where the sample No. 4 is not provided. For example, in the case of the circuit of FIG. 4, the peak of the resonance voltage V1 is 1150 V.
The peak of the resonance voltage V1 could be reduced to 600V. Since the switching loss of the main switching element Q1 is reduced by reducing the peak value of the resonance voltage V1, the heat radiation structure can be downsized. In addition, a low withstand voltage element can be selected as the main switching element Q1.

Also, as can be seen from the above description,
In the power supply circuit of the present embodiment, the flyback transformer F
For the BT, not only the set of the primary winding N0 for high voltage and the boost winding NHV but also the primary winding N1 for low voltage and the secondary winding N2 for low voltage
Also winds up, one set of flyback transformer FBT
DC high voltage EHV and secondary side DC output voltage E at the secondary side of
O1 is going to get. That is, in the present embodiment, as a component, the insulating converter transformer PI
The core as T will be omitted. This allows
In the present embodiment, the mounting area of the printed circuit board is reduced accordingly, and it is possible to further reduce the size and weight. Moreover, in the present embodiment, the insulation converter transformer P
Since a core as an IT is not required, the size and weight can be greatly reduced.

Further, the low voltage primary winding N1 and the low voltage secondary winding N2 are wound around the flyback transformer FBT, so that the core cross-sectional area increases as compared with the case where the winding is wound around the insulating converter transformer PIT. So that
The number of turns of each of these windings will be reduced, which will also shorten the winding process time.

Further, according to the structure of the flyback transformer FBT shown in FIG. 3, the degree of coupling between the primary winding N0 and the low-voltage secondary winding N2 is a sufficiently low coupling coefficient k ≒ 0.55. It is possible to obtain a state of connection. For this reason, the leakage inductance of the low-voltage secondary winding N2 increases, and the capacitance of the secondary-side parallel resonance capacitor C2 that forms a secondary-side parallel circuit together with the low-voltage secondary winding N2 can be small. It will be. If the capacitance is small, the components of the selected capacitor can be small, and in this regard, the circuit can be reduced in size and weight.

In addition, according to the structure shown in FIG. 3, since the two gaps G1 and G2 are both wound, the magnetic flux leaking from the gap is shielded by these windings. Will be. Further, since the switching frequency fs is synchronized with, for example, the horizontal synchronization signal frequency fH, interference with the horizontal deflection circuit components due to the leakage magnetic flux radiated from the flyback transformer FBT does not occur. Therefore, it is not necessary to provide a magnetic shield for the flyback transformer FBT.

For example, in the above-described embodiment, a bipolar transistor is adopted as a main switching element, but a MOS-FE
It is also conceivable to employ other elements such as T and IGBT. Here, when a MOS-FET or an IGBT is adopted, for example, a switching drive may be performed by a separately-excited system by using an oscillation drive circuit using a general-purpose IC. Also, the secondary-side rectifier circuit formed including the secondary-side resonance circuit is not limited to the configurations shown in the drawings as the embodiments, and other circuit configurations may be adopted. Not something.

[0105]

As described above, in the power supply circuit according to the present invention, a high-voltage generating transformer (flyback transformer)
And the high-voltage primary winding and the boost winding are wound so as to be tightly coupled, and the low-voltage primary winding and the low-voltage secondary winding are wound so as to be loosely coupled. . In each of these configurations, an operation as a composite resonance type switching converter can be obtained by forming a secondary side parallel resonance circuit on the low voltage secondary winding side. . Then, on the secondary side of the flyback transformer, a DC high voltage is generated using the alternating voltage excited by the boost winding, and the alternating voltage excited by the low voltage secondary winding is used. A DC low voltage is generated. As a result of adopting such a configuration, in order to obtain two types of secondary-side DC output voltages, that is, a DC high voltage and a DC low voltage, the power supply circuit of the present invention only needs to provide a high-voltage generating transformer. The converter transformer will be deleted. By removing the insulating converter transformer, at least the core for the transformer will be deleted as a component, but since the insulating converter transformer is a relatively large transformer, the core is By being removed, the substrate size can be significantly reduced, and as a result, it is possible to effectively promote downsizing and cost reduction.

In the high-voltage generating transformer, a winding winding in which a low-voltage primary winding is wound is connected in series to a series circuit of a high-voltage primary winding and a series resonance capacitor. The wire, the high voltage primary winding and the series resonance capacitor are connected in parallel with the low voltage primary winding. As a result, AC / DC power conversion efficiency can be improved, and AC input power can be reduced. Further, since the peak value of the voltage resonance pulse voltage applied to the main switching element can be reduced, there is an advantage that switching loss is reduced and a heat sink can be downsized.

Further, since the converter operation by the switching operation is synchronized with the horizontal synchronizing signal used in the cathode ray tube display device, the leakage magnetic flux of the converter transformer or the high voltage generating transformer interferes with the horizontal synchronizing signal, so that the power supply beat. Will not occur. Therefore,
There is no need to provide a short ring or a shield plate made of a copper plate for shielding the leakage magnetic flux in the converter transformer or the high-voltage generating transformer. This lowers the manufacturing cost of the high-voltage generating transformer (flyback transformer),
There is an effect that simplification of manufacturing and efficiency improvement can be realized.
Furthermore, not providing a short ring or shield plate
There is also an advantage that the temperature rise of the high-voltage generating transformer is reduced.

[Brief description of the drawings]

FIG. 1 is a circuit diagram illustrating a configuration example of a switching power supply circuit according to an embodiment of the present invention.

FIG. 2 is a waveform chart showing an operation of a main part in the power supply circuit of the embodiment.

FIG. 3 is a cross-sectional view illustrating a structure example of a flyback transformer provided in the power supply circuit according to the embodiment;

FIG. 4 is a circuit diagram showing a configuration of a switching power supply circuit as a prior art.

5 is a waveform chart showing an operation of a main part in the power supply circuit shown in FIG.

FIG. 6 is a cross-sectional view illustrating a structural example of an insulating converter transformer.

FIG. 7 shows an example of the structure of a conventional flyback transformer.
It is sectional drawing which shows the case where a boost winding is wound between layers.

FIG. 8 shows an example of the structure of a conventional flyback transformer.
It is sectional drawing which shows the case where a boost winding is dividedly wound.

[Explanation of symbols]

1A first control circuit, 1B second control circuit, 10 primary side active clamp circuit, 11 secondary side active clamp circuit, FBT flyback transformer, 40 high voltage generation circuit, Q1 main switching element, Cr primary side parallel resonance capacitor, C2 Secondary parallel resonance capacitor, C5 Primary series resonance capacitor, Q2, Q3
Auxiliary switching element, N0 high voltage primary winding, N1 low voltage primary winding, N2 low voltage secondary winding, NHV1 ~ NHV5
Boost winding, DHV1 to DHV5 High voltage rectifier diode, COHV
Smoothing capacitor

 ──────────────────────────────────────────────────続 き Continuing on the front page F term (reference)

Claims (2)

[Claims] 1. A switching means formed with a main switching element for intermittently outputting a DC input voltage, and a U-U in which two sets of U-shaped cores are joined via a gap.
A high-voltage primary winding wound around one of the magnetic legs of the U-U-shaped core and to which the switching output of the switching means is transmitted, and wound coaxially with the high-voltage primary winding. Then, the secondary side step-up winding is formed so as to obtain a required degree of coupling that is tightly coupled to the high-voltage primary winding, and is wound around the other magnetic leg of the UU-shaped core. The low voltage primary winding to which the switching output of the switching means is transmitted, and the same magnetic leg as the low voltage primary winding so as to obtain a required degree of loose coupling with the low voltage primary winding. A low-voltage secondary winding wound around, and a high-voltage generating transformer in which a winding winding in which the low-voltage primary winding is wound up is connected in series with the high-voltage primary winding. In order to make the operation of the switching means a voltage resonance type, A primary-side parallel resonance circuit having a primary-side parallel resonance capacitor connected in parallel with the low-voltage primary winding, a primary-side series resonance circuit with a primary-side series resonance capacitor connected in series to the high-voltage primary winding, A secondary-side parallel resonance circuit formed by connecting a secondary-side parallel resonance capacitor in parallel to the low-voltage secondary winding; and a secondary-side parallel resonance circuit formed including the secondary-side parallel resonance circuit; DC low voltage generating means configured to obtain a DC low voltage by performing a rectification operation on an alternating voltage obtained from the secondary winding for use, and a rectification operation on a high voltage obtained from the secondary side boost winding. Is performed, a DC high voltage generating means configured to obtain a DC high voltage, and a series connection circuit of a secondary-side clamp capacitor and a secondary-side auxiliary switching element. Secondary-side active clamp means formed so as to be connected in parallel to the secondary-side parallel resonance circuit; and controlling the conduction angle of the secondary-side auxiliary switching element according to the level of the DC low voltage. A first constant voltage control means for performing constant voltage control; and a series connection circuit of a primary side clamp capacitor and a primary side auxiliary switching element. The series connection circuit is connected to the low voltage primary winding. A primary-side active clamp means formed so as to be connected in parallel to the primary-side auxiliary switching element according to the level of the DC high voltage, thereby performing constant-voltage control. And a second constant voltage control means. 2. A synchronizing means for performing a switching operation in synchronization with a horizontal synchronizing signal frequency for the main switching element based on a signal synchronizing with a horizontal synchronizing signal used in a cathode ray tube display device. Item 2. The switching power supply circuit according to item 1.